W-band three-coil coupling switch transconductance low-noise mixer circuit

Abstract

The invention discloses a W-band three-coil coupling switch transconductance low-noise mixer circuit, and belongs to the technical field of radio frequency integrated circuits, the W-band three-coil coupling switch transconductance low-noise mixer circuit comprises a radio frequency input stage, a transconductance stage, a switch stage and a load stage, the radio frequency input stage comprises inductors Lin, Lg14 and Lg23 and a bias voltage VbRF, and radio frequency signal coupling and broadband impedance matching are realized through a three-winding coupling transformer; the transconductance stage comprises NMOS (N-channel metal oxide semiconductor) tubes M1-M4 and a plurality of groups of inductors, nodes x and y are set, transconductance multiplication and noise suppression are realized by means of reverse coupling of coils, and transconductance symbol periodic flipping is completed through node potential switching; the switch stage comprises an NMOS (N-channel metal oxide semiconductor) tube M5 and inductors Ls and Lg, receives a local oscillator signal to drive conduction, and controls the current path of the transconductance stage to turn over so as to realize down-conversion; and the load stage comprises inductors Ld1-Ld2 and capacitors CL1-CL2, converts the differential current signal into a voltage signal, and outputs an intermediate frequency signal through the LC resonance network. The circuit is suitable for a W-band low-power-consumption low-noise receiver, and the comprehensive performance of broadband matching, low noise and high gain can be achieved.

Description

Technical Field

[0001] This invention belongs to the field of radio frequency integrated circuit technology, specifically relating to a W-band three-coil coupled switching transconductance low-noise mixer circuit. Background Technology

[0002] With the rapid development of millimeter-wave communication, vehicle-mounted radar, and high-frequency wireless sensing systems, the front-end of radio frequency receivers places higher performance demands on mixer circuits, especially in the W-band (75–110 GHz, this invention operates at 80–100 GHz), where the need for low noise, high gain, and low power consumption is becoming increasingly prominent. As a key unit in the radio frequency receiving link, the downconversion mixer's noise figure and power consumption directly determine the system's sensitivity and energy efficiency.

[0003] Looking back, mixers are generally classified into passive and active types. Passive mixers offer good linearity but require high power from the local oscillator signal and suffer from high insertion loss, which is detrimental to signal amplification and transmission in millimeter-wave communication. Especially in the W-band, parasitic effects and high-loss silicon substrates in silicon-based processes make gain performance particularly important. On the other hand, active mixers, with their superior gain performance and port isolation, are highly favored in millimeter-wave applications. It's worth noting that while traditional Gilbert-type mixers can provide high conversion gain and maintain a certain noise level in the low-to-mid frequency or K / Ka bands, in the W-band, the transistor transconductance significantly decreases, and parasitic capacitance and distributed inductance effects increase, leading to a decrease in gain and bandwidth limitation. Simultaneously, the transconductance stage transistors must continuously carry DC bias current, which, when commutated by the switching transistors, significantly increases the output noise of the switching transistors, significantly worsening the mixer's noise figure.

[0004] To improve the noise performance of millimeter-wave active mixers, existing technologies, such as those described in C.-S. Lin, P.-S. Wu, H.-Y. Chang, and H. Wang, “A 9-50-GHz Gilbert-celldown-conversion mixer in 0.13-μm CMOS technology,” IEEE Microw. Wirel. Compon. Lett., vol. 16, no. 5, pp. 293–295, May 2006, doi:10.1109 / LMWC.2006.873492, propose a current injection structure to effectively reduce the bias current flowing through the switching transistor by shunting the current in the injection circuit path while maintaining a constant transconductor current. Correspondingly, the noise of the switching transistor is reduced accordingly. However, it is noted that this technology still achieves a gain of 7dB and a noise figure of 16.4dB in the 50GHz band; the high noise level makes it difficult to promote its application in practice. Furthermore, this technology uses a common balun structure for local oscillator RF port matching. Considering that the transistor gate sees a capacitive load, achieving wideband port matching is difficult, and no experimental results on related matching techniques have been disclosed in the literature. In summary, even with current injection techniques, traditional active mixers still suffer from high noise, and wideband port matching remains a significant technical challenge.

[0005] To address the above problems, this invention proposes a W-band three-coil coupled switching transconductance low-noise mixer circuit. Summary of the Invention

[0006] Based on the above-mentioned technical problems, this application discloses a W-band three-coil coupled switching transconductance low-noise mixer circuit, including an RF input stage, a transconductance stage, a switching stage and a load stage;

[0007] The RF input stage includes inductors Lin, Lg14, and Lg23, as well as a bias voltage VbRF.

[0008] The transconductance stage includes NMOS transistors M1, M2, M3, and M4, and inductors Ls1, Ls2, Ls3, Ls4, Ls0, and Lp0, and the transconductance stage is provided with nodes x and y;

[0009] The switching stage includes an NMOS transistor M5, inductors Ls and Lg, and a bias voltage VbLO.

[0010] The load stage includes inductors Ld1 and Ld2, and capacitors CL1 and CL2.

[0011] Preferably, in the RF input stage, one end of inductor Lin is connected to the RF input terminal and the other end is connected to ground potential; one end of inductor Lg14 is connected to the gate of NMOS transistors M1 and M4 and the other end is connected to the bias voltage VbRF; one end of inductor Lg23 is connected to the gate of NMOS transistors M2 and M3 and the other end is connected to the bias voltage VbRF.

[0012] Inductor Lin is in phase coupled with Lg14, and inductor Lin is in phase coupled with Lg23. After the RF input signal is input through inductor Lin, it is mutually coupled with inductors Lg14 and Lg23 respectively.

[0013] Preferably, in the transconductance stage, the drains of NMOS transistors M1 and M3 are connected to the differential intermediate frequency output terminal IF+; the drains of NMOS transistors M2 and M4 are connected to the differential intermediate frequency output terminal IF-.

[0014] One end of inductor Ls1 is connected to the source of NMOS transistor M1, and the other end is connected to node x; one end of inductor Ls2 is connected to the source of NMOS transistor M2, and the other end is connected to node x; one end of inductor Ls3 is connected to the source of NMOS transistor M3, and the other end is connected to node y; one end of inductor Ls4 is connected to the source of NMOS transistor M4, and the other end is connected to node y.

[0015] Preferably, in the transconductance stage, the center tap of inductor Ls0 is grounded, one end of which is connected to node x and the other end is connected to node y; one end of inductor Lp0 is connected to the drain of NMOS transistor M5 and the other end is connected to the power supply potential.

[0016] Inductors Ls0 and Lp0 are in phase coupled, inductors Lg14 and Ls1 are in reverse coupled, and Lg14 and Ls2 are in reverse coupled; inductors Lg23 and Ls3 are in reverse coupled, and Lg23 and Ls4 are in reverse coupled.

[0017] Preferably, in the switching stage, one end of the inductor Lg is connected to the gate of the NMOS transistor M5, and the other end is connected to the bias voltage VbLO; one end of the inductor Ls is connected to the source of the NMOS transistor M5, and the other end is connected to ground potential.

[0018] Inductors Lg and Ls are coupled in phase. The switching stage receives the local oscillator signal and is driven to conduct alternately, controlling the current path of the transconductance stage to flip, thereby realizing the down-conversion function.

[0019] Preferably, the drain of the NMOS transistor M5 is connected to the midpoint of Ls0, the source is grounded via Ls, and the gate receives the local oscillator signal LO via Lg and sets the DC operating point through the bias voltage VbLO, thereby realizing the dynamic potential reversal of the transconductance nodes x and y.

[0020] Preferably, in the load stage, one end of inductor Ld1 is connected to the power supply potential, and the other end is connected to the differential intermediate frequency output terminal IF+; one end of inductor Ld2 is connected to the power supply potential, and the other end is connected to the differential intermediate frequency output terminal IF-.

[0021] Capacitor CL1 is connected in parallel with inductor Ld1; capacitor CL2 is connected in parallel with inductor Ld2. The load stage converts the differential current signal output from the transconductance stage into a voltage signal, which is then output as an intermediate frequency signal through the resonant network.

[0022] Furthermore, the NMOS transistors M1, M2 and M3, M4 form two sets of differential units, which alternately switch the potentials of nodes x and y to achieve periodic flipping of the transconductance sign, thereby completing the mixing of the radio frequency signal and the local oscillator signal.

[0023] Compared with the prior art, the present invention has the following technical effects:

[0024] This invention employs a multi-winding mutual inductance input network Lin–Lg–Ls to achieve impedance transformation and differential drive of an external 50Ω signal source through magnetic coupling, achieving wideband input matching without relying on additional matching capacitors. Simulation results show that the circuit maintains good port matching performance in the 80–96GHz range, verifying the wideband matching capability of this invention in the W-band.

[0025] In this invention, the RF input signal is induced to the gate of the transconductance transistor via the Lin–Lg multi-winding structure, and then further reverse-coupled to the source of the transconductance transistor through inductors Ls12 and Ls34, realizing the transconductance multiplication effect of the transconductance transistor and significantly improving the gain. Simulation results show that the voltage conversion gain of the circuit at 88GHz is approximately 16dB, which presents a significant technical advantage compared to the low gain level of existing Gilbert-type mixers.

[0026] The reverse coupling of coils Lin–Lg14–Lg23 and Ls12, Ls34, besides generating a transconductance multiplication effect, also suppresses noise in the transconductor, improving the circuit's noise performance. Furthermore, the periodic commutation of the transconductor in this circuit under local oscillator large signal drive at millimeter waves is equivalent to a transconductance curve resembling a sine wave. This differs from the pulsed transconductance switching characteristic of traditional Gilbert mixers. The low harmonic components inevitably reduce the down-conversion effect of thermal noise at the local oscillator harmonic frequency of the RF port, thereby further reducing system noise. The final circuit achieves a low noise figure. Simulation results show that the noise figure of the circuit at an 8GHz intermediate frequency is approximately 3.3dB, a significant advantage over existing technologies.

[0027] This invention achieves broadband matching through a multi-winding mutual inductance network and uses the local oscillator signal to drive the periodic commutation of the large signal across the transconductor to achieve mixing. This structure features low noise, and also boasts significant advantages in gain and power consumption, making it suitable for W-band millimeter-wave receivers and low-power drive applications.

[0028] The above description is only an overview of the technical solution of this application. In order to better understand the technical means of this application and implement it in accordance with the contents of the specification, and to make the above and other objects, features and advantages of this application more obvious and understandable, the preferred embodiments of this application are described in detail below with reference to the accompanying drawings.

[0029] The above and other objects, advantages and features of this application will become more apparent to those skilled in the art from the following detailed description of specific embodiments in conjunction with the accompanying drawings. Attached Figure Description

[0030] To more clearly illustrate the technical solutions in the embodiments of this application or the prior art, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are some embodiments of this application. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort. In all drawings, similar elements or parts are generally identified by similar reference numerals. In the drawings, the elements or parts are not necessarily drawn to scale.

[0031] Based on the description of the figures and their corresponding technical content in the document, the titles of the figures are as follows:

[0032] Figure 1 This is the overall circuit schematic diagram of the low-noise mixer circuit described in this invention;

[0033] Figure 2 This is a schematic diagram of the layout structure of the input transformer of the present invention;

[0034] Figure 3 The graph shows the S-parameter simulation results of the mixer circuit of this invention.

[0035] Figure 4 The noise figure simulation results of the mixer circuit of this invention are shown in the graph.

[0036] Figure 5 This is a simulation curve of the gain of the mixer circuit of the present invention;

[0037] Figure 6 This is a linearity characteristic curve of the mixer circuit of the present invention. Detailed Implementation

[0038] To make the objectives, technical solutions, and advantages of the embodiments of this application clearer, the technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of this application, not all embodiments. In the following description, specific details such as specific configurations and components are provided merely to help fully understand the embodiments of this application. Therefore, those skilled in the art should understand that various changes and modifications can be made to the embodiments described herein without departing from the scope and spirit of this application. In addition, for clarity and brevity, descriptions of known functions and structures are omitted in the embodiments.

[0039] It should be understood that the phrase "an embodiment" or "this embodiment" throughout the specification means that a specific feature, structure, or characteristic related to the embodiment is included in at least one embodiment of this application. Therefore, "an embodiment" or "this embodiment" appearing throughout the specification does not necessarily refer to the same embodiment. Furthermore, these specific features, structures, or characteristics can be combined in any suitable manner in one or more embodiments.

[0040] Furthermore, reference numerals and / or letters may be repeated in different examples within this application. Such repetition is for the purpose of simplification and clarity and does not in itself indicate a relationship between the various embodiments and / or settings discussed.

[0041] In this article, the term "and / or" is merely a description of the relationship between related objects, indicating that three relationships can exist. For example, A and / or B can mean: A exists alone, B exists alone, and A and B exist simultaneously. The term " / and" in this article describes another type of relationship between related objects, indicating that two relationships can exist. For example, A / and B can mean: A exists alone, and A and B exist alone. In addition, the character " / " in this article generally indicates that the related objects before and after it are in an "or" relationship.

[0042] In this article, the term "at least one" is merely a description of the relationship between related objects, indicating that there can be three relationships. For example, "at least one of A and B" can mean: A exists alone, A and B exist simultaneously, or B exists alone.

[0043] It should also be noted that, in this document, relational terms such as "first" and "second" are used only to distinguish one entity or operation from another, and do not necessarily require or imply any such actual relationship or order between these entities or operations. Furthermore, the terms "comprising," "including," or any other variations thereof are intended to cover non-exclusive inclusion.

[0044] Example 1

[0045] like Figure 1 As shown, this embodiment discloses a switched transconductance type low-noise W-band mixer circuit, which achieves multiple functions such as transconductance multiplication and noise suppression by coupling two sets of mutual inductance networks between the RF input stage and the transconductance stage.

[0046] In the RF input stage, inductor Lin and mutual inductors Lg14 and Lg23 form a three-winding coupling transformer. Lin acts as the primary winding, connected to the RF input port RFin. Lg14 and Lg23 act as secondary windings, coupled to the gates of transconductance stage transistors M1, M4 and M2, M3, respectively. When an external RF signal is applied to Lin, the primary current generates a changing magnetic flux. Through mutual inductance, this induces voltages of equal amplitude and phase in Lg14 and Lg23. These voltages are then coupled to the source windings Ls12 and Ls34 in opposite directions, resulting in a transconductance multiplication effect on the transconductors. The formula is:

[0047]

[0048] Taking the case where the left half of the circuit is conducting as an example, The equivalent effective transconductance of a transconductance-level transistor, The transconductance multiplication factor, For the small-signal transconductance of the transistor, Expressed as the turns ratio of Lg14 and Ls12, This represents the coupling coefficient between the two; considering the layout and the transformer's quality factor at millimeter-wave frequencies, a turns ratio of approximately 1:1 is used here. Approximately 0.7; furthermore, the equivalent single-ended impedance seen at the RF port can be approximated as:

[0049]

[0050] in, This is the equivalent single-ended impedance of the RF input port. This refers to the gate-source parasitic capacitance of a transistor. As a complex frequency variable, it can be absorbed by the transformer coil resonance, resulting in a low impedance resistor seen at the RF port, which easily achieves broadband matching with the 50-ohm resistor at the source end. Similarly, a transformer coupling structure is also used at the local oscillator port, resulting in a low impedance and broadband matching with the source port.

[0051] The noise figure of the switching transconductance downconverter mixer disclosed in this invention application It can be characterized as:

[0052]

[0053] In the formula, the parameters This is the noise figure of a transistor, which is related to the transistor's bias state and is also closely related to the transistor's linewidth in the manufacturing process. It is generally beyond the control of circuit designers. This is the gate parasitic resistance of the transconductance-pair transistors (M1, M2). Through proper layout design, this resistance can be minimized. The parallel resonant impedance of the coil at the load is... The internal resistance and noise folding factor of the external radio frequency signal source. This describes the noise downconversion effect caused by LO harmonics. In the formula, the first term describes the noise contribution of the RF port transmitted to the IF output through the LO harmonics; the second, third, and fourth terms represent the noise contributions of the transconductance pair, parasitic gate resistance, and load resistance, respectively. Because this circuit uses a local oscillator sinusoidal signal to excite the transconductor, the equivalent transconductance has a sinusoidal shape and low harmonic characteristics, thus achieving a noise folding factor. Minimizing the noise level reduces the frequency conversion effect of thermal noise at the RF port harmonics. Simultaneously, the formula also shows that the transconductance multiplication effect generated by the transformer coupling in this circuit effectively suppresses noise. Furthermore, the noise from the local oscillator link M5 is canceled out as a common-mode signal at the mixer output. All these effects are the theoretical basis for achieving low noise in this invention.

[0054] This embodiment describes how M5 in the local oscillator link is biased near a preset voltage and operates in Class B state to achieve high efficiency; at the same time, it also ensures the isolation performance of RF-LO; overall, the transformer maintains a differential symmetrical layout to minimize the mismatch caused by the conversion of single-ended to differential signals and reduce the output distortion of the circuit.

[0055] Example 2

[0056] This embodiment is based on Embodiment 1 and further illustrates the performance of the mixer circuit of the present invention with simulation data; the circuit is designed and implemented using a 65nm CMOS process, with an overall power consumption of approximately 6mW and a supply voltage of 0.6V; transistor parameters: M1-4: 50μm / 60nm, M5: 14μm / 60nm; VbRF=0.2V, VbLO=0.3V; as Figure 2 The diagram shows the layout of the input three-coil transformer. The outermost metal layer (AP layer) is the primary coil Lin; the middle metal layer (AP layer) is the secondary coils Lg14 and Lg23; and the innermost metal layer (Metal9) is the third coil Ls12 and Ls34. The central area is the active device layout area, corresponding to transistors M1-M4. The width of the three-coil metal lines is 3μm, and the coil spacing is approximately 2μm. The top metal layer is chosen for the inductor transformer layout to obtain a high quality factor.

[0057] like Figure 3The figure shows the S-parameter simulation results of the mixer of the present invention; where S11 represents the input return loss of the RF port, S22 represents the output return loss of the IF port, and S33 represents the return loss of the local oscillator port; the results show that the circuit has good port matching and isolation performance in the 80–96 GHz frequency band, wherein the input return loss is... Output return loss Port isolation This result verifies the broadband matching effect achieved by the input mutual inductance transformer network, enabling the circuit to maintain stable high gain and excellent impedance matching performance over a wide operating bandwidth.

[0058] like Figure 4 The figure shows the simulation results of the noise figure of the mixer of the present invention. At an intermediate frequency of 8 GHz, the circuit achieves the best noise performance at 88 GHz, with a minimum noise figure of approximately 3.3 dB. Similarly, by maintaining an intermediate frequency of 8 GHz and changing the radio frequency and local oscillator frequency, low noise results were also achieved. This shows that the transconductance multiplication of the transconductance stage has a good effect on noise suppression and ensures the low noise figure of the circuit.

[0059] like Figure 5 The figure shows the frequency response curve of the mixer. The results show that at an intermediate frequency of 8 GHz, the voltage conversion gain at 88 GHz radio frequency is about 16 dB. Similarly, changing the radio frequency and local oscillator frequency while keeping the intermediate frequency constant also yielded similar high gain results. It can be seen that the circuit maintains a flat gain response in the 82-96 GHz radio frequency input range, which verifies the effectiveness of the transformer coupling transconductance multiplication in improving the gain.

[0060] like Figure 6 The figure shows the linearity results of the mixer; the third-order input intermodulation point (IIP3) obtained through two-tone simulation is approximately It exhibits good linearity; as shown in Table 1, simulation results show that the RF-LO isolation of this circuit is 45dB, and the LO-RF isolation is better than 30dB.

[0061] Table 1 Passive Component Parameters

[0062] Table 1 above shows the inductance parameter values ​​of the mixer circuit of the present invention;

[0063] Table 2 Comparison of Performance Parameters

[0064] The prior art listed in Table 2 above is C.-S. Lin, P.-S. Wu, H.-Y. Chang, and H. Wang, “A9-50-GHz Gilbert-celldown-conversion mixer in 0.13-μm CMOS technology,” IEEE Microw. Wirel. Compon. Lett., vol. 16, no. 5, pp. 293–295, May 2006, doi:10.1109 / LMWC.2006.873492. As can be seen from the table above, the present invention is significantly superior to the prior art in terms of noise characteristics, gain performance, and power consumption, demonstrating good application potential.

[0065] The above are merely preferred embodiments of the present invention and are not intended to limit the scope of protection of the present invention. For those skilled in the art, the present invention can have various modifications and variations. Any changes, modifications, substitutions, integrations, and parameter changes made to these embodiments within the spirit and principles of the present invention, without departing from the principles and spirit of the present invention, through conventional substitutions or to achieve the same function, shall fall within the scope of protection of the present invention.

Claims

1. A W-band three-coil coupled switching transconductance low-noise mixer circuit, characterized in that, It includes the RF input stage, transconductance stage, switching stage, and load stage; The RF input stage includes inductors Lin, Lg14, and Lg23, as well as a bias voltage VbRF. The transconductance stage includes NMOS transistors M1, M2, M3, and M4, and inductors Ls1, Ls2, Ls3, Ls4, Ls0, and Lp0, and the transconductance stage is provided with nodes x and y; The switching stage includes an NMOS transistor M5, inductors Ls and Lg, and a bias voltage VbLO. The load stage includes inductors Ld1 and Ld2, and capacitors CL1 and CL2.

2. The W-band three-coil coupled switching transconductance low-noise mixer circuit according to claim 1, characterized in that, In the RF input stage, one end of inductor Lin is connected to the RF input terminal, and the other end is connected to ground potential; one end of inductor Lg14 is connected to the gate of NMOS transistors M1 and M4, and the other end is connected to the bias voltage VbRF; one end of inductor Lg23 is connected to the gate of NMOS transistors M2 and M3, and the other end is connected to the bias voltage VbRF. Inductor Lin is in phase coupled with Lg14, and inductor Lin is in phase coupled with Lg23. After the RF input signal is input through inductor Lin, it is mutually coupled with inductors Lg14 and Lg23 respectively.

3. The W-band three-coil coupled switching transconductance low-noise mixer circuit according to claim 1, characterized in that, In the transconductance stage, the drains of NMOS transistors M1 and M3 are connected to the differential intermediate frequency output terminal IF+; the drains of NMOS transistors M2 and M4 are connected to the differential intermediate frequency output terminal IF-. One end of inductor Ls1 is connected to the source of NMOS transistor M1, and the other end is connected to node x; one end of inductor Ls2 is connected to the source of NMOS transistor M2, and the other end is connected to node x; one end of inductor Ls3 is connected to the source of NMOS transistor M3, and the other end is connected to node y; one end of inductor Ls4 is connected to the source of NMOS transistor M4, and the other end is connected to node y.

4. The W-band three-coil coupled switching transconductance low-noise mixer circuit according to claim 3, characterized in that, In the transconductance stage, the center tap of inductor Ls0 is grounded, one end of which is connected to node x and the other end is connected to node y; one end of inductor Lp0 is connected to the drain of NMOS transistor M5 and the other end is connected to the power supply potential. Inductors Ls0 and Lp0 are in phase coupled, inductors Lg14 and Ls1 are in reverse coupled, and Lg14 and Ls2 are in reverse coupled; inductors Lg23 and Ls3 are in reverse coupled, and Lg23 and Ls4 are in reverse coupled.

5. The W-band three-coil coupled switching transconductance low-noise mixer circuit according to claim 1, characterized in that, In the switching stage, one end of inductor Lg is connected to the gate of NMOS transistor M5, and the other end is connected to the bias voltage VbLO; one end of inductor Ls is connected to the source of NMOS transistor M5, and the other end is connected to ground potential. Inductors Lg and Ls are coupled in phase. The switching stage receives the local oscillator signal and is driven to conduct alternately, controlling the current path of the transconductance stage to flip, thereby realizing the down-conversion function.

6. The W-band three-coil coupled switching transconductance low-noise mixer circuit according to claim 5, characterized in that, The drain of the NMOS transistor M5 is connected to the midpoint of Ls0, the source is grounded via Ls, and the gate receives the local oscillator signal LO via Lg and sets the DC operating point through the bias voltage VbLO, thereby realizing the dynamic potential reversal of the transconductance nodes x and y.

7. The W-band three-coil coupled switching transconductance low-noise mixer circuit according to claim 1 or 3, characterized in that, In the load stage, one end of inductor Ld1 is connected to the power supply potential, and the other end is connected to the differential intermediate frequency output terminal IF+; one end of inductor Ld2 is connected to the power supply potential, and the other end is connected to the differential intermediate frequency output terminal IF-. Capacitor CL1 is connected in parallel with inductor Ld1; capacitor CL2 is connected in parallel with inductor Ld2. The load stage converts the differential current signal output from the transconductance stage into a voltage signal, which is then output as an intermediate frequency signal through the resonant network. Furthermore, the NMOS transistors M1, M2 and M3, M4 form two sets of differential units, which alternately switch the potentials of nodes x and y to achieve periodic flipping of the transconductance sign, thereby completing the mixing of the radio frequency signal and the local oscillator signal.

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