Low-power-consumption full-step-up-ratio boost circuit and control method thereof

Abstract

The invention relates to the technical field of direct current boost, in particular to a boost circuit with low power consumption and a full step-up ratio and a control method thereof. Synchronous on-off of the main switch tube and the auxiliary switch tube is controlled, half-cycle free oscillation of the resonant capacitor, the auxiliary capacitor and the resonant inductor is combined, the voltage of the resonant capacitor is forced to reverse and is stabilized at the input voltage Vin, and it is ensured that the drain-source voltage of the main switch tube gradually rises from zero to the output voltage Vout at the moment that the main switch tube is turned off; the turn-off voltage impact and the switching loss are thoroughly eliminated, the effect is not influenced by the step-up ratio value, and the technical problem that the ZVS is difficult to realize under the low step-up ratio of a traditional boost circuit is solved.

Description

Technical Field

[0001] This invention relates to the field of DC boost technology, and in particular to a low-power full boost ratio boost circuit and its control method. Background Technology

[0002] New energy sources all require DC-DC conversion, and the boost circuit is the most commonly used circuit. The boost circuit can increase the DC voltage by one to several times.

[0003] Existing boost ZVS boost circuits, such as Figure 1a As shown, a boost ratio >= 3 is required to guarantee complete ZVS during Q turn-off. Figure 1b The diagram shows the voltage-current plot of Q. Theoretically, the turn-off power consumption is 0, but when the boost ratio is less than 3, the voltage-current plot of Q during turn-off is as follows: Figure 1c As shown, there is an initial voltage Vo across Vds of Q, so Q still has turn-off power consumption.

[0004] Therefore, overcoming the shortcomings of the existing technology is an urgent problem to be solved in this technical field. Summary of the Invention

[0005] The technical problem to be solved by this invention is how to solve the problem of additional shutdown power consumption that is easily caused by the boost ratio in traditional boost ZVS boost circuits.

[0006] The present invention adopts the following technical solution: In the first aspect, a low-power full boost ratio boost circuit is provided, including an energy storage inductor, a main switching transistor and a main diode, as well as a resonant inductor, a resonant capacitor, an auxiliary capacitor, an auxiliary switching transistor, a second diode, a third diode and a fourth diode. One end of the energy storage inductor is connected to the positive input terminal and the cathode of the second diode, respectively, and the other end of the energy storage inductor is connected to the anode of the main diode, the anode of the fourth diode, one end of the resonant capacitor, and one end of the main switch. The cathode of the main diode is connected to the positive output terminal; the anode of the second diode is connected to the other end of the resonant capacitor to form a first connection node; one end of the auxiliary capacitor is connected to the cathode of the fourth diode to form a second connection node; the resonant inductor, the third diode, and the auxiliary switch are connected in series between the first connection node and the second connection node; the other end of the main switch and the other end of the auxiliary capacitor are respectively connected to the negative input terminal.

[0007] Preferably, the drain of the auxiliary switch is connected to one end of the fourth diode and one end of the auxiliary capacitor, respectively, and the source of the auxiliary switch is connected to one end of the resonant inductor.

[0008] Preferably, the drain of the auxiliary switch is connected to one end of the resonant inductor, and the source of the auxiliary switch is connected to the anode of the third diode.

[0009] Preferably, the drain of the auxiliary switch is connected to the cathode of the third diode, and the source of the auxiliary switch is connected to the anode of the second diode and one end of the resonant capacitor, respectively.

[0010] Preferably, the conduction timing of the auxiliary switch is consistent with that of the main switch.

[0011] Preferably, the capacitance of the auxiliary capacitor is greater than or equal to three times the capacitance of the resonant capacitor.

[0012] Preferably, in the LC resonant circuit formed by the resonant inductor, the resonant capacitor, and the auxiliary capacitor, half of its resonant period is less than the conduction time of the main switch.

[0013] Preferably, the output voltage is greater than the input voltage but less than twice the input voltage.

[0014] Preferably, the current of the energy storage inductor drops to 0 in each PWM operating cycle, so that the low-power full boost ratio boost circuit operates in discontinuous current mode.

[0015] Secondly, a control method for a low-power full boost ratio boost circuit is provided, for controlling the low-power full boost ratio boost circuit as described in the first aspect, the control method comprising: The main switch and the auxiliary switch are turned off by control, and the energy storage inductor charges the resonant capacitor and the auxiliary capacitor, so that the voltage of the resonant capacitor reaches Vout-Vin and the voltage of the auxiliary capacitor reaches Vout. The main switch and the auxiliary switch remain off, the current in the energy storage inductor drops to zero, the voltage on the resonant capacitor remains Vout-Vin, and the voltage on the auxiliary capacitor remains Vout. The main switch and the auxiliary switch are controlled to be turned on, so that the LC resonant circuit including the resonant capacitor, the auxiliary capacitor and the resonant inductor oscillates freely for half a cycle, so that the voltage on the resonant capacitor flips in the reverse direction from Vout-Vin to Vin, and the voltage on the auxiliary capacitor drops to zero, so that the main switch can achieve zero current conduction. Keep the main switch and the auxiliary switch on, so that the energy storage inductor absorbs energy from the input voltage and the current rises linearly; The main switch and the auxiliary switch are turned off by controlling the main switch and the auxiliary switch. The voltage across the resonant capacitor is Vin, and the drain-source voltage of the main switch rises from zero, thereby achieving zero-voltage turn-off. Here, Vin is the input voltage and Vout is the output voltage.

[0016] Compared with the prior art, the beneficial effects of the present invention are as follows: This invention controls the synchronous switching of the main switch and the auxiliary switch, combined with the half-cycle free oscillation of the resonant capacitor, auxiliary capacitor and resonant inductor, to force the resonant capacitor voltage to reverse and stabilize at the input voltage Vin. This ensures that the drain-source voltage of the main switch gradually rises from zero to the output voltage Vout at the moment of turn-off, completely eliminating turn-off voltage surges and switching losses. Moreover, this effect is not affected by the boost ratio, solving the technical pain point of traditional boost circuits that are difficult to achieve ZVS at low boost ratios. Attached Figure Description

[0017] To more clearly illustrate the technical solutions in the embodiments of the present invention or the prior art, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0018] Figure 1a This is a schematic diagram of a conventional boost ZVS boost circuit provided in an embodiment of the present invention; Figure 1b This is a waveform diagram of a boost ZVS boost circuit provided in an embodiment of the present invention; Figure 1c This is another waveform diagram of a boost ZVS boost circuit provided in an embodiment of the present invention; Figure 1 This is a schematic diagram of a conventional boost converter circuit provided in an embodiment of the present invention; Figure 2 This is a waveform diagram of a conventional boost circuit provided in an embodiment of the present invention; Figure 3 This is a schematic diagram of a high-efficiency boost circuit provided in an embodiment of the present invention; Figure 4 This is a waveform diagram of a high-efficiency boost circuit provided in an embodiment of the present invention; Figure 5This is a circuit and waveform diagram of stage one provided in an embodiment of the present invention; Figure 6 This is a schematic diagram of a circuit and waveforms for stage two provided in an embodiment of the present invention; Figure 7 This is a circuit and waveform diagram of stage three provided in an embodiment of the present invention; Figure 8 This is a circuit and waveform diagram of stage five provided in an embodiment of the present invention; Figure 9 This is a schematic diagram of the voltage and current waveforms of the main switching transistor in a high-efficiency boost circuit provided by an embodiment of the present invention; Figure 10 This is a schematic diagram of a circuit for cutting off the discharge of a resonant capacitor provided in an embodiment of the present invention; Figure 11 This is a schematic diagram showing the setting position of a flow interception unit according to an embodiment of the present invention; Figure 12 This is a circuit diagram of a diode and a MOSFET at position P1 provided in an embodiment of the present invention; Figure 13 This is a circuit diagram of a diode and MOSFET at position P2 provided in an embodiment of the present invention; Figure 14 This is a circuit diagram of a diode and MOSFET at position P3 provided in an embodiment of the present invention; Figure 15 This is a schematic diagram of a high-efficiency boost circuit with MOSFET T1 placed at position P4, provided by an embodiment of the present invention. Figure 16 This is a schematic diagram of a high-efficiency boost circuit with MOSFET T1 placed at position P8, provided by an embodiment of the present invention. Figure 17 This is a schematic diagram of a high-efficiency boost circuit with MOSFET T1 placed at position P5, provided by an embodiment of the present invention. Figure 18 This is a schematic diagram of a high-efficiency boost circuit with MOSFET T1 placed at position P7, provided by an embodiment of the present invention. Figure 19 This is a schematic diagram of a high-efficiency boost circuit with MOSFET T1 placed at position P6, provided by an embodiment of the present invention. Figure 20 This is a flowchart illustrating a control method for a high-efficiency boost circuit provided in an embodiment of the present invention; Figure 21 This is a schematic diagram of a low-power full boost ratio boost circuit provided in an embodiment of the present invention; Figure 22This is a schematic diagram of a second structure of a low-power full boost ratio boost circuit provided in an embodiment of the present invention; Figure 23 This is a schematic diagram of the third structure of a low-power full boost ratio boost circuit provided in an embodiment of the present invention; Figure 24 This is a schematic diagram of a waveform structure in stage one provided by an embodiment of the present invention; Figure 25 This is a schematic diagram of a waveform structure for stage two provided in an embodiment of the present invention; Figure 26 This is a schematic diagram of a waveform structure in stage three provided by an embodiment of the present invention; Figure 27 This is a schematic diagram of a waveform structure in stage four provided by an embodiment of the present invention; Figure 28 This is a schematic diagram of a waveform structure for stage five provided in an embodiment of the present invention; Figure 29 This is a schematic diagram of a current waveform from t3 to t4 provided in an embodiment of the present invention; Figure 30 This is a schematic diagram of the equivalent circuit structure at time t3 provided in an embodiment of the present invention; Figure 31 This is a schematic diagram of the equivalent circuit structure at time t4 provided in an embodiment of the present invention; Figure 32 This is a flowchart illustrating a control method for a low-power full boost ratio boost circuit provided in an embodiment of the present invention. Detailed Implementation

[0019] To make the objectives, technical solutions, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and not intended to limit the invention.

[0020] Unless the context otherwise requires, throughout the specification and claims, the term "comprising" is interpreted as openly inclusive, meaning "including, but not limited to." In the description of the specification, terms such as "one embodiment," "some embodiments," "exemplary embodiment," "example," "specific example," or "some examples" are intended to indicate that a particular feature, structure, material, or characteristic associated with that embodiment or example is included in at least one embodiment or example of this disclosure. The illustrative representations of the above terms do not necessarily refer to the same embodiment or example. Furthermore, the specific features, structures, materials, or characteristics mentioned may be included in any suitable manner in any one or more embodiments or examples; that is, although they may be incorporated into embodiments or examples using the above terms for reasons such as order and position, it does not limit them to be incorporated in combination by a single embodiment or example.

[0021] In the description of this invention, the terms "first" and "second" are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of indicated technical features. Thus, a feature defined with "first" or "second" may explicitly or implicitly include one or more of that feature. In the description of embodiments of this disclosure, unless otherwise stated, "a plurality of" means two or more. Furthermore, for example, the description may use the prefix "A" or "B" to describe the same type of nouns as two independent entities. In this case, the corresponding features defined with "A" and "B" are used only to distinguish between similar entities and should not be construed as indicating or implying relative importance or implicitly specifying the number of indicated technical features.

[0022] In describing some embodiments, the terms "coupled," "coupled," and "connected," and their derivative expressions, may be used. For example, the term "connected" may be used in describing some embodiments to indicate that two or more components have direct physical or electrical contact with each other. Similarly, the term "coupled" may be used in describing some embodiments to indicate that two or more components have direct physical or electrical contact. However, the terms "connected" or "coupled" may also refer to two or more components that do not have direct contact with each other but still cooperate or interact with each other, such as "optical coupling," "wireless connection," etc. The embodiments disclosed herein are not necessarily limited to the scope of this invention.

[0023] Furthermore, the technical features involved in the various embodiments of the present invention described below can be combined with each other as long as they do not conflict with each other.

[0024] Example 1: Before describing the low-power full boost ratio boost circuit, this embodiment first introduces a high-efficiency boost circuit.

[0025] In traditional boost converter circuits, the MOSFETs are hard-switched during both turn-on and turn-off, resulting in switching losses being the primary source of power loss and thus limiting the efficiency of the traditional boost converter circuit. In one embodiment, such as... Figure 1 The diagram shows a standard boost converter circuit, such as... Figure 2 The diagram shows the waveforms of the MOSFET current iT, voltage Vds, drive voltage Vgs, and inductor current iL when the boost circuit operates in DCM (Discontinuous Current Mode). Time t1 is the MOSFET turn-off moment. During turn-off, the MOSFET current decreases and the voltage increases, with a crossover time between the current and voltage. The power i×V during this time is the turn-off power loss. If we approximate the changes in i and V as linear, the power consumption during turn-off is approximately W=I×V / 2. If I=40A and V=400V, the turn-off power consumption is approximately W=40×400 / 2=8000W. Assuming the turn-off time is 0.1µs and the MOSFET's PWM period is 20µs, since the operating state is DCM and the turn-on is ZCS (Zero Current Turn-On), the turn-on loss is 0. Assuming the on-resistance is 0, the conduction loss is 0. The average power consumption per cycle consists only of the turn-off loss, and the average MOSFET loss is 8000 / 200=40 watts.

[0026] To address the aforementioned problems, this embodiment proposes a high-efficiency boost converter circuit. In one embodiment, such as... Figure 3 As shown, it includes an energy storage inductor (i.e., L), a main switch (i.e., T), and a main diode (i.e., D), as well as a resonant inductor (i.e., Lr), a resonant capacitor (i.e., Cr), a current-cutting unit (i.e., D1), a second diode (i.e., D2), and a third diode (i.e., D3).

[0027] One end of the energy storage inductor is connected to the positive input terminal, and the other end of the energy storage inductor is connected to the anode of the main diode, one end of the resonant capacitor, and one end of the main switch. The cathode of the main diode is connected to the positive output terminal. The other end of the resonant capacitor is connected to the anode of the second diode and one end of the resonant inductor. The cathode of the second diode is connected to the positive input terminal. The other end of the resonant inductor is connected to the cathode of the third diode. The anode of the third diode and the other end of the main switch are connected to the negative input terminal.

[0028] It is worth noting that since the current-cutting unit can be set in multiple locations, the current-cutting unit is not included in the above connection relationship. The above connection relationship describes the connection relationship between components whose relative positions remain unchanged.

[0029] Reference Figure 11As shown, the current-cutting unit is connected in series at any position in a series circuit including the input power supply, the energy storage inductor, the resonant capacitor, the resonant inductor, and the third diode; through the synergistic effect of the resonant capacitor, the resonant inductor, the current-cutting unit, the second diode, and the third diode, the main switch can achieve zero current conduction when turned on and zero voltage turn-off when turned off in high-frequency PWM operating mode.

[0030] In one embodiment, the high-efficiency boost circuit further includes an output capacitor (i.e., C) and a load resistor (i.e., R), one end of the output capacitor being connected to the cathode of the main diode and the other end of the output capacitor being connected to the negative input terminal; one end of the load resistor being connected to the cathode of the main diode and the other end of the load resistor being connected to the negative input terminal.

[0031] In one embodiment, based on the structure listed above, referring to Figure 3 A current-cutting unit is connected in series between the energy storage inductor and the main switching transistor, and a diode is used as an example of the current-cutting unit. This embodiment first introduces the working principle of the high-efficiency boost circuit. This embodiment divides its working principle into at least 5 stages. In one embodiment, such as... Figure 4 The diagram shown is a schematic of the main waveforms of the high-efficiency boost circuit, wherein: Phase 1 (t1-t2): such as Figure 5 As shown, when the main switch is turned off, the current of the energy storage inductor decreases, and the voltage of the energy storage inductor is Vout-Vin (where Vout is the output voltage and Vin is the input voltage). This process is to boost the voltage and transfer energy to the output capacitor and the load resistor. The voltage of the resonant capacitor is Vcr=Vout-Vin. Assume that Vcr>Vin.

[0032] Phase 2 (t2-t3): such as Figure 6 As shown, the main switch is still off. At this time, the current iL on the energy storage inductor is 0. Due to the first diode (i.e., the current-cutting unit, here we take the diode, which is connected in series between the energy storage inductor and the main switch), the voltage on the resonant capacitor remains unchanged, Vcr=Vout-Vin.

[0033] Phase 3 (t3-t4): such as Figure 7As shown, when the main switch is turned on, the current in the energy storage inductor begins to increase. At this time, the resonant capacitor and the resonant inductor form an LC resonant circuit through the third diode and begin to discharge. After half a resonant cycle, the voltage polarity of the resonant capacitor reverses, and its voltage value, due to the clamping effect of the second diode, is Vcr = -Vin ((Vout - Vin) >= Vin). At this time, the current of the main switch slowly increases from 0, while the voltage drops rapidly to 0. Therefore, the main switch is ZCS zero-current turn-on, and the turn-on power consumption is 0.

[0034] Phase 4 (t4-t5): The main switch remains on, the current in the energy storage inductor increases linearly, and the voltage Vcr = -Vin on the resonant capacitor remains unchanged.

[0035] Phase 5 (t5-t6): such as Figure 8 As shown, with the main switch off, since Vcr = -Vin, the energy storage inductor generates a current iL through the resonant capacitor. At this time, the voltage across the resonant capacitor gradually changes from -Vin to Vout. Simultaneously, the drain voltage across the main switch gradually rises from 0V to Vout, while the current of the main switch rapidly decreases to 0. (See...) Figure 9 Since the current fall time is a fixed value and the voltage rise time Vds is longer, the voltage value in the crossover range is lower, thus reducing the turn-off power consumption.

[0036] After Phase 5, we re-enter Phase 1 and begin the next cycle.

[0037] In one embodiment, Figure 9 To illustrate the voltage and current waveforms of the main switching transistor in a high-efficiency boost circuit, for example, if the current turn-off time is still 100ns, but the voltage rise time is delayed from 100ns to 1us, assuming the current and voltage waveforms change linearly, the turn-off power is I×(V / 10) / 2=40×(400 / 10) / 2=20×40=800W, and the average power is 800 / 200=4W. Therefore, the turn-off loss is reduced from 40W to 4W.

[0038] In one embodiment, the function of the current-cutting unit is to make the turn-off starting voltage of the main switch equal to 0. The current-cutting unit is connected in series at any position in the series circuit consisting of input power supply-energy storage inductor-resonant capacitor-resonant inductor-third diode, without affecting the on / off state of the main circuit.

[0039] In one embodiment, the key function of the current-cutting unit is to block improper discharge of the resonant capacitor. Specifically, in stage two, when the energy storage inductor current drops to zero, the presence of the second diode prevents the resonant capacitor from discharging through the energy storage inductor, input power supply, third diode, and resonant inductor (preventing the energy storage inductor current from becoming negative). This ensures that the voltage of the resonant capacitor reverses in stage three and reaches -Vin. Consequently, when the main switch is turned off in stage five, the drain voltage of the main switch, due to the presence of the resonant capacitor voltage -Vin, decreases from 0 ( Figure 9 The t5 in the middle slowly rises to Vout, reducing the turn-off loss of the main switch.

[0040] In one embodiment, if there is no current-cutting unit (corresponding to the removal of diode D1 in the above example), then in stage two, the discharge circuit of the resonant capacitor is as follows: Figure 10 As shown, after the discharge in stage two, the voltage on the resonant capacitor does not reach -Vin. Therefore, at time t5 in stage five, the Vds of the main switch transistor has an initial voltage V0. Figure 10 The starting voltage V0 (t5 in the equation) increases the turn-on power consumption of the main switch, thus weakening the ZVS function. The magnitude of the starting voltage V0 is related to Vin and Vout. Generally, the starting voltage V0 is greater than 0. The magnitude of the starting voltage V0 can be calculated using the following formula: When Vout < 2 × Vin, V0 = 2 × Vin - Vout; when 2 × Vin <= Vout < 3 × Vin, V0 = Vout - 2 × Vin; when 3 × Vin <= Vout < 4 × Vin, V0 = 4 × Vin - Vout; when 4 × Vin <= Vout, V0 = Vin.

[0041] After adding the cutoff unit, when Vout>=2×Vin, V0=0; when Vout<2×Vin, V0=2×Vin-Vout.

[0042] In one embodiment, the current-cutting unit is a diode or a MOSFET; when the current-cutting unit is connected in series between the positive input terminal and the energy storage inductor, the current-cutting unit is a diode or a MOSFET; when the current-cutting unit is connected in series between the energy storage inductor and the main switching transistor, the current-cutting unit is a diode or a MOSFET; when the current-cutting unit is connected in series between the energy storage inductor and the resonant capacitor, the current-cutting unit is a MOSFET; when the current-cutting unit is connected in series between the resonant capacitor and the resonant inductor, the current-cutting unit is a MOSFET; when the current-cutting unit is connected in series between the resonant inductor and the third diode, the current-cutting unit is a MOSFET; when the current-cutting unit is connected in series between the third diode and the negative input terminal, the current-cutting unit is a MOSFET.

[0043] Among them, from Figure 10 The flowchart shows that there are 8 positions to cut off the discharge current of the resonant capacitor, P1-P8. Figure 11 As shown. Positions P1-P3 can use either diodes or MOSFETs, while positions P4-P8 can only use MOSFETs. Figure 11 The thick black line at the top represents the high-current path. P1-P3 are on the high-current path, so the device losses are relatively large. Points P4-P8 are on the branch, where the current is very small, so the power consumption of the MOSFET is small. The direction of the MOSFET is the direction of the red arrow in the diagram. Setting the MOSFET at the five points P4-P8 has the same effect and the same power consumption. The specific position to choose can be considered from the perspective of driver design. Choose the position that is simple to drive.

[0044] In one embodiment, when the current-cutting unit is a diode, the conduction direction of the current-cutting unit is opposite to the reverse current direction of the resonant capacitor discharge in the series circuit.

[0045] The series circuit refers to the improper discharge circuit that the resonant capacitor may form during the stage when the current of the energy storage inductor returns to zero. Its path is: resonant capacitor → energy storage inductor → input voltage → third diode → resonant inductor → resonant capacitor. The reverse current of the resonant capacitor discharge is the current that releases energy from the resonant capacitor in the above improper discharge circuit (the direction is opposite to the resonant discharge current when the subsequent main switch is turned on, so it is a reverse current). The core function of the current-cutting unit is to block this reverse current, ensure that the voltage of the resonant capacitor is stable at Vout-Vin, and avoid voltage drop due to discharge, thereby destroying the ZVS condition.

[0046] In one embodiment, the diode has the characteristics of forward conduction and reverse cutoff. To block the aforementioned reverse current, the conduction direction of the diode must be opposite to the direction of the reverse current. At this time, when the reverse current flows through the diode, the diode is in a reverse bias state and is in a cutoff state, thereby cutting off the improper discharge circuit of the resonant capacitor. In another embodiment, if the conduction direction is the same as the reverse current direction, the diode will conduct forward, and the resonant capacitor will discharge rapidly through this circuit. The voltage cannot be maintained at Vout-Vin, and the ZVS condition (V0=0) when the main switch is turned off will fail.

[0047] In one embodiment, when the current-cutting unit is a MOS transistor, the switching control timing of the current-cutting unit is synchronized with the switching control timing of the main switching transistor.

[0048] In this embodiment, the current-cutting MOSFET (i.e., the current-cutting unit) turns on simultaneously at the instant the main switch is turned on. When the main switch is turned on, the resonant capacitor and resonant inductor need to form a complete resonant circuit. The synchronous turn-on of the current-cutting MOSFET can open the path from the resonant capacitor to the resonant inductor to the third diode to the main switch and then to the current-cutting MOSFET, allowing the electric field of the resonant capacitor to be released smoothly and the voltage polarity to be reversed, so that the current of the main switch starts from 0, thus realizing ZCS. In one embodiment, if the current-cutting MOSFET turns on late, it will cause the resonant circuit to be open, the resonant process cannot start, and ZCS will fail.

[0049] After the main switch is turned off, the energy storage inductor current returns to zero, and the resonant capacitor is charged to Vout-Vin. The synchronous turn-off of the current-cutting MOSFET can cut off the discharge circuit of Cr, preventing its voltage drop and ensuring that Cr can gradually rise from -Vin to Vout during the subsequent turn-off phase, providing a stable initial condition for the main switch's ZVS. If the turn-off of the current-cutting MOSFET is delayed, the resonant capacitor will discharge through its leakage current, the voltage cannot be maintained, and the ZVS condition is destroyed.

[0050] Meanwhile, the branch containing the current-cutting MOSFET is a low-current branch (average current ≤1.5A), and the timing design synchronized with the main switch does not require additional complex control, and can minimize branch losses, taking into account both circuit stability and efficiency.

[0051] In one embodiment, Figure 12 , Figure 13 and Figure 14 The circuit diagrams are for diodes and MOSFETs located at positions P1-P3, respectively.

[0052] Figure 12 , Figure 13 and Figure 14 In the right-hand diagram, the diode in the left-hand diagram is replaced with a MOS switch T1. The switching control of T1 is completely synchronized with the main switch, which can reduce the power consumption caused by the diode junction voltage drop.

[0053] Although Figure 12 , Figure 13 and Figure 14 Replacing the diode with a MOSFET T1 can reduce power consumption, but Figure 12 , Figure 13 and Figure 14 Since MOSFETs are located in circuits where large currents flow, the overall power consumption reduction depends on the performance of the MOSFET switching transistor. High-performance MOSFETs are also very expensive, making them uneconomical in practical use.

[0054] Figure 11Placing MOSFETs at positions P4-P8 removes the high-current path from these five locations, achieving the same effect as placing diodes or MOSFETs at positions P1-P3. However, the average current on this branch is only 1.5A, far lower than the average current of 12A in the main circuit. Therefore, the requirements for the parameters of the MOSFETs are very low, which can reduce the cost of the ZVS circuit. At the same time, the losses introduced by the MOSFETs are also very small. In actual production, one of the five points P4-P8 is selected to place a MOSFET to achieve low power consumption.

[0055] In one embodiment, such as Figure 15 and Figure 16 The diagram shows the schematic of a high-efficiency boost circuit with MOSFET T1 placed at points P4 and P8 respectively.

[0056] In one embodiment, such as Figure 17 and Figure 18 The diagram shows the schematic of a high-efficiency boost circuit with MOSFET T1 placed at points P5 and P7 respectively.

[0057] In one embodiment, such as Figure 19 The diagram shows the schematic of a high-efficiency boost circuit with MOSFET T1 placed at P6.

[0058] To ensure the normal operation of the high-efficiency boost circuit, in one embodiment, half of the resonant period of the LC resonant circuit formed by the resonant inductor and the resonant capacitor is less than the conduction time of the main switch.

[0059] In the LC resonant circuit, the reversal of the voltage polarity of the resonant capacitor must rely on a complete half-resonant cycle (the complete conversion of electric field energy → magnetic field energy → reverse electric field energy). This is the essential law of LC resonance and the prerequisite for the main switch to achieve zero current conduction (ZCS).

[0060] In one embodiment, the core objective of turning on the main switch is to allow the resonant capacitor to complete polarity reversal from the initial voltage Vout-Vin (Vout is the output voltage, Vin is the input voltage) and be clamped to -Vin by the second diode. If half a resonant period is greater than or equal to the main switch's on-time, the resonant process will not be completed (the capacitor voltage will not reverse or will not reverse sufficiently), the main switch current will not be able to start from 0, ZCS will fail, and the subsequent zero-voltage turn-off (ZVS) condition will be violated.

[0061] Furthermore, in one embodiment, the turn-on timing and turn-off timing of the main switch are synchronized with the resonance phase of the LC resonant circuit, wherein: the turn-on trigger timing of the main switch is when the current iL of the energy storage inductor is 0; the turn-off trigger timing of the main switch is when the voltage of the resonant capacitor stabilizes at -Vin, where Vin is the input voltage.

[0062] Specifically, the turn-on triggering time of the main switch is locked at the current iL of the energy storage inductor = 0. At this time, the circuit is at the end of stage two. The energy storage inductor has completed energy release, and there is no free current after the current drops to 0. When the main switch is turned on, it will not encounter the impact of reverse free current. At the same time, the voltage of the resonant capacitor has been charged to Vout-Vin in stage one and is kept stable by the action of the current-cutting unit, providing the ideal start-up conditions of constant voltage and initial current of 0 for the LC resonant circuit.

[0063] The turn-off trigger of the main switch is set when the voltage of the resonant capacitor stabilizes at -Vin. At this time, the circuit is in the early stage of stage four. The LC resonant circuit has completed half a cycle of oscillation. The voltage of the resonant capacitor has completed the polarity reversal from Vout to Vin and is clamped to -Vin by the second diode (the voltage is stable and there is no fluctuation). At the same time, the energy storage inductor has completed linear energy storage during the conduction of the main switch and the current is in a stable growth state. After the turn-off, the energy of the energy storage inductor can continue to flow through the LC resonant circuit.

[0064] In one embodiment, the current of the energy storage inductor drops to 0 in each PWM duty cycle, so that the high-efficiency boost circuit operates in discontinuous current mode.

[0065] For the BOOST boost circuit, the criterion for determining the discontinuous current mode is that the current of the energy storage inductor can drop to 0 in each PWM working cycle. This is different from the continuous current mode (CCM), where the current of the energy storage inductor does not drop to 0 and there is always freewheeling current, which cannot provide a stable initial operating condition for soft switching.

[0066] It is worth noting that in the above scheme, the output voltage is greater than or equal to twice the input voltage. The LC resonant circuit includes a resonant capacitor and a resonant inductor. The reversal of the voltage polarity of the resonant capacitor (from VCr = Vout - Vin to VCr = -Vin) depends entirely on its initial stored energy.

[0067] According to the law of conservation of energy, the minimum energy condition for a resonant capacitor to complete a polarity reversal is: 1 / 2 × Cr × (Vout - Vin) 2 ≥1 / 2Cr×Vin 2 After simplification, we get: Vout-Vin≥Vin, that is, Vout≥2Vin. Therefore, the output voltage is greater than or equal to twice the input voltage.

[0068] In one embodiment, if Vout < 2Vin (corresponding to a boost ratio of less than 2), the initial energy storage of the resonant capacitor is insufficient to support the polarity reversal of half a resonant cycle, and ultimately the voltage of the resonant capacitor cannot stabilize to -Vin. This directly leads to the simultaneous failure of the zero-current turn-on (ZCS) and zero-voltage turn-off (ZVS) conditions of the main switch, causing the circuit to degenerate into a hard-switching mode, and the switching losses at high frequencies to increase significantly. The explanation for a boost ratio of less than 2 will be provided in detail in subsequent embodiments.

[0069] This embodiment introduces an auxiliary resonant network consisting of a resonant capacitor, a resonant inductor, and multiple diodes into the traditional Boost circuit. Utilizing the LC resonance characteristics of the resonant capacitor and inductor, the current of the main switch increases from zero when it is turned on, and the voltage rises slowly and the current quickly returns to zero when it is turned off. This avoids switching losses caused by the overlap of current and voltage, solving the technical problem of large current and voltage overlap losses and low boost efficiency caused by hard switching of the main switch in traditional high-efficiency boost circuits. It achieves zero-voltage turn-off and zero-current turn-on of the main switch, significantly reducing switching losses and improving the circuit's operational stability and energy conversion efficiency under high-frequency conditions.

[0070] Example 2: This embodiment proposes a control method for a high-efficiency boost circuit. In one embodiment, such as... Figure 20 As shown, the control method includes: Step 101: Turn off the main switch, the energy storage inductor releases energy, and the resonant capacitor charges to Vout-Vin; the current of the energy storage inductor drops to zero, and the voltage on the resonant capacitor remains unchanged at Vout-Vin.

[0071] After the main switch is turned off, the circuit enters the energy release stage of the energy storage inductor and the voltage regulation stage of the resonant capacitor. On the one hand, since the current of the energy storage inductor cannot change abruptly, it supplies power to the load resistor and output capacitor at the output terminal through the main diode, completing the basic voltage boost function; on the other hand, the freewheeling current of the energy storage inductor will flow into the resonant capacitor, causing the voltage of the resonant capacitor to gradually rise to VCr=Vout-Vin.

[0072] As energy continues to be released, the current in the energy storage inductor eventually drops to 0. At this point, the current-cutting unit will block the discharge circuit of the resonant capacitor, keeping the voltage of the resonant capacitor stable at Vout-Vin, thus providing a stable initial voltage condition for the subsequent resonance process.

[0073] Step 102: Turn on the main switch transistor. The LC resonant circuit formed by the resonant capacitor and the resonant inductor resonates and discharges. The resonant capacitor is charged to -Vin through resonance, so that the main switch transistor can achieve zero current conduction.

[0074] In this process, at the instant the main switch is turned on, the resonant capacitor and resonant inductor immediately form a complete LC resonant circuit, initiating the resonant discharge process. The electric field energy stored in the resonant capacitor is first converted into the magnetic field energy of the resonant inductor, and then converted into the reverse electric field energy of the resonant capacitor. The entire process takes exactly half a resonant cycle. During this process, the voltage of the resonant capacitor gradually decreases from Vout-Vin to 0, and then reverse-charges to -Vin, where it is clamped by the second diode. At the same time, the current of the main switch follows the resonant law, slowly increasing from 0 without any current surge, thus achieving zero-current conduction and completely eliminating switching losses during the conduction phase.

[0075] Step 103: Keep the main switch on, the current of the energy storage inductor increases linearly, and the voltage on the resonant capacitor remains unchanged at -Vin.

[0076] When the main switch is on, the circuit enters the energy storage inductor and resonant capacitor voltage stabilization phase. The input power supply directly supplies power to the energy storage inductor. Since the inductor current cannot change abruptly, the current of the energy storage inductor increases linearly, continuously storing energy to prepare for the boost output in the next cycle. Meanwhile, the resonant capacitor, clamped by the second diode, maintains a stable voltage at -Vin, preventing fluctuations due to changes in the external circuit and ensuring stable circuit operation.

[0077] Step 104: Turn off the main switch again, and the voltage of the resonant capacitor gradually rises from -Vin to Vout, so that the main switch can achieve zero-voltage turn-off; where Vin is the input voltage and Vout is the output voltage.

[0078] When the main switch is turned off again, the current in the energy storage inductor, unable to change abruptly, immediately freewheels through the resonant circuit. This freewheeling current continuously charges the resonant capacitor, causing its voltage to gradually rise from -Vin to Vout. During this process, the drain-source voltage of the main switch rises slowly and synchronously with the voltage of the resonant capacitor, while the main switch current drops rapidly to 0. Since there is no effective overlap between voltage and current, zero-voltage turn-off is achieved, significantly reducing switching losses during the turn-off phase. At this point, the circuit completes one full PWM cycle and enters the next cycle.

[0079] In one embodiment, by setting the parameters of the resonant capacitor and the resonant inductor to match, and satisfying that when the circuit operates at Vout≥2Vin, after the resonance of the LC resonant circuit ends, the reverse voltage of the resonant capacitor can be clamped to -Vin by the second diode, so as to provide the initial voltage condition for the zero-voltage turn-off of the main switch.

[0080] The core objective of parameter matching is to ensure that half the resonant period of the LC resonant circuit is less than or equal to the on-time of the main switch. Only when this parameter matching relationship is satisfied can the resonant capacitor complete the full polarity reversal from Vout-Vin to the reverse voltage during the on-time of the main switch. Otherwise, the resonant process will be interrupted due to insufficient duration, and the reverse voltage will not reach the clamping threshold.

[0081] According to the law of conservation of energy, the necessary energy condition for a resonant capacitor to complete polarity reversal and reach a reverse voltage of -Vin is Vout ≥ 2Vin. If the boost ratio does not meet this constraint, the initial stored energy of the resonant capacitor is 1 / 2 × Cr × (Vout - Vin). 2 It is insufficient to support reverse charging to -Vin; even if the parameters are matched, the reverse voltage will be lower than the threshold.

[0082] When the LC resonant circuit completes half a cycle of oscillation and the reverse voltage of the resonant capacitor reaches -Vin, the second diode will conduct due to forward bias, precisely stabilizing the reverse voltage of the resonant capacitor at -Vin without fluctuation. This stable voltage is the key initial condition for achieving ZVS when the main switch is turned off. That is, after the main switch is turned off, the freewheeling current of the energy storage inductor will cause the resonant capacitor voltage to gradually rise from -Vin to Vout, and the drain-source voltage of the main switch will slowly rise accordingly, while the current has quickly returned to zero, ultimately achieving zero-loss turn-off with no crossover between voltage and current.

[0083] This embodiment introduces an auxiliary resonant network consisting of a resonant capacitor, a resonant inductor, and multiple diodes into a traditional boost circuit. Utilizing the LC resonance characteristics of the resonant capacitor and inductor, the current of the main switch increases from zero when it is turned on, and the voltage rises slowly while the current quickly returns to zero when it is turned off. This avoids switching losses caused by the overlap of current and voltage, solving the technical problem of large current and voltage overlap losses and low boost efficiency caused by hard switching of the main switch in traditional high-efficiency boost circuits. It achieves zero-voltage turn-off and zero-current turn-on of the main switch, significantly reducing switching losses and improving the circuit's operational stability and energy conversion efficiency under high-frequency conditions.

[0084] Example 3: In one embodiment, refer to Figure 3 and Figure 10 ,and Figure 1a Similar to the scheme shown, when its boost ratio is less than 2, the main switch always has an initial voltage V0 (at time t5) when it is turned off, so it cannot achieve zero-voltage turn-off.

[0085] To address the aforementioned issues, this embodiment proposes a low-power full boost ratio boost circuit. In one embodiment, such as... Figure 21As shown, it includes an energy storage inductor (i.e., L), a main switch (i.e., T), and a main diode (i.e., D), as well as a resonant inductor (i.e., Lr), a resonant capacitor (i.e., Cr), an auxiliary capacitor (i.e., C2), an auxiliary switch (i.e., T1), a second diode (i.e., D2), a third diode (i.e., D3), and a fourth diode (i.e., D4).

[0086] One end of the energy storage inductor is connected to the positive input terminal and the cathode of the second diode, respectively. The other end of the energy storage inductor is connected to the anode of the main diode, the anode of the fourth diode, one end of the resonant capacitor, and one end of the main switch. The cathode of the main diode is connected to the positive output terminal. The anode of the second diode is connected to the other end of the resonant capacitor to form a first connection node (i.e., point A). One end of the auxiliary capacitor is connected to the cathode of the fourth diode to form a second connection node (i.e., point B). The resonant inductor, the third diode, and the auxiliary switch are connected in series between the first connection node and the second connection node. The other end of the main switch and the other end of the auxiliary capacitor are connected to the negative input terminal, respectively.

[0087] The resonant inductor, the third diode, and the auxiliary switch can be arbitrarily connected in series, ensuring that the connection point is between the first and second connection nodes. (Refer to...) Figure 21 In this embodiment, one end of the auxiliary switch is connected to the second connection node, the other end of the auxiliary switch is connected to one end of the resonant inductor, the other end of the resonant inductor is connected to the anode of the third diode, and the cathode of the third diode is connected to the first connection node.

[0088] It is worth noting that the low-power full boost ratio boost circuit proposed in this embodiment differs from the high-efficiency boost circuit proposed in Embodiment 1 in that a fourth diode and an auxiliary capacitor are added. The fourth diode is mainly used to guide the current to the auxiliary capacitor to charge the auxiliary capacitor.

[0089] Reference Figure 11 In one embodiment, the auxiliary switching transistor mentioned in this embodiment corresponds to the current-cutting unit in Embodiment 1, which is located at positions P6, P7, or P8. As can be seen from Embodiment 1, when the current-cutting unit is located at positions P6, P7, or P8, the current-cutting unit is a switching transistor. Other equivalent variations of the low-power full-boost circuit are not described in detail in this embodiment.

[0090] In one embodiment, refer to Figure 21The low-power full boost ratio boost circuit further includes an output capacitor (i.e., C) and a load resistor (i.e., R). One end of the output capacitor is connected to the cathode of the main diode, and the other end of the output capacitor is connected to the negative input terminal. One end of the load resistor is connected to the cathode of the main diode, and the other end of the load resistor is connected to the negative input terminal.

[0091] In one embodiment, the auxiliary switch includes three positions, wherein, in one embodiment, reference is made to... Figure 21 As shown, the drain of the auxiliary switch is connected to one end of the fourth diode and one end of the auxiliary capacitor, respectively, and the source of the auxiliary switch is connected to one end of the resonant inductor. In one embodiment, as... Figure 22 As shown, the drain of the auxiliary switch is connected to one end of the resonant inductor, and the source of the auxiliary switch is connected to the anode of the third diode. In one embodiment, as... Figure 23 As shown, the drain of the auxiliary switch is connected to the cathode of the third diode, and the source of the auxiliary switch is connected to the anode of the second diode and one end of the resonant capacitor.

[0092] In one embodiment, the turn-on timing of the auxiliary switch is consistent with that of the main switch.

[0093] Based on the specific structure of the above circuit, Figure 21 For example, this embodiment first explains the working principle of the low-power full boost ratio boost circuit, whose workflow specifically includes five stages, among which: Phase 1 (t1-t2): The main switch is turned off, the auxiliary switch is turned off, the energy storage inductor releases energy and the resonant capacitor is charged.

[0094] In one embodiment, such as Figure 24 As shown, after both the main switch and the auxiliary switch are turned off, the current in the energy storage inductor cannot change abruptly, so it continues to flow through the main boost circuit. The current flows out of the energy storage inductor, through the main diode to the positive output terminal, supplies power to the load and charges the output capacitor, thus realizing the basic boost function.

[0095] At the same time, the freewheeling current of the energy storage inductor flows into the resonant auxiliary circuit, and the current passes through the fourth diode to charge the auxiliary capacitor, so that the voltage of the auxiliary capacitor is gradually charged to VC2=Vout.

[0096] Simultaneously, the resonant capacitor is charged sequentially through the resonant capacitor and the first switching transistor, gradually charging the voltage of the resonant capacitor until VCr = Vout. Vin.

[0097] Stage 2 (t2 - t3): As the energy continues to be released, the current in the energy storage inductor finally drops to 0; In one embodiment, as Figure 25 shown, at this time the third diode is reverse cut-off, the resonant capacitor has no discharge loop, and the voltage is stably maintained at Vout Vin, to provide an initial condition for the subsequent resonance process.

[0098] Stage 3 (t3 - t4): The main switch is turned on, and the auxiliary switch is turned on synchronously.

[0099] In one embodiment, as Figure 26 shown, the main switch and the auxiliary switch are turned on synchronously. The drain potential of the main switch is pulled to the negative input potential (approximately 0V). The conduction of the auxiliary switch打通 the LC resonance loop of the resonant capacitor - third diode - resonant inductor - auxiliary capacitor.

[0100] The initial energy storage of the resonant capacitor (Vout Vin) and the energy storage of the auxiliary capacitor participate in resonance together: in the first 1 / 4 cycle, the electric field energy of the resonant capacitor is converted into the magnetic field energy of the resonant inductor, and the capacitor voltage drops to 0; in the next 1 / 4 cycle, the magnetic field energy of the resonant inductor is reversely converted into the electric field energy of the resonant capacitor, causing the capacitor voltage polarity to reverse.

[0101] In one embodiment, under the condition of a low boost ratio (1 < n < 2), the energy storage of the auxiliary capacitor can supplement the energy gap of the resonant capacitor to ensure that its reverse voltage stably rises to Vin; under the condition of a high boost ratio (n ≥ 2), the resonant capacitor has sufficient self - energy storage, and the auxiliary capacitor does not affect the resonance process.

[0102] During this process, the current of the main switch follows the resonance law and slowly increases from 0, without current impact, achieving zero - current turn - on.

[0103] Stage 4 (t4 - t5): The main switch remains on, and the auxiliary switch remains on.

[0104] In one embodiment, as Figure 27 shown, the energy storage of the energy storage inductor and the resonance parameters are stable. The main switch and the auxiliary switch continue to be on. The input power supply directly supplies power to the energy storage inductor, and the current of the energy storage inductor shows a linear upward trend, continuously storing energy.

[0105] The reverse voltage of the resonant capacitor is clamped at Vin by the second diode to prevent the voltage from rising excessively and causing device breakdown; the resonant inductor and the auxiliary capacitor are both in an energy - stable state, and the entire resonant auxiliary network has no energy fluctuation, laying a foundation for the realization of ZVS in the subsequent turn - off stage.

[0106] Stage 5 (t5 - t6): The main switch is turned off, and the auxiliary switch is turned off synchronously.

[0107] In one embodiment, as Figure 28 shown, the main switch and the auxiliary switch are turned off synchronously. Due to the non - mutation of the current in the energy - storage inductor, the energy is immediately released through the resonant free - wheeling circuit to charge the resonant capacitor positively. The voltage of the resonant capacitor rises gradually from Vin to Vout, and the drain - source voltage of the main switch rises slowly synchronously with the voltage of the resonant capacitor. At the same time, the current of the main switch has quickly dropped to 0, and there is no cross - overlap region between the voltage and the current, achieving zero - voltage turn - off.

[0108] When the voltage of the resonant capacitor rises to Vout, the main diode conducts naturally, and the remaining energy of the energy - storage inductor continues to be released to the output terminal, completing the energy conversion of one PWM cycle.

[0109] The core innovation of this circuit lies in the energy - supplement function of the auxiliary capacitor and the timing control of the auxiliary switch: that is, when the boost ratio is high (n≥2), the initial energy storage of the resonant capacitor is sufficient, and the polarity inversion can be completed without the participation of the auxiliary capacitor, and the circuit operating characteristics are the same as those of the traditional resonant boost; when the boost ratio is low (1 < n < 2), the energy storage of the auxiliary capacitor can compensate for the energy gap of the resonant capacitor to ensure the integrity of the resonance process, breaking through the boost - ratio limit of the traditional scheme. Finally, through the precise cooperation of the resonant auxiliary network and the main circuit, low - power soft - switching operation within the full boost - ratio range is achieved.

[0110] To ensure that the LC resonant circuit can complete the voltage - polarity inversion of the resonant capacitor under the condition of low boost ratio (1 < n < 2), that is, the output voltage is greater than the input voltage and less than twice the input voltage, and to ensure that the zero - current - turn - on (ZCS) condition of the main switch is stably established, the capacitance value of the auxiliary capacitor needs to be accurately calculated through the principle of energy conservation.

[0111] In one embodiment, the optimal capacitance value of the auxiliary capacitor is when, in stage three, the auxiliary capacitor, the resonant capacitor, and the resonant inductor resonate, causing the voltage on the resonant capacitor to flip, and the flipped voltage Vcr is exactly equal to Vin, while the voltage on the auxiliary capacitor is equal to zero and the current on the second diode is exactly 0, that is, at this time, the second diode does not need to be clamped, and the power of the second diode is the minimum loss power.

[0112] In one embodiment, as Figure 29 、 Figure 30 and Figure 31 shown, are the initial states of the auxiliary capacitor, the resonant capacitor, and the resonant inductor, that is, the state at time t3, and the end - state of resonance, that is, the state at time t4.

[0113] The resonance process lasts for half a cycle. According to the law of conservation of energy, the energy of the auxiliary capacitor and the resonant capacitor at time t3 is equal to the energy of the resonant capacitor at time t4. Since the voltage across the auxiliary capacitor is 0 at time t4, all the energy is in the resonant capacitor.

[0114] The specific calculation method for the capacitance value C2 of the auxiliary capacitor is as follows: Ecr|t3+Ecr|t4=Ecr|t4|; Cr×Vcr 2 / 2=C2×V2 2 / 2+Cr×V1 2 / 2; Cr×Vcr 2 =C2×V2 2 +Cr×V1 2 .......(1) Initial conditions: V1 = Vout - Vin, V2 = Vout; Substituting into equation (1), we get: at the end of time t4, Vcr = -Vin; Cr×Vin 2 =C2×Vout 2 +Cr×(Vout-Vin) 2 .......(2) Substituting the capacitance ratio C2 / Cr=K into equation (2), we get: Vin 2 =K×Vout 2 +(Vout-Vin) 2 ; K=(Vin 2 -Vout 2 +2×Vout×Vin-Vin 2 ) / Vout 2 .......(3) Substituting the boost ratio n = Vout / Vin into equation (3) yields: K=C2 / Cr=2 / n-1........(4) Where C2 is the capacitance of the auxiliary capacitor, Cr is the capacitance of the resonant capacitor, Vin is the input voltage, and Vout is the output voltage. When n>=2, K<=0, so a solution only exists for n<2.

[0115] When n>=2 and C2=0, Cr is always Cr>=Vin. Therefore, this circuit does not need an auxiliary capacitor (and correspondingly, it does not need a fourth diode) when n>=2, and can guarantee Vcr>=Vin at time t4. Therefore, the circuit can be evolved as follows: Figure 3 The circuit shown in the diagram of the first scheme has lower losses. The specific working principle of this circuit is explained in subsequent embodiments.

[0116] In one embodiment, when 1 < n < 2, the optimal capacitance values of Cr and C2 can be obtained through formula (4). At this time, the current on D1 is equal to 0, and the circuit loss is minimized.

[0117] When n >= 2, Figure 21 Due to the clamping of the second diode, the circuit shown can still ensure that Vcr = Vin. The excess energy returns to the input voltage source through the second diode. The currents on the second diode and the third diode increase, and the power consumption of the second diode and the third diode is relatively large.

[0118] In order to balance the energy requirements of a low boost ratio, the stability of the full boost ratio operating conditions, and the reliability of the soft-switching conditions, in one embodiment, the capacitance value of the auxiliary capacitor is greater than or equal to 3 times the capacitance value of the resonant capacitor.

[0119] Without the auxiliary capacitor, the initial energy storage of the resonant capacitor in the low boost ratio operating condition (1 < n < 2) is 1 / 2 × Cr × (Vout - Vin) 2 , which is not sufficient to support its polarity reversal from VCr = Vout - Vin to VCr = -Vin. Therefore, it is necessary to rely on the auxiliary capacitor to supplement energy.

[0120] In one embodiment, according to the law of conservation of energy, the energy provided by the auxiliary capacitor needs to cover the energy storage difference of the resonant capacitor. After derivation, when the boost ratio (n = 1.5) (extremely low boost ratio), theoretically C2 ≥ 0.75Cr can meet the energy requirements. However, in actual engineering, sufficient redundancy needs to be reserved to offset the influences of device parameter deviations, line losses, electromagnetic interference, etc. Therefore, the conservative constraint of C2 ≥ 3Cr is taken to ensure that even in the most severe low boost ratio operating conditions, the energy storage of the auxiliary capacitor can fully compensate for the energy gap of the resonant capacitor, guaranteeing the integrity of the resonance process. Ensure that the resonant capacitor reaches -Vin within half a resonant period and is locked by the second diode, providing a reliable basis for the voltage slow rise when the main switch tube is turned off, and ultimately guaranteeing the stable realization of ZVS and avoiding the increase in switching losses caused by inaccurate voltage clamping.

[0121] In one embodiment, in the LC resonance circuit composed of the resonant inductor, the resonant capacitor, and the auxiliary capacitor, its half resonant period is less than the on-time of the main switch tube.

[0122] Among them, in the LC resonance circuit, the voltage polarity reversal of the resonant capacitor must rely on a complete half resonant period (the complete conversion of electric field energy → magnetic field energy → reverse electric field energy), which is the essential law of LC resonance and also the prerequisite for the main switch tube to achieve zero-current conduction (ZCS).

[0123] In one embodiment, the core objective of turning on the main switch is to allow the resonant capacitor to complete polarity reversal from the initial voltage Vout-Vin (Vout is the output voltage, Vin is the input voltage) and be clamped to -Vin by the second diode. If half a resonant period is greater than or equal to the main switch's on-time, the resonant process will not be completed (the capacitor voltage will not reverse or will not reverse sufficiently), the main switch current will not be able to start from 0, ZCS will fail, and the subsequent zero-voltage turn-off (ZVS) condition will be violated.

[0124] In one embodiment, the current of the energy storage inductor drops to 0 in each PWM duty cycle, so that the low-power full boost ratio boost circuit operates in discontinuous current mode.

[0125] The criterion for determining the discontinuous current mode is that the current of the energy storage inductor can drop to 0 in each PWM working cycle. This is different from the continuous current mode, where the current of the energy storage inductor does not drop to 0 and there is always freewheeling current, which cannot provide a stable initial operating condition for soft switching.

[0126] In summary, this embodiment controls the synchronous switching of the main switch and the auxiliary switch, combined with the half-cycle free oscillation of the resonant capacitor, auxiliary capacitor and resonant inductor, to force the resonant capacitor voltage to reverse and stabilize at the input voltage Vin. This ensures that the drain-source voltage of the main switch gradually rises from zero to the output voltage Vout at the moment of turn-off, completely eliminating turn-off voltage surges and switching losses. Moreover, this effect is not affected by the boost ratio, solving the technical pain point of traditional boost circuits that are difficult to achieve ZVS at low boost ratios.

[0127] Example 4: This embodiment proposes a control method for a low-power full boost ratio boost circuit. In one embodiment, such as... Figure 32 As shown, the control method includes: Step 101: Control the main switch and the auxiliary switch to turn off, and the energy storage inductor charges the resonant capacitor and the auxiliary capacitor so that the voltage of the resonant capacitor reaches Vout-Vin and the voltage of the auxiliary capacitor reaches Vout.

[0128] When both the main and auxiliary switches are turned off, the circuit enters the energy release and capacitor charging phase of the energy storage inductor. Since the current in the energy storage inductor cannot change abruptly, it releases the stored energy to the resonant capacitor and the auxiliary capacitor. On one hand, current flows into the resonant capacitor, gradually increasing its voltage to Vout-Vin; on the other hand, current simultaneously charges the auxiliary capacitor, bringing its voltage to Vout.

[0129] Step 102: Keep the main switch and the auxiliary switch off, the current of the energy storage inductor drops to zero, the voltage on the resonant capacitor remains Vout-Vin, and the voltage on the auxiliary capacitor remains Vout.

[0130] During this process, the main and auxiliary switching transistors remain off, and the energy storage inductor continuously releases energy. Its current gradually decreases as energy is consumed, eventually dropping to zero. Throughout this process, the circuit has no additional energy input or discharge path. The resonant capacitor, relying on its non-discharge characteristic, maintains a stable voltage at Vout-Vin, and the voltage of the auxiliary capacitor also remains constant at Vout.

[0131] Step 103: Control the main switch and the auxiliary switch to be turned on, so that the LC resonant circuit including the resonant capacitor, the auxiliary capacitor and the resonant inductor oscillates freely for half a cycle, so that the voltage on the resonant capacitor flips in the reverse direction from Vout-Vin to Vin, and the voltage of the auxiliary capacitor drops to zero, so that the main switch can achieve zero current conduction.

[0132] In this process, the main switch and auxiliary switch are synchronously turned on, and the resonant capacitor, auxiliary capacitor, and resonant inductor immediately form a complete LC resonant circuit, initiating a free oscillation process for half a cycle. During the first quarter of the oscillation cycle, the electric field energy of the resonant capacitor and auxiliary capacitor are converted into the magnetic field energy of the resonant inductor, causing the voltage of the resonant capacitor to drop from Vout-Vin to 0, and the voltage of the auxiliary capacitor to drop to 0 simultaneously. During the second quarter of the oscillation cycle, the magnetic field energy of the resonant inductor is converted back into the electric field energy of the resonant capacitor, causing the voltage of the resonant capacitor to reverse to Vin. Throughout this process, the current of the main switch follows the resonant law, slowly increasing from 0 without any current surge, thus achieving zero-current conduction.

[0133] Step 104: Keep the main switch and the auxiliary switch on, so that the energy storage inductor absorbs energy from the input voltage and the current rises linearly.

[0134] In this circuit, the main and auxiliary switching transistors remain on, creating a path between the input power supply and the energy storage inductor. The inductor then begins to absorb energy from the input voltage. Since the inductor current cannot change abruptly, it increases linearly, continuously storing energy to prepare for the boost output and capacitor charging in the next PWM cycle. At this point, the voltage of the resonant capacitor is clamped to Vin, and the resonant circuit is in a stable energy state, preventing interference with the energy storage process of the inductor.

[0135] Step 105: Control the shutdown of the main switch and the auxiliary switch. The voltage across the resonant capacitor is Vin. The drain-source voltage of the main switch rises from zero, thereby achieving zero-voltage shutdown. Wherein, Vin is the input voltage and Vout is the output voltage.

[0136] In this circuit, the main and auxiliary switches are turned off synchronously. At this point, the voltage across the resonant capacitor stabilizes at Vin, and the drain-source voltage of the main switch slowly rises from zero as the resonant capacitor voltage changes. Simultaneously, the current in the energy storage inductor, unable to change abruptly, freewheels through the resonant circuit, while the current in the main switch rapidly drops to zero. With no overlap between voltage and current, turn-off losses are completely eliminated, ultimately achieving zero-voltage turn-off. At this point, the circuit completes one full PWM cycle and enters the next cycle.

[0137] In summary, this embodiment controls the synchronous switching of the main switch and the auxiliary switch, combined with the half-cycle free oscillation of the resonant capacitor, auxiliary capacitor and resonant inductor, to force the resonant capacitor voltage to reverse and stabilize at the input voltage Vin. This ensures that the drain-source voltage of the main switch gradually rises from zero to the output voltage Vout at the moment of turn-off, completely eliminating turn-off voltage surges and switching losses. Moreover, this effect is not affected by the boost ratio, solving the technical pain point of traditional boost circuits that are difficult to achieve ZVS at low boost ratios.

[0138] For the specific structure of the low-power full boost ratio boost circuit, please refer to Embodiment 3, which will not be repeated here.

[0139] The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. A low-power full boost ratio boost circuit, comprising an energy storage inductor, a main switching transistor, and a main diode, characterized in that, It also includes a resonant inductor, a resonant capacitor, an auxiliary capacitor, an auxiliary switching transistor, a second diode, a third diode, and a fourth diode; One end of the energy storage inductor is connected to the positive input terminal and the cathode of the second diode, respectively, and the other end of the energy storage inductor is connected to the anode of the main diode, the anode of the fourth diode, one end of the resonant capacitor, and one end of the main switch. The cathode of the main diode is connected to the positive output terminal; the anode of the second diode is connected to the other end of the resonant capacitor to form a first connection node; one end of the auxiliary capacitor is connected to the cathode of the fourth diode to form a second connection node; the resonant inductor, the third diode, and the auxiliary switch are connected in series between the first connection node and the second connection node; the other end of the main switch and the other end of the auxiliary capacitor are respectively connected to the negative input terminal.

2. The low-power full boost ratio boost circuit according to claim 1, characterized in that, The drain of the auxiliary switch is connected to one end of the fourth diode and one end of the auxiliary capacitor, respectively, and the source of the auxiliary switch is connected to one end of the resonant inductor.

3. The low-power full boost ratio boost circuit according to claim 1, characterized in that, The drain of the auxiliary switch is connected to one end of the resonant inductor, and the source of the auxiliary switch is connected to the anode of the third diode.

4. The low-power full boost ratio boost circuit according to claim 1, characterized in that, The drain of the auxiliary switch is connected to the cathode of the third diode, and the source of the auxiliary switch is connected to the anode of the second diode and one end of the resonant capacitor.

5. The low-power full boost ratio boost circuit according to any one of claims 2 to 4, characterized in that, The conduction timing of the auxiliary switch is consistent with that of the main switch.

6. The low-power full boost ratio boost circuit according to claim 1, characterized in that, The capacitance of the auxiliary capacitor is greater than or equal to three times the capacitance of the resonant capacitor.

7. The low-power full boost ratio boost circuit according to claim 1, characterized in that, In the LC resonant circuit formed by the resonant inductor, the resonant capacitor, and the auxiliary capacitor, half of its resonant period is less than the conduction time of the main switch.

8. The low-power full boost ratio boost circuit according to claim 1, characterized in that, The output voltage is greater than the input voltage but less than twice the input voltage.

9. The low-power full boost ratio boost circuit according to claim 1, characterized in that, The current of the energy storage inductor drops to 0 in each PWM operating cycle, so that the low-power full boost ratio boost circuit operates in discontinuous current mode.

10. A control method for a low-power full boost ratio boost circuit, characterized in that, The control method for controlling the low-power full boost ratio boost circuit as described in any one of claims 1-9 includes: The main switch and the auxiliary switch are turned off by control, and the energy storage inductor charges the resonant capacitor and the auxiliary capacitor, so that the voltage of the resonant capacitor reaches Vout-Vin and the voltage of the auxiliary capacitor reaches Vout. The main switch and the auxiliary switch remain off, the current in the energy storage inductor drops to zero, the voltage on the resonant capacitor remains Vout-Vin, and the voltage on the auxiliary capacitor remains Vout. The main switch and the auxiliary switch are controlled to be turned on, so that the LC resonant circuit including the resonant capacitor, the auxiliary capacitor and the resonant inductor oscillates freely for half a cycle, so that the voltage on the resonant capacitor flips in the reverse direction from Vout-Vin to Vin, and the voltage on the auxiliary capacitor drops to zero, so that the main switch can achieve zero current conduction. Keep the main switch and the auxiliary switch on, so that the energy storage inductor absorbs energy from the input voltage and the current rises linearly; The main switch and the auxiliary switch are turned off by controlling the main switch and the auxiliary switch. The voltage across the resonant capacitor is Vin, and the drain-source voltage of the main switch rises from zero, thereby achieving zero-voltage turn-off. Here, Vin is the input voltage and Vout is the output voltage.

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