Permanent magnet synchronous motor phase current limiting method, system, device and medium

Abstract

The application relates to the technical field of motor control, and provides a phase current limiting method, system, device and medium of a permanent magnet synchronous motor. The implementation scheme is as follows: operation parameters and a current commutation state of the permanent magnet synchronous motor in a current control period are acquired; each current prediction model is constructed based on a three-phase voltage equation and a three-phase flux linkage equation of the permanent magnet synchronous motor, and a target current prediction model is determined from each current prediction model based on the current commutation state; the operation parameters and the current commutation state are input into the target current prediction model to predict a predicted current in a next control period; if the predicted current is greater than a preset current limiting threshold, the target current prediction model is inversely solved based on the current limiting threshold and the operation parameters to obtain a target duty cycle; and the inverter bridge is controlled to drive the permanent magnet synchronous motor to operate according to a pulse width modulation signal generated by the target duty cycle and the current commutation state, so as to limit the phase current of the permanent magnet synchronous motor. The embodiment of the application can realize smooth control of the phase current.

Description

Technical Field

[0001] This invention relates to the field of motor control technology, and in particular to a method, system, device and medium for limiting phase current of a permanent magnet synchronous motor. Background Technology

[0002] Permanent magnet synchronous motors (PMSMs) are widely used in new energy vehicles, power tools, and industrial drives due to their high power density, high control precision, and high operating efficiency. In the control process of PMSMs, phase current is a crucial parameter affecting the motor's torque output and operational stability. To prevent the phase current from exceeding the tolerance range of the devices or motor windings, a phase current limiting mechanism is typically implemented in the control system.

[0003] In existing technologies, overcurrent protection for phase current limiting in permanent magnet synchronous motors often employs methods based on current sampling and hardware comparison circuits. This involves controlling the inverter bridge to shut down or adjusting the drive signal after detecting a phase current reaching a preset threshold. However, this approach suffers from hysteresis, requiring the current-limiting action to occur only after the current exceeds the threshold, leading to abrupt changes in the drive waveform between adjacent control cycles. Furthermore, the involvement of hardware comparison circuits alters the drive timing within some control cycles, potentially causing discontinuities in the pulse width modulation waveform and consequently resulting in unsmooth phase current.

[0004] Therefore, how to achieve smooth phase current control in the six-step commutation control process of a permanent magnet synchronous motor is a technical problem that urgently needs to be solved by those skilled in the art. Summary of the Invention

[0005] This invention provides a method, system, device, and medium for limiting phase current of a permanent magnet synchronous motor, which can solve the technical problem of unsmooth phase current limiting process in existing permanent magnet synchronous motors.

[0006] In a first aspect, embodiments of the present invention provide a method for limiting the phase current of a permanent magnet synchronous motor, comprising: Obtain the operating parameters and current commutation status of the permanent magnet synchronous motor within the current control cycle; Based on the three-phase voltage equation and three-phase flux linkage equation of the permanent magnet synchronous motor, a current prediction model corresponding to each commutation state is constructed, and based on the current commutation state, a target current prediction model is determined from each of the current prediction models. The operating parameters and the current commutation state are input into the target current prediction model to predict the predicted current for the next control cycle. If the predicted current is greater than the preset current limiting threshold, the target current prediction model is solved in reverse based on the current limiting threshold and the operating parameters to obtain the target duty cycle; The control inverter bridge generates a pulse width modulation signal based on the target duty cycle and the current commutation state to drive the permanent magnet synchronous motor to operate, thereby limiting the phase current of the permanent magnet synchronous motor.

[0007] Secondly, embodiments of the present invention provide a phase current limiting system for a permanent magnet synchronous motor, comprising: The acquisition module is used to acquire the operating parameters and current commutation status of the permanent magnet synchronous motor in the current control cycle; The current prediction model construction module is used to construct current prediction models corresponding to each commutation state based on the three-phase voltage equation and the three-phase flux linkage equation of the permanent magnet synchronous motor, and to determine the target current prediction model from each of the current prediction models based on the current commutation state. The prediction module is used to input the operating parameters and the current commutation state into the target current prediction model to predict the predicted current for the next control cycle. The reverse solution module is used to reverse solve the target current prediction model based on the current limiting threshold and the operating parameters if the predicted current is greater than the preset current limiting threshold, so as to obtain the target duty cycle. The phase current limiting module is used to control the inverter bridge to drive the permanent magnet synchronous motor to operate based on the target duty cycle and the current commutation state, so as to limit the phase current of the permanent magnet synchronous motor.

[0008] Thirdly, embodiments of the present invention also provide a computer device, including: at least one processor; and a memory communicatively connected to the at least one processor; wherein the memory stores instructions executable by the at least one processor, the instructions being executed by the at least one processor to enable the at least one processor to perform the method described in any one of the embodiments of the present invention.

[0009] Fourthly, embodiments of the present invention also provide a non-transitory computer-readable storage medium storing computer instructions, wherein the computer instructions are used to cause a computer to perform the method described in any one of the embodiments of the present invention.

[0010] The present invention first acquires the operating parameters and current commutation state of the permanent magnet synchronous motor (PMSM) within the current control cycle. Based on the three-phase voltage equation and three-phase flux linkage equation of the PMSM, a current prediction model corresponding to each commutation state is constructed. This allows for the determination of the target current prediction model corresponding to the current conducting loop, based on the current commutation state. Further, the operating parameters are input into the target current prediction model to predict the phase current for the next control cycle, thus obtaining the current change trend before the phase current reaches the current limiting threshold. Subsequently, when the predicted current exceeds the preset current limiting threshold, the target current prediction model is solved in reverse based on the current limiting threshold and the operating parameters to obtain the target duty cycle corresponding to the target current. This transforms the pulse width modulation signal adjustment process from instantaneous shutdown to duty cycle adjustment. Finally, the control inverter bridge generates a pulse width modulation signal based on the target duty cycle and the current commutation state to drive the PMSM. This ensures continuous change of the drive signal between adjacent control cycles, thereby gradually adjusting the phase current according to the preset trend, avoiding sudden changes in the phase current, and achieving smooth phase current control.

[0011] It should be understood that the description in this section is not intended to identify key or essential features of the embodiments of the present invention, nor is it intended to limit the scope of the invention. Other features of the invention will become readily apparent from the following description. Attached Figure Description

[0012] The accompanying drawings are provided for a better understanding of this solution and do not constitute a limitation of the invention. Wherein: Figure 1 This is a flowchart of a phase current limiting method for a permanent magnet synchronous motor according to an embodiment of the present invention; Figure 2 This is a structural block diagram of a phase current limiting system for a permanent magnet synchronous motor according to an embodiment of the present invention; Figure 3 This is a schematic block diagram of a computer device used to implement the methods of the embodiments of the present invention. Detailed Implementation

[0013] The following description, in conjunction with the accompanying drawings, illustrates exemplary embodiments of the present invention, including various details to aid understanding. These details should be considered merely exemplary. Therefore, those skilled in the art will recognize that various changes and modifications can be made to the embodiments described herein without departing from the scope of the invention. Similarly, for clarity and brevity, descriptions of well-known functions and structures are omitted in the following description.

[0014] This invention provides a method, system, device, and medium for limiting the phase current of a permanent magnet synchronous motor. The execution entity of this method can be the phase current limiting system for a permanent magnet synchronous motor provided in this invention, or a computer device integrating the phase current limiting system. The phase current limiting system can be implemented in hardware or software, and the computer device can be a terminal or a server.

[0015] In a specific application scenario, the method of this invention is applied to a permanent magnet synchronous motor drive system employing six-step commutation control. This drive system includes a main control unit, a position detection unit, a current sampling unit, a voltage sampling unit, and a three-phase inverter bridge. The main control unit drives the three-phase inverter bridge according to the six-step commutation control method, and in each control cycle, it acquires the bus voltage, bus current, rotor electrical angle, and rotor electrical angular velocity, while simultaneously obtaining the current commutation state. The main control unit selects the corresponding target current prediction model based on the current commutation state and inputs the acquired operating parameters into the target current prediction model to obtain the predicted current for the next control cycle. When the predicted current exceeds a preset current limiting threshold, the main control unit reverse-engineers the current prediction model based on the current limiting threshold and operating parameters to obtain the target duty cycle, and controls the three-phase inverter bridge to output the corresponding pulse width modulation signal according to the target duty cycle. Therefore, under conditions such as start-up, commutation, or sudden load changes in the electric actuator, the phase current of the permanent magnet synchronous motor can be adjusted according to the prediction results.

[0016] Figure 1 This is a flowchart of a phase current limiting method for a permanent magnet synchronous motor according to an embodiment of the present invention.

[0017] like Figure 1 As shown, the phase current limiting method for this permanent magnet synchronous motor may include: S110: Obtain the operating parameters and current commutation status of the permanent magnet synchronous motor in the current control cycle.

[0018] For example, operating parameters refer to parameters used to characterize the electrical state of the motor and the motion state of the rotor within the current control cycle, including bus voltage, bus current, rotor electrical angle, and rotor electrical angular velocity. Specifically, the bus voltage is obtained by a bus voltage sampling circuit through voltage division sampling and analog-to-digital conversion; the bus current is obtained by a bus current sampling circuit through a sampling resistor, Hall current sensor, or current transformer; the rotor electrical angle is calculated from the rotor position signal output by a Hall sensor, encoder, or resolver; and the rotor electrical angular velocity is calculated from the difference in rotor electrical angle between adjacent control cycles and the control cycle duration.

[0019] In this example, the main control unit triggers the analog-to-digital converter at a preset sampling time during the PWM (Pulse Width Modulation) counting cycle to read the bus voltage and bus current sampling values, and converts them into bus voltage and bus current according to the scaling factor of the sampling circuit. The scaling factor of the sampling circuit is a pre-calibrated parameter (e.g., the bus voltage sampling ratio is 1 / 20 to 1 / 200, preferably 1 / 100 in this embodiment; the bus current sampling ratio is determined by the sampling resistor value and the amplification factor, with the sampling resistor value ranging from 0.5mΩ (milliohm) to 20mΩ, preferably 5mΩ in this embodiment). The main control unit reads the mechanical angle output by the rotor position detection unit and calculates the rotor electrical angle by combining it with the number of motor pole pairs. ;in, For rotor electrical angle, The number of pole pairs of the motor (e.g., 1 to 10, with 4 in a preferred embodiment). This refers to the rotor mechanical angle.

[0020] The rotor's electric angular velocity is calculated using the following formula: ; in, The rotor electrical angular velocity for the current control cycle. The rotor electrical angle for the current control cycle. The rotor electrical angle of the previous control cycle. To control the cycle duration (e.g., from 20 μs to 200 μs, with a specific example of 100 μs).

[0021] For example, the current commutation state refers to the combination of phase bridge arms currently in the conducting state in the six-step commutation control. The main control unit obtains the current commutation state by looking up a table based on the electrical angle range of the rotor electrical angle or the status code of the Hall sensor. For example, the current commutation state includes phase A connected to the positive terminal of the bus and phase B connected to the negative terminal of the bus, phase A connected to the positive terminal of the bus and phase C connected to the negative terminal of the bus, phase B connected to the positive terminal of the bus and phase C connected to the negative terminal of the bus, phase B connected to the positive terminal of the bus and phase A connected to the negative terminal of the bus, phase C connected to the positive terminal of the bus and phase A connected to the negative terminal of the bus, and phase C connected to the positive terminal of the bus and phase B connected to the negative terminal of the bus. In this way, the subsequent current prediction can correspond to the currently conducting phase loop, avoiding the mixing of current paths in different commutation states.

[0022] S120: Based on the three-phase voltage equation and three-phase flux linkage equation of the permanent magnet synchronous motor, a current prediction model corresponding to each commutation state is constructed, and based on the current commutation state, a target current prediction model is determined from each current prediction model.

[0023] For example, a current prediction model refers to a discrete model used to describe the correspondence between bus voltage, bus current, back EMF, resistor voltage drop, inductor voltage and current in the next control cycle under a certain commutation state.

[0024] In this example, the three-phase voltage equations are first established: ; in, , , These are the stator voltages for phases A, B, and C, respectively. , , These are the stator currents for phases A, B, and C, respectively. The stator resistance is from 0.01Ω (ohm) to 20Ω, with a specific example being 0.2Ω. , , These are the magnetic flux linkages for phases A, B, and C, respectively.

[0025] The three-phase flux linkage relationship is expressed as follows: ; in, The stator inductance is from 0.01 mH (millihenry) to 10 mH, with a specific embodiment being 0.5 mH. The flux linkage is for permanent magnets (e.g., 0.001 Wb (Weber) to 0.2 Wb, specifically 0.01 Wb in this embodiment). The above flux linkage relationship is modeled primarily based on the stator winding self-inductance. Mutual inductance of the three-phase windings is not included in the calculation if it meets a preset neglect condition relative to self-inductance. The preset neglect condition is that the ratio of mutual inductance to self-inductance is less than a set ratio threshold (e.g., 0.01 to 0.2, specifically 0.05 in this embodiment).

[0026] Substituting the three-phase flux linkage relationship into the three-phase voltage equation, we obtain the dynamic relationship of the three-phase current: ; Subsequently, the main control unit constructs a current prediction model corresponding to each commutation state based on the conducting phase combinations under the six-step commutation control. For example, when the current commutation state is that phase A is connected to the positive terminal of the bus and phase B is connected to the negative terminal of the bus, phase A and phase B form a conducting phase loop, and phase C is a floating phase, satisfying: ; in, Bus voltage Let be the bus current. Substituting the above relationship into the three-phase current dynamic relationship, we obtain the continuous current model of the AB conducting phase loop: ; The continuous current model is then discretized to obtain the current prediction model corresponding to the AB commutation state: ; in, The predicted current for the next control cycle, This is the back electromotive force characteristic of the current commutation state. For the AB conducting phase circuit, for: ; For other commutation states, the corresponding current prediction model is generated in the same way from the phase voltage difference, phase current direction, and back EMF phase of the corresponding conducting phase. In this way, each current prediction model is consistent with the current direction and back EMF phase of the actual conducting phase circuit, which can provide a clear physical calculation chain for subsequent predictions.

[0027] For example, the current prediction models for the other five switching states are shown in Table 1 below: S130 inputs the operating parameters and the current commutation state into the target current prediction model to predict the predicted current for the next control cycle.

[0028] In this example, the main control unit first determines the back EMF phase identifier in the target current prediction model based on the current commutation state. For example, the back EMF phase identifier for phase A is 0, and the back EMF phase identifier for phase B is... The back electromotive force phase of phase C is identified as follows: .

[0029] Taking phase A connected to the positive terminal of the busbar and phase B connected to the negative terminal of the busbar as an example, the main control unit first calculates the back EMF characterization quantity: .

[0030] Then calculate the effect of the back electromotive force: ;in, This represents the back EMF influence quantity corresponding to the AB conducting phase circuit.

[0031] Next, calculate the voltage drop across the resistance corresponding to the conducting phase loop: ;in, This is the voltage drop across the resistor. Since two phases are conducting in the six-step commutation state, and the conducting phase circuit includes the positive conducting phase winding and the negative conducting phase winding, the voltage drop across the resistor is calculated based on the resistance of the two phase windings.

[0032] Then the effective voltage is calculated: ;in, It is the effective voltage applied to the equivalent inductance of the conducting phase circuit.

[0033] Calculate the rate of change of current based on the relationship between inductor voltage and rate of change of current: .

[0034] Then calculate the current change based on the control cycle duration: .

[0035] Finally, the predicted current for the next control cycle is obtained: .

[0036] For example, in the AB conducting phase circuit, the main control unit substitutes the currently collected bus voltage, bus current, rotor electrical angle, and rotor electrical angular velocity into the above calculation link to obtain the back EMF characterization quantity, back EMF influence quantity, resistance voltage drop quantity, effective voltage quantity, current change rate, and predicted current in sequence. Thus, the predicted current is not simply extrapolated from the currently sampled current, but simultaneously considers the bus voltage, resistance voltage drop, inductance energy storage change, and back EMF influence, and the prediction process corresponds to the current electrical state of the permanent magnet synchronous motor.

[0037] S140, if the predicted current is greater than the preset current limiting threshold, then the target current prediction model is solved in reverse based on the current limiting threshold and the operating parameters to obtain the target duty cycle.

[0038] For example, the current limiting threshold refers to the current constraint value that the conducting phase loop current is allowed to reach. The current limiting threshold is determined by the rated phase current of the motor, the rated current of the inverter bridge switching devices, winding thermal limiting, or system protection strategy (e.g., 0.5 to 1.5 times the rated phase current of the motor, specifically 1.0 times the rated phase current in a specific embodiment).

[0039] For example, the target duty cycle refers to the duty cycle used to generate the pulse width modulation signal within the current control cycle. The target duty cycle ranges from 0 to 1, where 0 indicates that no effective modulation voltage is output within the current PWM cycle, and 1 indicates that the maximum equivalent bus voltage is output according to the current commutation state within the current PWM cycle.

[0040] In this example, when the predicted current satisfies: The main control unit sets the target current to the current limiting threshold. The bus voltage in the current prediction model is replaced with the equivalent bus voltage modulated by the duty cycle: ;in, The rate limiting threshold, Bus voltage This is the equivalent bus voltage. This is the duty cycle for limiting the flow.

[0041] Taking the AB conducting phase circuit as an example, The current constraint value for the next control cycle is written into the current prediction model: ; right By performing a reverse solution, we obtain: ; in, This is the current-limiting duty cycle. If the calculated... If it is less than 0, then The amplitude is limited to 0; if the calculated amplitude is 0... If it is greater than 1, then The amplitude limit is set to 1. The amplitude limit processing is as follows: ; in, This is the duty cycle after the bandwidth is limited.

[0042] When the motor also has speed closed-loop control, the main control unit also calculates the speed regulation duty cycle. Specifically, it obtains the target speed. and actual speed Calculate the speed deviation: ; in, The target rotational speed is (e.g., 100 r / min (revolutions per minute) to 20,000 r / min, specifically 3,000 r / min in this embodiment). This refers to the actual rotational speed. This refers to the speed deviation.

[0043] Input the speed deviation into the proportional-integral controller to obtain the speed loop output: ; in, To adjust the duty cycle for speed, This is a scaling factor (e.g., 0.001 to 1, 0.02 in a specific embodiment). This is the integration coefficient (e.g., 0.001 to 10, 0.1 in a specific embodiment). For Perform amplitude limiting processing within the range of 0 to 1.

[0044] The target duty cycle is determined by the following formula: ; in, The target duty cycle is set as follows: when the duty cycle required by the speed loop causes the current in the next control cycle to exceed the current limiting threshold, the target duty cycle is constrained by the current limiting duty cycle; when the duty cycle required by the speed loop does not trigger the current limiting, the target duty cycle follows the speed regulation requirements.

[0045] Understandably, when the speed loop outputs a large speed regulation duty cycle within a certain control cycle, and the current prediction model determines that the current will exceed the current limiting threshold in the next control cycle, the main control unit obtains a smaller current limiting duty cycle through the aforementioned reverse solution process, and uses the smaller value as the target duty cycle. Since the target duty cycle is obtained by the reverse solution of the current prediction model, rather than directly turning off the bridge arm after overcurrent, the voltage regulation in adjacent control cycles is more continuous.

[0046] S150 controls the inverter bridge to drive the permanent magnet synchronous motor to operate based on the pulse width modulation signal generated by the target duty cycle and the current commutation state, so as to limit the phase current of the permanent magnet synchronous motor.

[0047] In this example, the main control unit analyzes the current commutation state to determine the upper bridge arm connected to the positive terminal of the bus, the lower bridge arm connected to the negative terminal of the bus, and the corresponding suspended bridge arm in the three-phase inverter bridge. For example, when the current commutation state is that phase A is connected to the positive terminal of the bus and phase B is connected to the negative terminal of the bus, the main control unit controls the upper bridge arm of phase A to output a PWM signal according to the target duty cycle, controls the lower bridge arm of phase B to conduct or operate in a complementary PWM mode, and controls the corresponding bridge arm of phase C to be in the off state. As another example, when the current commutation state is that phase C is connected to the positive terminal of the bus and phase A is connected to the negative terminal of the bus, the main control unit controls the upper bridge arm of phase C and the lower bridge arm of phase A to form a conducting phase loop, and keeps phase B in a suspended state.

[0048] The conduction time of the PWM signal is determined by the following formula: ; in, For PWM on-time, The PWM period is (e.g., 20μs to 200μs, with a specific example being 100μs). The target duty cycle.

[0049] The main control unit writes the target duty cycle into the PWM comparator register, causing the inverter bridge to output the corresponding modulation voltage according to the current commutation state within the current control cycle. After entering the next control cycle, the main control unit re-executes S110 to S150, updating the predicted current and target duty cycle based on the new operating parameters and the new commutation state.

[0050] According to the above implementation method, S110 enables the controller to obtain the bus voltage, bus current, rotor electrical angle, rotor electrical angular velocity, and commutation state within the current control cycle, providing input for model calculation; S120 ensures that each commutation state has a current prediction model corresponding to the conducting phase circuit, avoiding model mixing between different conducting phase combinations; S130 obtains the predicted current for the next control cycle through resistor voltage drop, back EMF influence, inductance conversion, and discrete integration, giving the current limiting judgment an advance amount; S140 reversely solves the target duty cycle when the predicted current exceeds the current limiting threshold, allowing the duty cycle to be continuously adjusted according to the motor's electrical state; S150 implements the target duty cycle into the PWM drive signal of the inverter bridge, enabling the phase current limiting action to be executed synchronously with the current commutation state. Thus, smooth control of the phase current of the permanent magnet synchronous motor can be achieved during the six-step commutation drive process.

[0051] In one implementation, a current prediction model corresponding to each commutation state is constructed based on the three-phase voltage equation and the three-phase flux linkage equation of the permanent magnet synchronous motor, including: The three-phase flux linkage equation is simplified based on the stator winding self-inductance to obtain the simplified flux linkage relationship. Substituting the simplified flux linkage relationship into the three-phase voltage equation, we obtain the dynamic relationship of the three-phase current. Obtain the conducting phase combination and the suspended phase corresponding to each commutation state under the six-step commutation control; Based on the dynamic relationship of three-phase current and the combination of conducting phases and floating phases corresponding to each commutation state, a current prediction model corresponding to each commutation state is constructed.

[0052] For example, the three-phase voltage equation is used to characterize the correspondence between the phase voltage, phase current, and phase flux linkage changes in each phase winding of a permanent magnet synchronous motor. The three-phase voltage equation is as shown in the previous example, and will not be repeated in this example.

[0053] The three-phase flux linkage equations may include stator winding self-inductance terms, stator winding mutual inductance terms, and permanent magnet flux linkage terms. When constructing the current prediction model, it is first determined whether the stator winding mutual inductance meets a preset simplification condition. The preset simplification condition is that the ratio of mutual inductance to self-inductance is less than a mutual inductance neglect threshold (e.g., 0.01 to 0.2, 0.05 in a specific embodiment). When the preset simplification condition is met, the three-phase flux linkage equations are simplified based on the stator winding self-inductance to obtain a simplified flux linkage relationship (as shown in the previous example). Through the above processing, the online computational load caused by the mutual inductance term can be reduced while retaining the influence of stator resistance, inductance, and permanent magnet back EMF.

[0054] Furthermore, by substituting the simplified flux linkage relationship into the three-phase voltage equation and taking the time derivative of the permanent magnet flux linkage term, the dynamic relationship of the three-phase current is obtained (as shown in the previous example). Therefore, a current prediction model consistent with the conducting phase loop can be constructed for different commutation states.

[0055] Furthermore, the conducting phase combinations and suspended phases corresponding to each commutation state under the six-step commutation control are obtained. The conducting phase combinations include positive conducting phases connected to the positive end of the bus and negative conducting phases connected to the negative end of the bus. Suspended phases are phases that do not participate in the conducting loop under the current commutation state. The commutation states under the six-step commutation control can be determined according to a preset commutation table (for example, the commutation table includes six states: A+B-, A+C-, B+C-, B+A-, C+A-, and C+B-; in specific embodiments, these six states are sequentially switched). Wherein, A+B- indicates that phase A is connected to the positive end of the bus, phase B is connected to the negative end of the bus, and phase C is a suspended phase; A+C- indicates that phase A is connected to the positive end of the bus, phase C is connected to the negative end of the bus, and phase B is a suspended phase; the remaining states are determined according to the same rules.

[0056] Furthermore, for each commutation state, the dynamic relationship of the three-phase current is converted into the dynamic relationship of the bus current in the conducting phase loop. Taking the A+ B- state as an example, phase A and phase B form a conducting phase loop, and phase C is left floating, satisfying: ; in, This refers to the bus voltage. Let be the bus current. Substituting the above relationship into the three-phase current dynamic relationship, we obtain the bus current dynamic relationship under states A+ and B-: ; Discretizing the dynamic relationship of the bus current yields the current prediction model corresponding to states A+ and B-: ; in, The predicted current for the next control cycle; To control the cycle duration (e.g., from 20μs to 200μs, with a specific example of 100μs); It is the back potential characterization quantity corresponding to the A+ B- state.

[0057] For the A+ C- state, phase A and phase C form a conducting phase loop, while phase B is left floating, allowing for the construction of a corresponding current prediction model: .

[0058] For the B+ and C- states, a corresponding current prediction model can be constructed: .

[0059] For states B+ A-, C+ A-, and C+ B-, corresponding current prediction models are constructed according to the phase voltage difference, phase current direction, and back EMF phase identifier between the positive and negative conducting phases. Specifically, the back EMF phases of the positive and negative conducting phases are first determined, and then the back EMF components of the two conducting phases are combined according to the direction of the conduction loop to obtain the corresponding back EMF characterization quantity. Then, the bus voltage, resistance voltage drop term, back EMF influence term, and inductor current change term are written into the discretization relation to form the current prediction model for the corresponding commutation state.

[0060] For example, each commutation state in the preset commutation table can be configured as a model record. The model record includes the state number, positive conducting phase, negative conducting phase, floating phase, positive conducting back EMF phase, negative conducting back EMF phase, and current prediction model index. The main control unit reads the motor parameters during the initialization phase. The system generates six current prediction models based on the commutation table. During operation, the corresponding current prediction model is called based on the current commutation state, and the current operating parameters are substituted to make predictions.

[0061] According to the above implementation method, the three-phase current dynamic relationship with physical meaning is first obtained from the three-phase voltage equation and the three-phase flux linkage equation. Then, the three-phase current dynamic relationship is transformed according to different combinations of conducting phases and floating phases in the six-step commutation, so that the current prediction model corresponding to each commutation state is matched with the actual conducting phase circuit. In this way, during motor operation, the controller can select the corresponding model for current prediction according to the current commutation state, avoiding the inconsistency between the predicted quantity and the actual current path caused by using the same model to process different conducting circuits. This provides a model basis for subsequent duty cycle reverse calculation and phase current smoothing control.

[0062] In one implementation, operating parameters and the current commutation state are input into a target current prediction model to predict the current for the next control cycle. The operating parameters include bus voltage, bus current, rotor electrical angle, and rotor electrical angular velocity. The target current prediction model includes a resistance voltage drop term, a back EMF term, an inductance conversion term, a voltage margin term, and discrete prediction terms, including: Based on the back EMF phase identifier corresponding to the current commutation state, the rotor electrical angle is phase matched, and the back EMF term is used to perform trigonometric function mapping on the phase-matched rotor electrical angle to obtain the back EMF characterization quantity corresponding to the current commutation state. The rotor electric angular velocity and permanent magnet flux linkage are input into the back EMF term to obtain the back EMF amplitude. The back EMF amplitude and the back EMF characterization quantity are summed to obtain the back EMF influence quantity corresponding to the conducting phase circuit. By converting the voltage drop of the bus current according to the stator resistance corresponding to the conduction phase identifier using the resistance voltage drop term, the resistance voltage drop corresponding to the conduction phase circuit can be obtained. By inputting the bus voltage, back EMF influence, and resistance voltage drop into the voltage margin term, the effective voltage of the conducting phase circuit within the current control cycle is obtained. The effective voltage is converted into the rate of change of current by the inductance conversion term, and the rate of change of current of the conducting phase circuit in the current control cycle is obtained. The current change rate and the duration of the current control cycle are input into the discrete prediction term to perform discrete integration on the current change rate. The result of the discrete integration is then superimposed with the bus current to obtain the predicted current for the next control cycle.

[0063] For example, in the target current prediction model, the resistance voltage drop term is used to calculate the voltage drop generated by the stator winding resistance in the conducting phase circuit, the back EMF term is used to calculate the influence of the back EMF generated by the permanent magnet flux linkage as the rotor rotates, the voltage margin term is used to calculate the voltage acting on the equivalent inductance of the conducting phase circuit, the inductance conversion term is used to convert the effective voltage into the current change rate, and the discrete prediction term is used to convert the current change rate in the current control cycle into the predicted current in the next control cycle.

[0064] Specifically, the main control unit first reads the conduction phase identifier corresponding to the current commutation state. The conduction phase identifier includes a positive conduction phase identifier and a negative conduction phase identifier. The positive conduction phase identifier is used to characterize the phase connected to the positive end of the bus, and the negative conduction phase identifier is used to characterize the phase connected to the negative end of the bus. For example, when the current commutation state is A+ B-, the positive conduction phase is phase A, the negative conduction phase is phase B, and phase C is a suspended phase; when the current commutation state is C+ A-, the positive conduction phase is phase C, the negative conduction phase is phase A, and phase B is a suspended phase.

[0065] Furthermore, a back-potential phase identifier is assigned to each phase. The back-potential phase identifier for phase A is 0, and the back-potential phase identifier for phase B is... The phase of the opposite potential of C is marked as The back EMF phase identifier can be stored in the commutation state table, which includes the state number, positive conducting phase, negative conducting phase, floating phase, positive conducting phase identifier, and negative conducting phase identifier.

[0066] Given the current commutation state as A+ B-, phase matching is performed on the rotor electrical angles to obtain the angle values ​​corresponding to A and B: ;in, The rotor electrical angle; Let A be the angle of the opposite potential. The angle is the opposite potential of B.

[0067] Then, the back EMF term is sinusoidally mapped onto the phase-matched angle quantity to obtain the back EMF characterization quantity corresponding to the current commutation state: ; Substituting the states A+ and B-, we get: ; in, This is a characterization of the back potential. For the A+ C- state, the characterization of the back potential is... For the B+ C- state, the back potential is characterized by: Other commutation states are treated the same way as the positive and negative conduction phases. In this way, the back EMF calculation corresponds to the actual conduction phase combination, avoiding the introduction of floating back EMF into the conduction phase circuit calculation.

[0068] Furthermore, by inputting the rotor electric angular velocity and permanent magnet flux linkage into the back EMF term, the magnitude of the back EMF is obtained: ;in, This is the magnitude of the back electromotive force; The flux linkage is a permanent magnet (e.g., 0.001 Wb to 0.2 Wb, with a specific embodiment being 0.01 Wb). Let be the rotor electric angular velocity. The back EMF term is then multiplied by the back EMF amplitude and the back EMF characteristic quantity to obtain the back EMF influence quantity corresponding to the conducting phase circuit: ; in, This is the quantity affected by the back electromotive force (EMF). Using multiplication can maintain dimensional consistency; that is, the magnitude of the back EMF is a voltage quantity, and the quantity representing the back EMF is a dimensionless trigonometric function result. The product of the two is still a voltage quantity.

[0069] Furthermore, the resistance voltage drop term reads the stator resistance corresponding to the positive and negative conduction phases according to the conduction phase identification, and performs voltage drop conversion on the bus current: ; in, This is the voltage drop across the resistor; For the positive conducting phase stator resistance, The stator resistance is for the negative conducting phase; This represents the bus current. If the three-phase stator resistances are the same, then: ; in, This refers to the single-phase stator resistance (e.g., 0.01Ω to 20Ω, specifically 0.2Ω in this embodiment). For example, in the A+B- state, the resistance voltage drop term reads the stator resistance of phase A and phase B, and the voltage drop is calculated using the bus current as the current in the conducting phase loop. This separates the voltage consumed by the winding resistance from the bus voltage, providing a basis for subsequent calculations of the inductor voltage.

[0070] Furthermore, the voltage margin term receives the bus voltage, the back EMF effect, and the resistance voltage drop, and calculates the effective voltage acting on the equivalent inductance of the conducting phase loop during the current control cycle: ; in, Effective voltage quantity; The value represents the bus voltage. The sign of the back EMF influence quantity in this formula is already represented by the back EMF characteristic quantity corresponding to the current commutation state. That is to say, under different commutation states, the back EMF term forms a corresponding positive and negative relationship through phase matching and trigonometric function mapping, and the voltage margin term obtains the effective voltage quantity according to a unified calculation rule.

[0071] Understandably, when the direction of the back EMF opposes the rise of the current, the back EMF characteristic quantity reduces the effective voltage; when the direction of the back EMF matches the current change direction, the back EMF characteristic quantity increases the effective voltage.

[0072] Furthermore, the inductance conversion term converts the effective voltage to the rate of change of current based on the equivalent inductance of the conducting phase circuit. For a six-step commutation circuit with two conducting phases and identical inductance in all three phases, the equivalent inductance of the conducting phase circuit is the sum of the stator inductances of the two phases: ; in, To conduct the equivalent inductance of the phase circuit, For positive conduction stator inductance, The stator inductor is the negative conducting phase.

[0073] If the three-phase stator inductances are the same, then: The rate of change of current is: ; in, The inductance is a single-phase stator inductance (e.g., 0.01mH to 10mH, specifically 0.5mH in this embodiment). Through the inductance conversion term, the effective voltage can be converted into the rate of change of the current in the conducting phase loop within the current control cycle, ensuring that the current prediction conforms to the physical relationship between the inductor voltage and the rate of change of current.

[0074] Furthermore, the discrete prediction term is discretely integrated over the current change rate according to the duration of the current control cycle to obtain the current change: ; in, This refers to the change in current. The duration of the current control cycle (e.g., 20μs to 200μs, 100μs in a specific embodiment).

[0075] The discrete prediction term is then superimposed with the bus current to obtain the predicted current for the next control cycle: ;in, This is the predicted current for the next control cycle.

[0076] As a concrete example, when the current commutation state is A+ B-, the target current prediction model can be written as: ; When the current commutation state is A+ C-, the target current prediction model can be written as: ; When the current commutation state is B+ C-, the target current prediction model can be written as: ; For the B+ A-, C+ A-, and C+ B- states, a back potential characterization quantity is generated according to the back potential phase identifier of the corresponding positive and negative conduction phases, and then substituted into the same prediction structure.

[0077] According to the above implementation method, the target current prediction model is not a black-box model that only receives operating parameters and outputs results. Instead, it sequentially completes the physical quantity conversion between voltage, current, and back EMF through back EMF, resistance voltage drop, voltage margin, inductance conversion, and discrete prediction terms. The back EMF term aligns the prediction process with the current commutation state and rotor position; the resistance voltage drop term reflects the consumption of available voltage by the winding resistance; the voltage margin term obtains the effective voltage acting on the inductor; the inductance conversion term converts the effective voltage into the current change rate; and the discrete prediction term converts the current change rate into the predicted current for the next control cycle. Therefore, the predicted current can reflect the actual electrical state of the currently conducting phase circuit, providing a clear calculation basis for subsequent reverse calculation of the target duty cycle based on the current limiting threshold.

[0078] In one implementation, the rotor electrical angle is phase-matched according to the back EMF phase identifier corresponding to the current commutation state, and the phase-matched rotor electrical angle is mapped by a trigonometric function using the back EMF term to obtain the back EMF characterization quantity corresponding to the current commutation state, including: Based on the conduction phase identifier corresponding to the current commutation state, determine the positive conduction phase connected to the positive end of the bus and the negative conduction phase connected to the negative end of the bus in the current control cycle; Based on the back EMF phase identifier corresponding to the current commutation state, the rotor electrical angle is phase shifted in the positive conduction phase and the negative conduction phase respectively to obtain the first back EMF angle corresponding to the positive conduction phase and the second back EMF angle corresponding to the negative conduction phase. The first and second back EMF angle quantities are converted into sinusoidal values ​​to obtain the first and second back EMF components. The first and second back EMF components are summed to obtain the back EMF characteristic quantity corresponding to the current commutation state.

[0079] For example, the conducting phase identifier refers to the identifier used to characterize the phase winding participating in the current conducting loop and its connection direction to the bus in the six-step commutation control. The conducting phase identifier includes a positive conducting phase identifier and a negative conducting phase identifier. The positive conducting phase identifier corresponds to the phase winding connected to the positive end of the bus, and the negative conducting phase identifier corresponds to the phase winding connected to the negative end of the bus. The conducting phase identifier can be determined by the commutation state table, which can be pre-stored in the memory of the main control unit (for example, the commutation state table includes six states: A+B-, A+C-, B+C-, B+A-, C+A-, and C+B-; in specific embodiments, the above six states are switched sequentially in a cyclic manner).

[0080] Specifically, the main control unit reads the current commutation state and queries the commutation state table to determine the positive and negative conducting phases within the current control cycle. For example, when the current commutation state is A+ B-, the positive conducting phase is phase A, the negative conducting phase is phase B, and phase C is a floating phase; when the current commutation state is C+ A-, the positive conducting phase is phase C, the negative conducting phase is phase A, and phase B is a floating phase. Through this process, it is clear that the back EMF calculation is performed only on the two phase windings in the current conducting loop, avoiding the introduction of the back EMF component of the floating phase into the conducting loop calculation.

[0081] For example, the back EMF phase identifier refers to the phase offset of the back EMF of each phase winding relative to the rotor electrical angle. For a three-phase permanent magnet synchronous motor, the back EMF phase identifiers of phases A, B, and C can be set as follows: ; in, The opposite potential phase identifier for A. This is the phase identifier for the opposite potential of B. The phase identifier is the opposite potential phase identifier for phase C. This phase identifier can also be stored in degrees (e.g., 0° for phase A, -120° for phase B, and +120° for phase C), and converted to radians during calculation. The main control unit can store the phase identifier using a lookup table (e.g., lookup accuracy of 1° to 10°, 1° in a specific embodiment), or it can be configured as a constant in the program.

[0082] Furthermore, the main control unit performs phase shifting processing on the rotor electrical angle based on the back EMF phase identifiers corresponding to the positive and negative conduction phases. If the back EMF phase identifier corresponding to the positive conduction is... The phase identifier corresponding to negative conduction is: Then the first back electromotive force angle and the second back electromotive force angle are respectively: ; in, This is the angle of the first back electromotive force corresponding to positive conduction. The second back electromotive force angle is the corresponding value for negative conduction. For rotor electrical angle, This is a phase indicator for the opposite potential during positive conduction. This indicates the phase of the negative conduction opposite potential.

[0083] For example, when the current commutation state is A+ B-, , ,but: ; If the current commutation state is C+ A-, then... , ,but: ; By using phase shift processing, the same rotor electrical angle can be converted into the back EMF angle corresponding to different phase windings, so that the subsequent trigonometric function mapping is consistent with the spatial phase relationship of the three-phase windings.

[0084] Furthermore, the back potential term is sinusoidally converted to the first and second back potential angle quantities respectively, yielding the first and second back potential components: ; in, This is the first back potential component. This is the second back electromotive force component. Sine conversion can be achieved by the main control unit calling the trigonometric function operation unit, or by using a preset sine lookup table. The angle range of the preset sine lookup table is one electrical angle cycle (e.g., 0 to...). (Specific implementations range from 0 to 360°), and the number of entries can be set according to the control precision (e.g., 256 to 4096 entries, specific implementations range from 1024 entries).

[0085] When the angular amount after phase shift exceeds one electrical angle period, period normalization can be performed first: ;in, The normalized angle quantity. This represents the angle quantity after phase shift. Through period normalization, the trigonometric function mapping can be executed within one electrical angle period, facilitating table lookup calculations by the embedded controller.

[0086] Furthermore, by summing the first and second back EMF components, we obtain the back EMF characteristic quantity corresponding to the current commutation state: ; in, This is the back EMF characteristic quantity corresponding to the current commutation state. The back EMF characteristic quantity is dimensionless and is used to characterize the combination relationship between the two opposite EMF components in the currently conducting phase circuit. Multiplying the back EMF characteristic quantity by the permanent magnet flux linkage and the rotor electric angular velocity yields the back EMF influence quantity corresponding to the conducting phase circuit.

[0087] For example, in the A+ B- state, the back potential is characterized by: ; In the A+ C- state, the back potential is characterized by: ; In the B+ C- state, the back potential is characterized by: ; In the B+ A- state, the back potential is characterized by: ; In the C+ A- state, the back potential is characterized by: ; In the C+ B- state, the back potential is characterized by: ; It should be noted that the sign relationship of the above-mentioned back EMF characterization quantities is consistent with the definition of phase voltage difference in the current prediction model. If the target current prediction model uses the positive conduction phase voltage minus the negative conduction phase voltage as the phase-to-phase voltage, then the back EMF characterization quantities are configured according to the direction of that phase-to-phase voltage; if the target current prediction model uses the opposite direction of the phase-to-phase voltage, then the sign of the back EMF component is adjusted accordingly. This ensures that the positive and negative directions of the back EMF characterization quantities are consistent with those in the voltage margin calculation, avoiding deviations in the predicted current due to inconsistent sign directions.

[0088] According to the above implementation method, the positive and negative conducting phases are first determined based on the current commutation state. Then, the rotor electrical angle is phase-shifted according to the back EMF phase identifier corresponding to each conducting phase. The back EMF components of the two conducting phases are obtained through sinusoidal conversion. Finally, the back EMF components of the two conducting phases are combined into a back EMF characterization quantity. Thus, the back EMF characterization quantity can change synchronously with the current commutation state and rotor electrical angle, making the back EMF term in the current prediction model correspond to the actual conducting phase circuit, thereby improving the consistency of subsequent calculations for predicting current and duty cycle in reverse order.

[0089] In one implementation, the target current prediction model is solved inversely based on the current limiting threshold and operating parameters to obtain the target duty cycle, including: The current limiting threshold and operating parameters are used as the predicted current input to the target current prediction model, and the target current prediction model is algebraically transformed so that the target duty cycle is used as the variable to be determined. Based on the operational relationships between the resistance voltage drop term, back EMF term, inductance conversion term, voltage margin term and discrete prediction term in the target current prediction model, the variable to be solved is solved in reverse to obtain the current limiting duty cycle. Calculate the speed adjustment duty cycle based on the target speed and the actual speed of the permanent magnet synchronous motor; The minimum value between the current limiting duty cycle and the speed regulation duty cycle is taken as the target duty cycle.

[0090] For example, the current limiting threshold can be determined based on the rated phase current of the permanent magnet synchronous motor, the rated current of the inverter bridge switching devices, the winding thermal limitation, or the protection level of the control system (e.g., 0.5 to 1.5 times the rated phase current, specifically 1.0 times the rated phase current in a specific embodiment). The target duty cycle ranges from 0 to 1.

[0091] Specifically, when the predicted current exceeds the current limiting threshold, the main control unit uses the current limiting threshold as the target predicted current for the next control cycle and replaces the bus voltage in the target current prediction model with the equivalent bus voltage modulated by the duty cycle. The equivalent bus voltage can be expressed as: ; in, This is the equivalent bus voltage. To limit the duty cycle, This is the bus voltage.

[0092] The target current prediction model can be expressed as: ; in, To predict the current, Bus current, This is the voltage drop across the resistor. The effect of back electromotive force. To conduct the equivalent inductance of the phase circuit, This represents the duration of the current control cycle. For a six-step commutation circuit with two phases conducting and all three phases having identical parameters, , .

[0093] When solving in reverse, Replace with rate limiting threshold and will Replace with ,get: ; Treating variables By performing algebraic transformations, the current-limiting duty cycle is obtained: ; When the three-phase stator resistance and stator inductance are the same, it can be written as: ; in, For single-phase stator resistance, It is a single-phase stator inductor. It is a permanent magnet flux linkage. The rotor's electric angular velocity, This is the back electromotive force characteristic of the current commutation state.

[0094] Furthermore, the duty cycle of the current limiting is limited: ; in, This is the current-limited duty cycle after amplitude limiting. Through this amplitude limiting process, the current-limited duty cycle can be made to meet the effective duty cycle range of the pulse width modulation signal.

[0095] Furthermore, the main control unit calculates and adjusts the duty cycle based on the target speed and the actual speed of the permanent magnet synchronous motor. The target speed can be given by the host controller, throttle pedal signal, actuator position controller, or preset operating command (e.g., 100 r / min to 20000 r / min, specifically 3000 r / min in this embodiment). The actual speed can be obtained by converting the position signal output by the rotor position detection unit. The main control unit calculates the speed deviation: ; in, For speed deviation, For the target speed, This is the actual rotational speed.

[0096] Input the speed deviation into the speed regulator to obtain the speed regulation duty cycle: ; in, To adjust the duty cycle for speed, This is a scaling factor (e.g., 0.001 to 1, 0.02 in a specific embodiment). The integral coefficient is used (e.g., 0.001 to 10, 0.1 in a specific embodiment). The speed regulation duty cycle is also limited within the range of 0 to 1.

[0097] Finally, the main control unit compares the current-limiting duty cycle after amplitude limiting with the speed regulation duty cycle, and takes the smaller of the two as the target duty cycle: ;in, The target duty cycle.

[0098] Understandably, when the current commutation state is A+ B-, the main control unit first calculates the back EMF influence, resistance voltage drop, and equivalent inductance of the conducting phase circuit based on the bus voltage, bus current, rotor electrical angle, rotor electrical angular velocity, stator resistance, stator inductance, and permanent magnet flux linkage. When the predicted current exceeds the current limiting threshold, the current limiting threshold is substituted into the target current prediction model, and the duty cycle is solved as an unknown to obtain the current limiting duty cycle. If the speed regulation duty cycle output by the speed regulator is greater than the current limiting duty cycle, the current limiting duty cycle is used as the target duty cycle; if the speed regulation duty cycle is less than or equal to the current limiting duty cycle, the speed regulation duty cycle is used as the target duty cycle.

[0099] According to the above implementation method, the current-limiting duty cycle is obtained by inversely solving the target current prediction model. The calculation process sequentially retains the physical relationships between resistor voltage drop, back EMF influence, inductance conversion, and discrete prediction, avoiding direct reduction of the duty cycle solely through empirical proportions. The speed regulation duty cycle is used to meet speed regulation requirements, while the current-limiting duty cycle is used to constrain the phase current growth in the next control cycle. The smaller of the two values ​​is taken as the target duty cycle, ensuring that the inverter bridge output voltage is simultaneously constrained by speed control requirements and phase current limiting requirements, thereby maintaining the continuity of duty cycle changes during the current-limiting process.

[0100] In one implementation, the speed regulation duty cycle is calculated based on the target speed and the actual speed of the permanent magnet synchronous motor, including: In response to a target speed command, which carries the target speed; The acquired speed detection signal is sampled and filtered to obtain the actual speed of the permanent magnet synchronous motor; The speed deviation is determined based on the difference between the target speed and the actual speed; The speed deviation is input into the speed regulator, which performs proportional-integral adjustment on the speed deviation to obtain the speed loop output. The speed loop output is subjected to duty cycle mapping and limiting processing to obtain the speed regulation duty cycle.

[0101] For example, a target speed command refers to a control command used to indicate the desired operating speed of a permanent magnet synchronous motor. The target speed command can be generated by a host controller, vehicle controller, actuator control unit, or local control program. The target speed can be expressed in revolutions per minute or radians per second (e.g., 100 r / min to 20000 r / min, specifically 3000 r / min in this embodiment).

[0102] Specifically, after receiving the target speed command, the main control unit parses the target speed from the data field of the command. If the target speed command is transmitted in the form of a communication message, the main control unit reads the target speed field according to a preset message format and converts the target speed field into an actual physical quantity based on a proportionality coefficient. The preset message format may include a command identifier, a target speed field, and a check field (for example, the target speed field is a two-byte or four-byte integer, and the proportionality coefficient is from 1 r / min / bit (revolutions per minute per bit, the number of revolutions per minute corresponding to each bit) to 10 r / min / bit, with a specific embodiment being 1 r / min / bit). Through this processing, the speed regulator can have a clear speed control target.

[0103] For example, a speed detection signal refers to a detection signal that reflects changes in rotor position or rotor speed. The speed detection signal can originate from a Hall sensor, encoder, resolver, or a sensorless estimation module. For instance, when using an encoder, the main control unit reads the encoder count increment within a fixed sampling period; when using a Hall sensor, the main control unit records the time interval between adjacent Hall transition edges; when using a resolver, the main control unit reads the decoded rotor angle and calculates the angle change.

[0104] In one example, the actual rotational speed is calculated based on the mechanical angle at adjacent sampling times: ; in, The actual rotational speed at the current sampling moment; The mechanical angle at the current sampling moment; The mechanical angle at the previous sampling time; The speed sampling period is (e.g., 0.1ms (millisecond) to 10ms, with a specific implementation of 1ms).

[0105] When using a Hall sensor, the actual rotational speed can also be calculated based on the time interval between adjacent transition edges: ; in, The number of effective Hall effect jumps corresponding to one revolution of the rotor (e.g., 6 to 120, 24 in a specific embodiment). This is the time interval between adjacent valid transition edges.

[0106] Furthermore, to reduce the impact of sampling jitter on the speed regulation duty cycle, the main control unit filters the speed sample values. The filtering can employ a first-order low-pass filter: ; in, This is the actual rotational speed after filtering. The actual rotational speed after filtering at the previous sampling time. The filtering coefficient is (e.g., 0.5 to 0.98, 0.8 in a specific embodiment). The filtered actual rotational speed is used as the input for subsequent speed closed-loop calculation. Through sampling and filtering, quantization errors and instantaneous disturbances in the speed detection signal can be reduced.

[0107] Furthermore, the main control unit determines the speed deviation based on the difference between the target speed and the actual speed: ; in, This represents the rotational speed deviation at the current sampling moment. For the target speed, This is the actual speed after filtering. If speed direction control is used, the motor rotation direction can be determined based on the sign of the target speed, and the sign of the speed deviation can be used as the basis for acceleration or deceleration adjustment.

[0108] Furthermore, the speed deviation is input into the speed regulator. The speed regulator can be a proportional-integral (PI) regulator. The proportional term generates an instantaneous adjustment based on the current speed deviation, and the integral term generates a steady-state adjustment based on the accumulated speed deviation. ; in, This is the integral of the speed deviation. For the speed loop output, This is a scaling factor (e.g., 0.001 to 1, 0.02 in a specific embodiment). This is the integral coefficient (e.g., 0.001 to 10, 0.1 in a specific embodiment).

[0109] To prevent the integral from accumulating and causing the speed loop output to exceed the available duty cycle range, an integral limit can be applied to the integral: ; in, This is the integral amount after the amplitude is limited. and These are the upper and lower limits of the integral (e.g., -1000 to 1000, or -300 to 300 in a specific embodiment). The speed regulator uses the limited integral to calculate the speed loop output. Through proportional-integral control, the speed regulation duty cycle can simultaneously respond to the current speed deviation and the accumulated deviation.

[0110] Furthermore, duty cycle mapping and limiting are performed on the speed loop output. If the speed loop output already uses duty cycle dimensions, then the speed loop output is directly used as the duty cycle to be limited; if the speed loop output uses voltage dimensions, then the duty cycle is converted based on the bus voltage. ; in, To adjust the duty cycle for the limiting speed, This is the bus voltage.

[0111] Subsequently, the duty cycle of the speed limiter to be adjusted is limited to the preset duty cycle range: ; in, To adjust the duty cycle for speed, This is the upper limit of the duty cycle (e.g., 0.5 to 1, specifically 0.95 in this embodiment). This represents the lower limit of the duty cycle (e.g., 0 to 0.2, specifically 0 in this embodiment). In scenarios where dead time or safety margins for switching devices need to be maintained, the upper limit of the duty cycle is less than 1.

[0112] For example, if the main control unit receives a target speed command of 3000 r / min, and the encoder detects the current actual speed, which is filtered to be 2800 r / min, then the speed deviation is 200 r / min. The main control unit inputs this speed deviation into a proportional-integral controller to obtain the speed loop output, and maps the speed loop output to a speed regulation duty cycle within the range of 0 to 1. This speed regulation duty cycle is then compared with the current limiting duty cycle, and the smaller value is used in the generation of the PWM signal.

[0113] According to the above implementation method, the target speed is explicitly given by the target speed command, and the actual speed is obtained by sampling and filtering the speed detection signal. The speed deviation is adjusted by proportional-integral control to form the speed loop output, which is then mapped and limited by duty cycle to obtain the speed regulation duty cycle. In this way, the generation process of the speed regulation duty cycle has a clear data source, calculation path, and limiting rules, and can participate in the determination of the target duty cycle together with the current limiting duty cycle, so that the phase current limiting process does not deviate from the speed control requirements of the permanent magnet synchronous motor.

[0114] In one implementation, controlling the inverter bridge to drive the permanent magnet synchronous motor based on a pulse width modulation signal generated according to the target duty cycle and the current commutation state includes: Generate a pulse width modulation signal corresponding to the current commutation state based on the target duty cycle; The pulse width modulation signal is output to the power switching devices in the inverter bridge corresponding to the positive and negative conduction phases to control the positive conduction phase to connect to the positive terminal of the bus, the negative conduction phase to connect to the negative terminal of the bus, and keep the corresponding bridge arm of the floating phase off. A modulation voltage corresponding to the target duty cycle is applied to the permanent magnet synchronous motor through the conduction loop between the positive and negative conduction phases to drive the permanent magnet synchronous motor to run.

[0115] For example, the inverter bridge is a three-phase inverter bridge, including phase A arm, phase B arm, and phase C arm. Each phase arm includes an upper arm power switch and a lower arm power switch. The upper arm power switch is used to connect the corresponding phase winding to the positive terminal of the DC bus, and the lower arm power switch is used to connect the corresponding phase winding to the negative terminal of the DC bus. The power switch can be a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), an IGBT (Insulated Gate Bipolar Transistor), or a silicon carbide power device (e.g., a voltage rating from 40V to 1200V, specifically a 100V MOSFET).

[0116] In this example, the main control unit reads the positive conducting phase, negative conducting phase, and floating phase from a preset drive table based on the current commutation state. The preset drive table may include six commutation states (e.g., A+ B-, A+ C-, B+ C-, B+ A-, C+ A-, C+ B-; in specific embodiments, these six states are switched sequentially). Here, A+ B- indicates that the upper arm of phase A participates in connecting to the positive end of the busbar, the lower arm of phase B participates in connecting to the negative end of the busbar, and phase C is a floating phase.

[0117] Furthermore, the main control unit generates a pulse width modulation (PWM) signal based on the target duty cycle. The period of the PWM signal is a preset PWM period (e.g., 20μs to 200μs, 100μs in a specific embodiment), and the target duty cycle ranges from 0 to 1. The main control unit calculates the effective on-time within the current PWM period based on the target duty cycle: ; in, To ensure effective conduction time, For the target duty cycle, This is the PWM period.

[0118] When using a counter comparison method to generate the PWM signal, the main control unit calculates the comparison register value based on the PWM counting period value and the target duty cycle: ; in, For the PWM compare register value, This is the value of the PWM period register. The main control unit will... Write the PWM channel corresponding to the current commutation state to enable the corresponding power switching device to conduct according to the target duty cycle.

[0119] Furthermore, the main control unit outputs the pulse width modulation signal to the power switching devices in the inverter bridge corresponding to the positive and negative conduction phases. Taking the A+ B- state as an example, the main control unit outputs a PWM signal to the upper bridge arm power switching device of phase A, controlling the phase A winding to connect to the positive terminal of the bus during the effective PWM conduction time; at the same time, it controls the lower bridge arm power switching device of phase B to conduct or outputs the corresponding drive signal in a complementary manner, so that the phase B winding connects to the negative terminal of the bus; and sets both the upper and lower bridge arms of phase C to the off state, so that phase C remains floating.

[0120] In one implementation, the upper arm of the positive conducting phase uses PWM modulation, the lower arm of the negative conducting phase remains on, and the upper and lower arms of the floating phase remain off. In another implementation, the upper arm of the positive conducting phase and the lower arm of the negative conducting phase use complementary PWM modulation, with a dead time added between the complementary signals. The dead time is used to prevent the power switching devices on and off of the same arm from being turned on simultaneously. The dead time can be set according to the turn-on and turn-off times of the power switching devices (e.g., from 100 ns (nanoseconds) to 5 μs, with a specific embodiment being 500 ns).

[0121] Furthermore, a conduction loop is formed between the positive and negative conducting phases. Taking the A+ B- state as an example, when the upper arm of phase A and the lower arm of phase B are in the conducting state, the current path is: positive terminal of the DC bus, upper arm of phase A, phase A winding, phase B winding, lower arm of phase B, and negative terminal of the DC bus. The main control unit changes the equivalent voltage across this conduction loop within one PWM cycle by adjusting the target duty cycle. The equivalent modulation voltage can be expressed as: ; in, For modulation voltage, This is the bus voltage. This modulated voltage acts on the conducting phase circuit corresponding to the current commutation state to drive the permanent magnet synchronous motor.

[0122] For example, when the current commutation state is C+ A-, the main control unit outputs a PWM signal corresponding to the target duty cycle to the upper arm of phase C, controlling the lower arm of phase A to conduct and controlling the upper and lower arms of phase B to turn off. At this time, the current path is: positive terminal of the DC bus, upper arm of phase C, phase C winding, phase A winding, lower arm of phase A, and negative terminal of the DC bus. When the next commutation state arrives, the main control unit reselects the arms to participate in the drive based on the new current commutation state and applies the same target duty cycle generation logic to the new combination of conducting phases.

[0123] To ensure that the power device drive requirements are met during bridge arm switching, the main control unit first shuts down the power switching device corresponding to the previous commutation state at the moment of commutation, and then turns on the power switching device corresponding to the next commutation state after a preset commutation protection time. The preset commutation protection time can be set according to the power device turn-off delay and the driver response time (e.g., 100ns to 10μs, specifically 1μs in this embodiment). This process avoids abnormal short circuits caused by overlapping states of different bridge arms during commutation.

[0124] According to the above implementation method, the main control unit first determines the effective conduction time of the PWM based on the target duty cycle, and then selects the power switching devices corresponding to the positive conduction phase, negative conduction phase, and floating phase based on the current commutation state, so that the inverter bridge forms a defined conduction loop according to the six-step commutation logic. Since the target duty cycle is derived from the current limiting calculation and speed regulation calculation of the current prediction model, the modulation voltage output by the inverter bridge corresponds to the current current limiting requirement and the commutation state, thereby enabling the permanent magnet synchronous motor to receive the modulation voltage according to the target duty cycle during operation, achieving smooth limitation of the phase current.

[0125] Figure 2 This is a structural block diagram of a phase current limiting system for a permanent magnet synchronous motor according to an embodiment of the present invention.

[0126] like Figure 2 As shown, the phase current limiting system of the permanent magnet synchronous motor may include: The acquisition module 510 is used to acquire the operating parameters and current commutation status of the permanent magnet synchronous motor in the current control cycle; The current prediction model construction module 520 is used to construct current prediction models corresponding to each commutation state based on the three-phase voltage equation and the three-phase flux linkage equation of the permanent magnet synchronous motor, and to determine the target current prediction model from each of the current prediction models based on the current commutation state. The prediction module 530 is used to input the operating parameters and the current commutation state into the target current prediction model to predict the predicted current of the next control cycle. The reverse solving module 540 is used to reverse solve the target current prediction model based on the current limiting threshold and the operating parameters if the predicted current is greater than the preset current limiting threshold, so as to obtain the target duty cycle. The phase current limiting module 550 is used to control the inverter bridge to drive the permanent magnet synchronous motor to operate according to the target duty cycle and the current commutation state, so as to limit the phase current of the permanent magnet synchronous motor.

[0127] In one implementation, the current prediction model building module includes: The equation simplification unit is used to simplify the three-phase flux linkage equation based on the stator winding self-inductance to obtain the simplified flux linkage relationship; The substitution unit is used to substitute the simplified magnetic flux relationship into the three-phase voltage equation to obtain the dynamic relationship of the three-phase current. The acquisition unit is used to acquire the conducting phase combination and the suspended phase corresponding to each of the six-step commutation control states; The construction unit is used to construct a current prediction model corresponding to each commutation state based on the dynamic relationship of the three-phase current and the combination of conducting phases and floating phases corresponding to each commutation state.

[0128] In one embodiment, the operating parameters include bus voltage, bus current, rotor electrical angle, and rotor electrical angular velocity; the target current prediction model includes a resistance voltage drop term, a back EMF term, an inductance conversion term, a voltage margin term, and a discrete prediction term; and the prediction module includes: The matching and mapping unit is used to perform phase matching on the rotor electrical angle according to the back EMF phase identifier corresponding to the current commutation state, and to perform trigonometric function mapping on the phase-matched rotor electrical angle through the back EMF term to obtain the back EMF characterization quantity corresponding to the current commutation state. The back EMF term unit is used to input the rotor electric angular velocity and permanent magnet flux linkage into the back EMF term to obtain the back EMF amplitude, and to sum the back EMF amplitude and the back EMF characterization quantity to obtain the back EMF influence quantity corresponding to the conducting phase circuit. The voltage drop conversion unit is used to convert the bus current to voltage drop based on the stator resistance corresponding to the conduction phase identifier using the resistance voltage drop term, so as to obtain the resistance voltage drop corresponding to the conduction phase circuit. The voltage margin term unit is used to input the bus voltage, the back EMF influence amount and the resistor voltage drop amount into the voltage margin term to obtain the effective voltage amount of the conducting phase circuit in the current control cycle; The inductance conversion unit is used to convert the effective voltage quantity into the current change rate through the inductance conversion item, so as to obtain the current change rate of the conducting phase circuit in the current control cycle. The discrete prediction term unit is used to input the current change rate and the duration of the current control cycle into the discrete prediction term, to perform discrete integration processing on the current change rate, and to superimpose the discrete integration processing result with the bus current to obtain the predicted current for the next control cycle.

[0129] In one embodiment, the matching and mapping unit includes: The conduction phase determination subunit is used to determine the positive conduction phase connected to the positive end of the bus and the negative conduction phase connected to the negative end of the bus in the current control cycle based on the conduction phase identifier corresponding to the current commutation state. The phase offset processing subunit is used to perform phase offset processing on the rotor electrical angle on the positive conduction phase and the negative conduction phase respectively according to the back EMF phase identifier corresponding to the current commutation state, so as to obtain the first back EMF angle corresponding to the positive conduction and the second back EMF angle corresponding to the negative conduction. The sinusoidal conversion subunit is used to perform sinusoidal conversion on the first back EMF angle and the second back EMF angle respectively to obtain the first back EMF component and the second back EMF component. The summation subunit is used to sum the first back EMF component and the second back EMF component to obtain the back EMF characterization quantity corresponding to the current commutation state.

[0130] In one implementation, the inverse solver module includes: An algebraic deformation unit is used to input the current limiting threshold and the operating parameters as predicted current into the target current prediction model, and to perform algebraic deformation on the target current prediction model so that the target duty cycle is used as the variable to be determined. The variable inverse solution unit is used to solve the variable inversely according to the operational relationship between the resistance voltage drop term, back EMF term, inductance conversion term, voltage margin term and discrete prediction term in the target current prediction model, and to obtain the current limiting duty cycle. The speed regulation duty cycle calculation unit is used to calculate the speed regulation duty cycle based on the target speed and the actual speed of the permanent magnet synchronous motor. The target duty cycle calculation unit is used to take the minimum value between the current limiting duty cycle and the speed regulation duty cycle as the target duty cycle.

[0131] In one embodiment, the speed regulation duty cycle calculation unit includes: The instruction response subunit is used to respond to a target speed instruction, wherein the target speed instruction carries the target speed. The sampling and filtering processing subunit is used to sample and filter the acquired speed detection signal to obtain the actual speed of the permanent magnet synchronous motor. The speed deviation calculation subunit is used to determine the speed deviation based on the difference between the target speed and the actual speed; The proportional-integral adjustment subunit is used to input the speed deviation into the speed regulator, and the speed regulator performs proportional-integral adjustment on the speed deviation to obtain the speed loop output. The mapping and limiting processing subunit is used to perform duty cycle mapping and limiting processing on the speed loop output to obtain the speed adjustment duty cycle.

[0132] In one embodiment, the phase current limiting module includes: A pulse width modulation signal generation unit is used to generate a pulse width modulation signal corresponding to the current commutation state based on the target duty cycle. The control unit is used to output the pulse width modulation signal to the power switching devices in the inverter bridge corresponding to the positive conduction phase and the negative conduction phase, so as to control the positive conduction phase to be connected to the positive terminal of the bus, the negative conduction phase to be connected to the negative terminal of the bus, and keep the corresponding bridge arm of the floating phase off. The drive unit is used to apply a modulation voltage corresponding to the target duty cycle to the permanent magnet synchronous motor through the conduction loop between the positive conduction phase and the negative conduction phase, so as to drive the permanent magnet synchronous motor to run.

[0133] The specific functions and examples of each module and submodule of the system in this embodiment of the invention can be found in the relevant descriptions of the corresponding steps in the above method embodiments, and will not be repeated here.

[0134] The acquisition, storage, and application of user personal information involved in the technical solution of this invention all comply with the provisions of relevant laws and regulations and do not violate public order and good morals.

[0135] This invention also provides a computer device, comprising: At least one processor; and a memory communicatively connected to said at least one processor; The memory stores instructions that can be executed by the at least one processor, which, when executed by the at least one processor, enables the at least one processor to perform the method described in any one of the embodiments of the present invention.

[0136] The beneficial effects of the computer device in this embodiment of the invention are equivalent to the beneficial effects of the phase current limiting method of the permanent magnet synchronous motor described above, and will not be repeated here.

[0137] This invention also provides a non-transitory computer-readable storage medium storing computer instructions, wherein the computer instructions are used to cause a computer to perform the method described in any one of the embodiments of this invention.

[0138] The beneficial effects of the storage medium of the present invention are equivalent to the beneficial effects of the phase current limiting method of the permanent magnet synchronous motor described above, and will not be repeated here.

[0139] Figure 3A schematic block diagram of an example computer device 800 that can be used to implement embodiments of the present invention is shown. Computer device 800 is intended to represent various forms of digital computers, such as laptop computers, desktop computers, workstations, personal digital assistants, servers, blade servers, mainframe computers, and other suitable computers. Computer device 800 may also represent various forms of mobile devices, such as personal digital assistants, cellular phones, smartphones, wearable devices, and other similar computing devices. The components shown herein, their connections and relationships, and their functions are merely illustrative and are not intended to limit the implementation of the invention described and / or claimed herein.

[0140] like Figure 3 As shown, the computer device 800 includes a computing unit 801, which can perform various appropriate actions and processes according to a computer program stored in a read-only memory (ROM) 802 or a computer program loaded from a storage unit 808 into a random access memory (RAM) 803. The RAM 803 may also store various programs and data required for the operation of the computer device 800. The computing unit 801, ROM 802, and RAM 803 are interconnected via a bus 804. An input / output (I / O) interface 805 is also connected to the bus 804.

[0141] Multiple components in computer device 800 are connected to I / O interface 805, including: input unit 806, such as keyboard, mouse, etc.; output unit 807, such as various types of monitors, speakers, etc.; storage unit 808, such as disk, optical disk, etc.; and communication unit 809, such as network card, modem, wireless transceiver, etc. Communication unit 809 allows computer device 800 to exchange information / data with other devices through computer networks such as the Internet and / or various telecommunications networks.

[0142] The computing unit 801 can be a variety of general-purpose and / or special-purpose processing components with processing and computing capabilities. Some examples of the computing unit 801 include, but are not limited to, a central processing unit (CPU), a graphics processing unit (GPU), various special-purpose artificial intelligence (AI) computing chips, various computing units running machine learning model algorithms, a digital signal processor (DSP), and any suitable processor, controller, microcontroller, etc. The computing unit 801 performs the various methods and processes described above, such as the phase current limiting method for a permanent magnet synchronous motor. For example, in some embodiments, the phase current limiting method for a permanent magnet synchronous motor can be implemented as a computer software program, which is tangibly contained in a machine-readable medium, such as storage unit 808. In some embodiments, part or all of the computer program can be loaded and / or installed on the computer device 800 via ROM 802 and / or communication unit 809. When the computer program is loaded into RAM 803 and executed by the computing unit 801, one or more steps of the phase current limiting method for a permanent magnet synchronous motor described above can be performed. Alternatively, in other embodiments, the computing unit 801 may be configured to perform a phase current limiting method for a permanent magnet synchronous motor by any other suitable means (e.g., by means of firmware).

[0143] Various embodiments of the systems and techniques described above herein can be implemented in digital electronic circuit systems, integrated circuit systems, field-programmable gate arrays (FPGAs), application-specific integrated circuits (ASICs), application-specific standard products (ASSPs), systems-on-a-chip (SoCs), payload-programmable logic devices (CPLDs), computer hardware, firmware, software, and / or combinations thereof. These various embodiments may include implementations in one or more computer programs that can be executed and / or interpreted on a programmable system including at least one programmable processor, which may be a dedicated or general-purpose programmable processor, capable of receiving data and instructions from a storage system, at least one input device, and at least one output device, and transmitting data and instructions to the storage system, the at least one input device, and the at least one output device.

[0144] The program code used to implement the methods of the present invention can be written in any combination of one or more programming languages. This program code can be provided to a processor or controller of a general-purpose computer, special-purpose computer, or other programmable data processing device, such that when executed by the processor or controller, the program code causes the functions / operations specified in the flowcharts and / or block diagrams to be implemented. The program code can be executed entirely on the machine, partially on the machine, as a standalone software package partially on the machine and partially on a remote machine, or entirely on a remote machine or server.

[0145] In the context of this invention, a machine-readable medium can be a tangible medium that may contain or store a program for use by or in conjunction with an instruction execution system, apparatus, or device. A machine-readable medium can be a machine-readable signal medium or a machine-readable storage medium. Machine-readable media can include, but are not limited to, electronic, magnetic, optical, electromagnetic, infrared, or semiconductor systems, apparatus, or devices, or any suitable combination of the foregoing. More specific examples of machine-readable storage media include electrical connections based on one or more wires, portable computer disks, hard disks, random access memory (RAM), read-only memory (ROM), erasable programmable read-only memory (EPROM or flash memory), optical fibers, portable compact disk read-only memory (CD-ROM), optical storage devices, magnetic storage devices, or any suitable combination of the foregoing.

[0146] To provide interaction with a user, the systems and techniques described herein can be implemented on a computer having: a display device for displaying information to the user (e.g., a CRT (cathode ray tube) or LCD (liquid crystal display) monitor); and a keyboard and pointing device (e.g., a mouse or trackball) through which the user provides input to the computer. Other types of devices can also be used to provide interaction with the user; for example, feedback provided to the user can be any form of sensory feedback (e.g., visual feedback, auditory feedback, or tactile feedback); and input from the user can be received in any form (including sound input, voice input, or tactile input).

[0147] The systems and technologies described herein can be implemented in computing systems that include backend components (e.g., as a data server), or computing systems that include middleware components (e.g., an application server), or computing systems that include frontend components (e.g., a user computer with a graphical user interface or web browser through which a user can interact with implementations of the systems and technologies described herein), or any combination of such backend, middleware, or frontend components. The components of the system can be interconnected via digital data communication of any form or medium (e.g., a communication network). Examples of communication networks include local area networks (LANs), wide area networks (WANs), and the Internet.

[0148] Computer systems can include clients and servers. Clients and servers are generally located far apart and typically interact via communication networks. Client-server relationships are created by computer programs running on the respective computers and having a client-server relationship with each other. Servers can be cloud servers, servers in distributed systems, or servers incorporating blockchain technology.

[0149] It should be understood that the various forms of processes shown above can be used to reorder, add, or delete steps. For example, the steps described in this invention can be executed in parallel, sequentially, or in different orders, as long as the desired result of the technical solution disclosed in this invention can be achieved, and this is not limited herein.

[0150] The specific embodiments described above do not constitute a limitation on the scope of protection of this invention. Those skilled in the art should understand that various modifications, combinations, sub-combinations, and substitutions can be made according to design requirements and other factors. Any modifications, equivalent substitutions, and improvements made within the principles of this invention should be included within the scope of protection of this invention.

Claims

1. A method for limiting phase current in a permanent magnet synchronous motor, characterized in that, include: The operating parameters and current commutation status of the permanent magnet synchronous motor are obtained within the current control cycle. The operating parameters include bus voltage, bus current, rotor electrical angle, and rotor electrical angular velocity. Based on the three-phase voltage equation and three-phase flux linkage equation of the permanent magnet synchronous motor, a current prediction model corresponding to each commutation state is constructed. Based on the current commutation state, a target current prediction model is determined from each of the current prediction models. The target current prediction model includes a resistance voltage drop term, a back EMF term, an inductance conversion term, a voltage margin term, and a discrete prediction term. The operating parameters and the current commutation state are input into the target current prediction model to predict the predicted current for the next control cycle. If the predicted current is greater than the preset current limiting threshold, the target current prediction model is solved in reverse based on the current limiting threshold and the operating parameters to obtain the target duty cycle; The control inverter bridge generates a pulse width modulation signal based on the target duty cycle and the current commutation state to drive the permanent magnet synchronous motor to operate, thereby limiting the phase current of the permanent magnet synchronous motor; Based on the back EMF phase identifier corresponding to the current commutation state, the rotor electrical angle is phase matched, and the back EMF term is used to perform trigonometric function mapping on the phase-matched rotor electrical angle to obtain the back EMF characterization quantity corresponding to the current commutation state. The rotor electric angular velocity and permanent magnet flux linkage are input into the back EMF term to obtain the back EMF amplitude. The back EMF amplitude and the back EMF characterization quantity are summed to obtain the back EMF influence quantity corresponding to the conducting phase circuit. The voltage drop of the bus current is calculated by converting the voltage drop of the resistance voltage drop term according to the stator resistance corresponding to the conduction phase identifier, and the resistance voltage drop corresponding to the conduction phase circuit is obtained. The conduction phase identifier corresponds to the current commutation state. The bus voltage, the back EMF influence, and the resistor voltage drop are input into the voltage margin term to obtain the effective voltage of the conducting phase circuit in the current control cycle. The effective voltage quantity is converted into the current change rate by the inductance conversion term to obtain the current change rate of the conducting phase circuit in the current control cycle. The current change rate and the duration of the current control cycle are input into the discrete prediction term to perform discrete integration on the current change rate. The result of the discrete integration is then superimposed on the bus current to obtain the predicted current for the next control cycle.

2. The method according to claim 1, characterized in that, The current prediction model for each commutation state, constructed based on the three-phase voltage equation and three-phase flux linkage equation of the permanent magnet synchronous motor, includes: The three-phase flux linkage equation is simplified based on the stator winding self-inductance to obtain the simplified flux linkage relationship. Substituting the simplified magnetic flux relationship into the three-phase voltage equation, we obtain the dynamic relationship of the three-phase current; Obtain the conducting phase combination and the suspended phase corresponding to each of the six-step commutation control states; Based on the dynamic relationship of the three-phase current and the combination of conducting phases and the floating phases corresponding to each commutation state, a current prediction model corresponding to each commutation state is constructed.

3. The method according to claim 1, characterized in that, The step of performing phase matching on the rotor electrical angle based on the back EMF phase identifier corresponding to the current commutation state, and then performing trigonometric function mapping on the phase-matched rotor electrical angle through the back EMF term to obtain the back EMF characterization quantity corresponding to the current commutation state includes: Based on the conduction phase identifier corresponding to the current commutation state, determine the positive conduction phase connected to the positive end of the bus and the negative conduction phase connected to the negative end of the bus in the current control cycle; Based on the back EMF phase identifier corresponding to the current commutation state, the rotor electrical angle is phase-shifted on the positive conduction phase and the negative conduction phase respectively to obtain the first back EMF angle corresponding to the positive conduction and the second back EMF angle corresponding to the negative conduction. The first back EMF angle and the second back EMF angle are converted into sinusoidal values ​​to obtain the first back EMF component and the second back EMF component. The first back EMF component and the second back EMF component are summed to obtain the back EMF characterization quantity corresponding to the current commutation state.

4. The method according to claim 1, characterized in that, The process of inversely solving the target current prediction model based on the current limiting threshold and the operating parameters to obtain the target duty cycle includes: The current limiting threshold and the operating parameters are used as the predicted current input to the target current prediction model, and the target current prediction model is algebraically transformed so that the target duty cycle is used as the variable to be determined. Based on the operational relationships between the resistance voltage drop term, back EMF term, inductance conversion term, voltage margin term and discrete prediction term in the target current prediction model, the variable to be determined is solved in reverse to obtain the current limiting duty cycle. The speed adjustment duty cycle is calculated based on the target speed and the actual speed of the permanent magnet synchronous motor. The minimum value between the current limiting duty cycle and the speed adjustment duty cycle is taken as the target duty cycle.

5. The method according to claim 4, characterized in that, The step of calculating the speed adjustment duty cycle based on the target speed and the actual speed of the permanent magnet synchronous motor includes: In response to a target speed command, the target speed command carries the target speed; The acquired speed detection signal is sampled and filtered to obtain the actual speed of the permanent magnet synchronous motor. Based on the difference between the target speed and the actual speed, the speed deviation is determined; The speed deviation is input into the speed regulator, which performs proportional-integral adjustment on the speed deviation to obtain the speed loop output. The speed loop output is subjected to duty cycle mapping and limiting processing to obtain the speed regulation duty cycle.

6. The method according to claim 1, characterized in that, The control inverter bridge drives the permanent magnet synchronous motor to operate based on the target duty cycle and the current commutation state, including: Generate a pulse width modulation signal corresponding to the current commutation state based on the target duty cycle; The pulse width modulation signal is output to the power switching devices in the inverter bridge corresponding to the positive conduction phase and the negative conduction phase, so as to control the positive conduction phase to be connected to the positive terminal of the bus, the negative conduction phase to be connected to the negative terminal of the bus, and keep the corresponding bridge arm of the floating phase off. A modulation voltage corresponding to the target duty cycle is applied to the permanent magnet synchronous motor through the conduction loop between the positive conduction phase and the negative conduction phase to drive the permanent magnet synchronous motor to run.

7. A phase current limiting system for a permanent magnet synchronous motor, characterized in that, include: The acquisition module is used to acquire the operating parameters and current commutation status of the permanent magnet synchronous motor in the current control cycle. The operating parameters include bus voltage, bus current, rotor electrical angle, and rotor electrical angular velocity. The current prediction model construction module is used to construct current prediction models corresponding to each commutation state based on the three-phase voltage equation and the three-phase flux linkage equation of the permanent magnet synchronous motor, and to determine the target current prediction model from each of the current prediction models based on the current commutation state. The target current prediction model includes a resistance voltage drop term, a back EMF term, an inductance conversion term, a voltage margin term, and a discrete prediction term. The prediction module is used to input the operating parameters and the current commutation state into the target current prediction model to predict the predicted current for the next control cycle. The reverse solution module is used to reverse solve the target current prediction model based on the current limiting threshold and the operating parameters if the predicted current is greater than the preset current limiting threshold, so as to obtain the target duty cycle. The phase current limiting module is used to control the inverter bridge to drive the permanent magnet synchronous motor to operate based on the target duty cycle and the current commutation state, so as to limit the phase current of the permanent magnet synchronous motor. The prediction module includes: The matching and mapping unit is used to perform phase matching on the rotor electrical angle according to the back EMF phase identifier corresponding to the current commutation state, and to perform trigonometric function mapping on the phase-matched rotor electrical angle through the back EMF term to obtain the back EMF characterization quantity corresponding to the current commutation state. The back EMF term unit is used to input the rotor electric angular velocity and permanent magnet flux linkage into the back EMF term to obtain the back EMF amplitude, and to sum the back EMF amplitude and the back EMF characterization quantity to obtain the back EMF influence quantity corresponding to the conducting phase circuit. The voltage drop conversion unit is used to perform voltage drop conversion on the bus current according to the stator resistance corresponding to the conduction phase identifier through the resistance voltage drop item, so as to obtain the resistance voltage drop corresponding to the conduction phase circuit. The conduction phase identifier corresponds to the current commutation state. The voltage margin term unit is used to input the bus voltage, the back EMF influence amount and the resistor voltage drop amount into the voltage margin term to obtain the effective voltage amount of the conducting phase circuit in the current control cycle; The inductance conversion unit is used to convert the effective voltage quantity into the current change rate through the inductance conversion item, so as to obtain the current change rate of the conducting phase circuit in the current control cycle. The discrete prediction term unit is used to input the current change rate and the duration of the current control cycle into the discrete prediction term, to perform discrete integration processing on the current change rate, and to superimpose the discrete integration processing result with the bus current to obtain the predicted current for the next control cycle.

8. A computer device, characterized in that, include: At least one processor; and a memory communicatively connected to the at least one processor; The memory stores instructions that can be executed by the at least one processor to enable the at least one processor to perform the method of any one of claims 1-6.

9. A non-transitory computer-readable storage medium storing computer instructions, characterized in that, The computer instructions are used to cause the computer to perform the method according to any one of claims 1-6.

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