Inverter control device and semiconductor module

Abstract

Inverter control device (1), comprising: a main control circuit (11) which applies a control voltage to a semiconductor switching element (SW) provided in an inverter output stage and controls a current delivery to a load, thereby switching the semiconductor switching element (SW) on and off, wherein the semiconductor switching element (SW) is switched off when a control unit of the inverter control device detects an overcurrent; a clamping diode (ZD) that clamps a voltage of an electromotive counterforce applied to the semiconductor switching element (SW) when the operation of the main control circuit (11) stops; a voltage divider resistor (RA, RB) that resistively divides and detects a voltage proportional to a current flowing through the clamping diode (ZD) when the electromotive force voltage is clamped; and an auxiliary control circuit (12) which generates a control voltage according to the voltage detected by the voltage divider resistor (RA, RB), wherein the control voltage switches on the semiconductor switching element (SW) in an abnormal condition of the inverter, wherein when the auxiliary control circuit (12) generates the control voltage which switches on the semiconductor switching element (SW), a control voltage is applied to the semiconductor switching element (SW) as a drive voltage, wherein the drive voltage is divided via an internal impedance (RG) of the main control circuit (11) and via an output resistance (R1) of the auxiliary control circuit (12).

Description

GENERAL STATE OF THE ART1. Field of the invention

[0001] The present invention relates to an inverter control device that switches a semiconductor switching element provided in an inverter output stage on and off and controls the current delivery to a consumer, and relates to a semiconductor module in which the inverter control device and the semiconductor switching element are integrally included. 2. Description of the state of the art

[0002] An inverter 10, which drives a single-phase motor, three-phase motor or the like, includes in an output stage a semiconductor switching element SW that controls the current output to a load, and is configured to include an inverter control device 1 that switches the semiconductor switching element SW on and off. Fig. Figure 3 is a diagram showing a schematic configuration of the inverter 10, which drives a three-phase motor M acting as a load. The inverter 10 comprises the semiconductor switching element SW, comprising upper arm IGBTs 2u, 2v, and 2w and lower arm IGBTs 3u, 3v, and 3w, which are connected and driven in a complementary manner as totem poles in a U-phase, V-phase, and W-phase, respectively. Freewheeling diodes 4u, 4v, 4w, 5u, 5v, and 5w are connected antiparallel between an emitter and a collector of the upper arm IGBTs 2u, 2v, and 2w and the lower arm IGBTs 3u, 3v, and 3w, respectively.

[0003] Here, the totem-pole connection of the upper arm IGBTs 2u, 2v, and 2w and the lower arm IGBTs 3u, 3v, and 3w denotes a circuit configuration in which the emitters of the upper arm IGBTs 2u, 2v, and 2w are connected to the collectors of the lower arm IGBTs 3u, 3v, and 3w, respectively. Each series connection of the upper arm IGBTs 2u, 2v, and 2w and the lower arm IGBTs 3u, 3v, and 3w connected as totem poles forms a half-bridge circuit.

[0004] Furthermore, the inverter control device includes 1 upper arm control circuits (HVICs) 6u, 6v and 6w, which switch the upper arm IGBTs 2u, 2v and 2w respectively on and off, and a lower arm control circuit (LVIC) 7, which switches each of the lower arm IGBTs 3u, 3v and 3w on and off. The control circuits of the upper arm 6u, 6v and 6w and the control circuit of the lower arm 7 receive control signals, in particular U-phases, V-phases and W-phase PWM signals, which are individually provided by a control device CONT, which is formed, for example, from a PWM control microcomputer and thereby switches the IGBTs of the upper arm 2u, 2v and 2w and the IGBTs of the lower arm 3u, 3v and 3w on and off with predetermined phase differences.

[0005] Furthermore, a current-sensing resistor RS is interposed in a power supply path of the semiconductor switching element SW, which is formed from the upper arm IGBTs 2u, 2v and 2w and the lower arm IGBTs 3u, 3v and 3w connected as totem poles. The current-sensing resistor RS detects a voltage proportional to the current flowing in the inverter 10 as current information, and the detected current information is fed into the control device CONT and the control circuit of the lower arm 7, respectively.

[0006] For example, an irregularity, such as a short circuit between phases caused by an insulation fault or incorrect wiring in the output wiring of inverter 10, is detected from the current information. Specifically, the lower arm 7 drive circuit includes an overcurrent protection function that immediately and directly switches off the lower arm IGBTs 3u, 3v, and 3w when an excessive current is detected, thereby interrupting the current flowing in these IGBTs. Furthermore, the control device CONT includes an overcurrent function that outputs control current information to the lower arm drive circuits 6u, 6v, and 6w when an excessive current is detected, thereby controlling the upper arm IGBTs 2u, 2v, and 2w, respectively, to switch them off.

[0007] This document provides a simple description of the control circuits of the upper arm 6u, 6v and 6w and the control circuit of the lower arm 7 in the inverter device 1. Fig. Figure 4 is a diagram of a schematic configuration, where the inverter control device 1 of a single phase, in this case the U-phase, of inverter 10 is extracted. The inverter control devices 1 of the V-phase and W-phase are configured in the same way.

[0008] The control circuit of the upper arm 6 (6u, 6v, 6w) comprises a P-MOS 8a and an N-MOS 8b, connected in series and switched on and off in a complementary manner, as output stage transistors that switch the IGBT of the upper arm 2u (2v, 2w) on and off. Here, the output stage transistors formed from the P-MOS 8a and N-MOS 8b switch the IGBT of the upper arm 2u (2v, 2w) on and off by switching on and off in a complementary manner, using a voltage at a junction of the IGBT of the upper arm 2u (2v, 2w) and the IGBT of the lower arm 3u (3v, 3w), which are connected as totem poles; that is, an intermediate voltage Vs serves as a reference potential.

[0009] Furthermore, the control circuit of the upper arm 6 (6u, 6v, 6w) comprises an input filter 8c, which receives a control signal (PWM signal) provided by the control device CONT, and a level-shifting circuit 8d, which shifts the level of the control signal (PWM signal) received via the input filter 8c to the level of the operating reference potential of the output stage transistor. The control circuit of the upper arm 6 (6u, 6v, 6w) is also configured to switch the output stage transistor (P-MOS 8a, N-MOS 8b) on and off using the control signal (PWM signal) whose level has been shifted by the level-shifting circuit 8d.

[0010] When the control device CONT detects an overcurrent from the current information (sensed voltage) measured via the current sensing resistor RS, it stops the output of the control signal (PWM signal). As a result, the P-MOS 8a and the N-MOS 8b stop, and the lower arm IGBT 2u (2V, 2W) is controlled to forcibly switch off.

[0011] Meanwhile, the U-phase (V-phase, W-phase) of the lower arm 7 drive circuit comprises a P-MOS 9a and an N-MOS 9b connected in series and switched on and off in a complementary manner, as output stage transistors that switch the lower arm 3u (3v, 3w) IGBT on and off. The output stage transistors formed from the P-MOS 9a and N-MOS 9b control the lower arm 3u (3v, 3w) IGBT by switching on and off in a complementary manner, with ground potential (GND) as a reference potential.

[0012] Furthermore, the control circuit of the lower arm 7 comprises an input filter 9c, which receives a control signal (PWM signal) provided by the control device CONT, and an AND gate circuit 9d, which controls the output of the control signal (PWM) received via the input filter to the output stage transistor (P-MOS 9a, N-MOS 9b). The AND gate circuit 9d performs the function of switching the P-MOS 9a and N-MOS 9b on and off in a complementary manner by outputting the control signal (PWM signal) to the output stage transistor (P-MOS 9a, N-MOS 9b) only when an output of a latching circuit 9e is "high".

[0013] Here, the current information (detected voltage) detected via the current sensing resistor RS is provided to a comparator 9f and compared with a predetermined reference voltage 9g. If the detected voltage exceeds the reference voltage 9g, the comparator 9f interprets this as an overcurrent and sets the output of the latching circuit 9e to "L". By setting the output of the latching circuit 9e to "L" due to the overcurrent detection, the AND gate circuit 9d is closed, and the control signal (PWM signal) of the output stage transistor (P-MOS 9a, N-MOS 9b) is forcibly prevented. This controls the lower arm IGBT 3u (3V, 3W) so that it is forcibly switched off when an overcurrent is detected.

[0014] Herein lies the inverter 10, which is configured to include the inverter control device 1, configured as previously described, such that if, for example, a short circuit occurs between phases in the output wiring and an overcurrent (short-circuit current) flows, the control circuit of the lower arm 7 immediately detects the occurrence of the overcurrent and controls the IGBTs of the lower arm 3u, 3v, and 3w to switch them off. Meanwhile, the control device CONT detects an occurrence of an overcurrent (short-circuit current) and stops the main current of the control signal (PWM), which is why a slight delay cannot be denied in the control circuits of the upper arm 6u, 6v, and 6w, which control the IGBTs of the upper arm 2u, 2v, and 2w to switch them off.

[0015] If the upper arm IGBTs 2u, 2v, and 2w and the lower arm IGBTs 3u, 3v, and 3w are forcibly switched off, a reverse current flows into the upper arm IGBTs 2u, 2v, and 2w due to an inductance component present in the internal wiring of the upper arm 6 (6u, 6v, 6w) drive circuit. Since the current that is forcibly switched off immediately before the upper arm IGBTs 2u, 2v, and 2w and the lower arm IGBTs 3u, 3v, and 3w is an overcurrent (short-circuit current), the reverse current flowing at that moment is ten times or more greater than a reverse current flowing when the inverter is operating normally.

[0016] Therefore, the magnitude of the current change (-dlc / dt) when the upper arm IGBTs 2u, 2v, and 2w are forcibly switched off is 1000 A / µs or more, which is ten times or more greater than the magnitude of the current change (-dlc / dt) during normal operation. As a result, an electromotive force caused by the inductance component present in the internal wiring and the magnitude of the current change (-Idc / dt) remains unchanged on the upper arm IGBTs 2u, 2v, and 2w. Furthermore, if the electromotive force exceeds a breakdown voltage between the collectors and emitters of the upper arm IGBTs 2u, 2v and 2w and a breakdown voltage between the cathodes and anodes of the freewheeling diodes 4u, 4v and 4w, then there is concern that the upper arm IGBTs 2u, 2v and 2w will experience an overvoltage failure.

[0017] To combat this type of problem, it is considered to increase the breakdown voltage between the collectors and emitters of the upper arm IGBTs 2u, 2v, and 2w, and the breakdown voltage between the cathodes and anodes of the freewheeling diodes 4u, 4v, and 4w, relative to the electromotive force generated by the magnitude of the current change (-dlc / dt) when the current is interrupted. However, since the breakdown voltage between the collectors and emitters of the upper arm IGBTs 2u, 2v, and 2w and their conduction losses are in a compensating relationship, a new problem arises: the loss in the upper arm IGBTs 2u, 2v, and 2w increases when the upper arm 6 drive circuit (6u, 6v, 6w) operates normally, and the operating performance of the inverter 10 deteriorates.

[0018] In this regard, a case in which a clamping diode (Zener diode) ZD, which counteracts the electromotive force, and a current-return blocking diode (blocking diode) D are connected in series between the collector and a gate of the upper arm IGBT 2u (2v, 2w), as shown by the broken line in Fig. 4 shown, and the voltage of the electromotive force applied to the upper arm IGBT 2u (2v, 2w) is clamped by the clamping diode ZD, for example as disclosed in JP-A-2009-253484.

[0019] According to the inverter 10, which is configured to include this type of clamping diode ZD and blocking diode D, the energy of the electromotive force applied to the IGBT of the upper arm 2u (2v, 2w) can be caused to flow from the gate side of the IGBT of the upper arm 2u (2v, 2w) as a breakdown current Ir of the clamping diode ZD into the drive circuit of the upper arm 6 (6u, 6v, 6w). Therefore, a voltage of equivalent internal impedance of the drive circuit of the upper arm 6 (6u, 6v, 6w) is generated by the breakdown current Ir flowing through the clamping diode ZD into the drive circuit of the upper arm 6 (6u, 6v, 6w), and this voltage is applied to the gate of the IGBT of the upper arm 2u (2v, 2w).

[0020] Therefore, the internal impedance (an equivalent gate resistance RG) of the upper arm 6 (6u, 6v, 6w) drive circuit, as seen from the upper arm 2u (2v, 2w) IGBT, is set such that, for example, a voltage applied to the gate of the upper arm 2u (2v, 2w) IGBT exceeds an operating limit of the upper arm 2u (2v, 2w) IGBT, and a collector current flows due to saturated operation of the upper arm 2u (2v, 2w) IGBT. Thus, the upper arm 2u (2v, 2w) IGBT is switched on in a saturated operating state, which is why the energy of the electromotive force applied to the upper arm 2u (2v, 2w) IGBT flows through the upper arm 2u (2v, 2w) IGBT.

[0021] This allows the energy of the electromotive force to be dissipated as heat energy by the IGBT of the upper arm 2u (2v, 2w). Consequently, the voltage applied to the IGBT of the upper arm 2u (2v, 2w) due to the electromotive force can be limited by the clamping diode ZD, effectively preventing overvoltage breakdown of the IGBT of the upper arm 2u (2v, 2w).

[0022] It is noted that a gate voltage of approximately 6V is required to switch on the IGBT of the upper arm 2u (2v, 2w). Furthermore, the internal impedance (equivalent gate resistance RG) of the upper arm 6 (6u, 6v, 6w) driver circuit is generally in the range of 10 to 50Ω. Therefore, to obtain a gate voltage of approximately 6V, the breakdown current Ir flowing through the clamping diode ZD must be a maximum of 600 mA. Consequently, to reduce the clamping resistance of the clamping diode ZD, it is necessary to use a chip area of ​​the same size as that of the IGBT of the upper arm 2u (2v, 2w) for the clamping diode ZD. This presents problems due to the increased circuit area, higher system costs, and other similar issues.

[0023] However, if a clamping diode ZD with a small chip area, which can possibly be embedded in the chip of the upper arm 6 (6u, 6v, 6w) driver circuit, is used, the breakdown current Ir flowing into the clamping diode decreases to, for example, the range of 100µA. Therefore, in order to generate a gate voltage of approximately 6V when the upper arm 2u (2v, 2w) IGBT is switched on in a saturated operating state, it is necessary that the internal impedance (equivalent gate resistance RG) of the upper arm 6 (6v, 6w) driver circuit be in the range of, for example, 60kΩ.

[0024] If the internal impedance of the upper arm 6 IGBT (6u, 6v, 6w) is increased in this way, the switching loss in the upper arm 2u IGBT (2v, 2w) typically increases when the inverter 10 is operating normally. Furthermore, the amount of heat generated when the upper arm 2u IGBT (2v, 2w) switches on increases, and switching operations at or above 10 kHz, which is a common switching frequency in the inverter 10, become difficult.

[0025] US 2013 / 0021083 A1 discloses a clamping circuit with a first switching element, a first diode, a first resistor, a first control circuit, and a second control circuit. The first diode is connected to the first switching element and breaks down when an overvoltage is applied across it. The first resistor is connected to the first diode and detects a current flowing through it. The first control circuit is configured to amplify a voltage drop across the first resistor and control the current through the first switching element. The second control circuit is configured to control the conductivity of the first switching element according to the voltage drop across the first resistor.

[0026] DE 43 00 100 A1 discloses a semiconductor device comprising a switching device such as a MOSFET or an IGBT, and an avalanche device for protecting the switching device by generating an avalanche breakdown current when an overvoltage is applied to the switching device. The avalanche device shares a drift layer, i.e., an epitaxial layer, with the switching device. In this arrangement, the avalanche breakdown voltage of the avalanche device follows changes in the breakdown voltages of the switching device that are due to variations in the thickness or impurity concentration of the epitaxial layer or the temperature. This allows the safety margin between the avalanche breakdown voltage of the avalanche device and the breakdown voltage of the switching device to be reduced while still reliably protecting the switching device from damage. BRIEF SUMMARY OF THE INVENTION

[0027] The invention, which was developed taking into account these types of situations, has an objective of providing an inverter control device such that, while limiting an undesired loss when a semiconductor switching element provided in an inverter output stage is operating normally, an overvoltage breakdown of the semiconductor switching element due to an electromotive force applied to the semiconductor switching element at the time of an abnormal condition can be reliably prevented.

[0028] At the same time, the invention has an objective of providing a semiconductor module configured to integrally include the semiconductor switching element provided in the inverter output stage and an inverter control device that switches the semiconductor switching element on and off.

[0029] To solve the problems described above, an inverter control device according to claim 1 is provided according to a first aspect of the invention. A second aspect of the invention relates to a semiconductor module according to claim 9. Further aspects of the invention are the subject of the dependent claims, the drawings, and the description of exemplary embodiments.

[0030] In this circuit, the semiconductor switching element is, for example, an IGBT, and the main driver circuit applies the drive voltage to a gate of the IGBT, thereby switching the IGBT on and off. Additionally, the auxiliary driver circuit applies the control voltage to the gate of the IGBT, thereby switching the IGBT on in a saturated operating region. In this circuit, the clamping diode is a Zener diode, which has a cathode-anode breakdown voltage that is lower than the collector-emitter breakdown voltage of the IGBT.

[0031] Specifically, the main drive circuit applies the drive voltage to the gate of the IGBT via a gate resistor when the inverter is operating normally, and the auxiliary drive circuit applies the control voltage to the gate of the IGBT via an output resistor when the inverter is in an abnormal state. Furthermore, the auxiliary drive circuit is connected in parallel to the main drive circuit.

[0032] Preferably, the semiconductor switching element is, for example, an upper-arm IGBT and a lower-arm IGBT connected as totem poles and driven alternately. The main drive circuit consists of an upper-arm drive circuit that switches the upper-arm IGBT on and off, and a lower-arm drive circuit that switches the lower-arm IGBT on and off. Furthermore, the auxiliary drive circuit is provided in the upper-arm drive circuit and performs the role of protecting the upper-arm IGBT from the electromotive force applied to it when the upper-arm IGBT is switched off and the lower-arm IGBT is in an off state.

[0033] Furthermore, a semiconductor module according to the invention is characterized in that it integrally comprises a semiconductor switching element, which is provided in an inverter output stage and controls the current output to a load, and the inverter control device of the previously described configuration, which switches the semiconductor switching element on and off. Alternatively, a semiconductor module according to the invention is characterized in that two phases or three phases of the semiconductor switching element and the inverter control device of the previously described configuration, which switches the semiconductor switching element on and off, are integrally provided in parallel.

[0034] In this system, a multiple of the inverter control device, which is provided in parallel for two or three phases, controls a multiple of the semiconductor switching element, which is provided in parallel with a predetermined phase difference, on and off.

[0035] According to the inverter control device and the semiconductor module of the previously described configurations, the semiconductor switching element (for example, an IGBT) can be switched on in a saturated operating range by the auxiliary control circuit even when an abnormal return current flows into the semiconductor element. This ensures that the semiconductor switching element (IGBT) is forcibly switched off when the inverter is in an abnormal state. Furthermore, the energy of an electromotive force applied to the semiconductor switching element due to the abnormal return current can be effectively dissipated by the semiconductor switching element. As a result, overvoltage breakdown of the semiconductor switching element due to the electromotive force caused by the abnormal return current can be effectively prevented.

[0036] Furthermore, since the auxiliary drive circuit does not drive the semiconductor switching element instead of the main drive circuit when the inverter is operating normally, the presence of the auxiliary drive circuit is not a factor in increasing losses in the semiconductor switching element. Consequently, a large number of practical advantages are gained, such as the ability to reliably prevent overvoltage breakdown of the semiconductor switching element in the event of an abnormal condition, such as a short circuit of the inverter output, while limiting losses in the semiconductor switching element during normal operation and maintaining the efficiency of the inverter. BRIEF DESCRIPTION OF THE DRAWINGS Fig. Figure 1 is a diagram showing a schematic configuration of a main part of an inverter control device according to an embodiment of the invention; Fig. Figure 2 is a diagram showing a schematic configuration of a semiconductor module according to an embodiment of the invention, in which a control inverter of a three-phase motor is constructed; Fig. Figure 3 is a configuration diagram showing an example of a control inverter for a three-phase motor; and Fig. Figure 4 is a diagram showing a schematic configuration of an existing typical inverter control device. DETAILED DESCRIPTION OF THE INVENTION

[0037] The following is a description, with reference to the drawings, of an inverter control device according to an embodiment of the invention, wherein a U-phase inverter control device in a control inverter of a three-phase motor serves as an example. The same reference numerals are assigned to parts that are identical to those in a version shown in the Fig. 3 and Fig. The devices shown in Figure 4 already exist, and a description of them is omitted. Furthermore, V-phase and W-phase inverter control devices in the control inverter of a three-phase motor are configured in the same way as the U-phase inverter control device described here.

[0038] An inverter control device 1 according to an embodiment of the invention, as shown in a schematic configuration thereof in Fig. Figure 1 shows a main control circuit 11 which applies a control voltage to a semiconductor switching element SW which is provided in an output stage of an inverter 10 and controls the current delivery to a consumer, thereby switching the semiconductor switching element SW on and off.

[0039] The semiconductor switching element SW consists of an upper-arm IGBT 2u (2v, 2w) and a lower-arm IGBT 3u (3v, 3w), which are connected as totem poles and driven in a complementary manner. Freewheeling diodes 4u, 4v, 4w, 5u, 5v and 5w are connected antiparallel between the emitter and collector of the upper-arm IGBT 2u (2v, 2w) and the lower-arm IGBT 3u (3v, 3w), respectively. Furthermore, the main control circuit 11 is formed from an upper arm control circuit 6u (6v, 6w), which switches the upper arm IGBT 2u (2v, 2w) on and off, and a lower arm control circuit 7, which switches the lower arm IGBT 3u (3v, 3w) on and off. Only a U-phase control circuit is extracted from the lower arm control circuit 7, which includes the U-phase, V-phase, and W-phase control circuits, and is shown in Fig. 1 shown.

[0040] The inverter control device 1 according to the embodiment of the invention is characterized by comprising a clamping diode ZD, which clamps the voltage of an electromotive force applied to the semiconductor switching element SW, in particular the IGBT of the upper arm 2u (2v, 2w), when the operation of the main control circuit 11 stops, and by voltage divider resistors RA and RB, which resistively divide and detect a voltage proportional to a current flowing through the clamping diode when the voltage of the electromotive force is clamped by the clamping diode ZD. A blocking diode D, which blocks the reverse current, is connected in series with the clamping diode ZD.

[0041] In particular, the clamping diode ZD is configured such that one of its cathodes is connected to a collector of the IGBT of the upper arm 2u (2v, 2w), and one of its anodes is connected to the anode of the clamping diode ZD. Furthermore, one of the cathodes of the clamping diode is connected, via the series-connected voltage divider resistors RA and RB, to a power supply line carrying an intermediate voltage Vs, which regulates a reference potential for the drive circuit of the upper arm 6u (6v, 6w). Consequently, a current generated by the clamping diode ZD, which clamps the voltage of the electromotive force, flows through the clamping diode D into the voltage divider resistors RA and RB, and the voltage divider resistors RA and RB divide and detect a voltage proportional to this current.

[0042] Furthermore, the inverter control device 1 is characterized by including an auxiliary control circuit 12, which generates a control voltage according to the voltage detected by the voltage divider resistors RA and RB and applies the control voltage to the IGBT of the upper arm 2u (2v, 2w) in the control circuit of the upper arm 6u (6v, 6w) instead of the main control circuit 11. The auxiliary control circuit 12 performs the role of switching on the IGBT of the upper arm 2u (2v, 2w) in a saturated operating range, for example, when the IGBT of the lower arm 3u (3v, 3w) is controlled such that it is forcibly switched off due to the generation of a short-circuit current, whereby the electromotive force is simultaneously applied to the IGBT of the upper arm 2u (2v, 2w).

[0043] That is, the switching control of the upper arm 6u (6v, 6w) is configured to include in parallel the main control circuit 11, which switches the IGBT of the upper arm 2u (2v, 2w) on and off at a time of normal operation, and the auxiliary control device 12, which switches the IGBT of the upper arm 2u (2v, 2w) on in a saturated range when the electromotive counterforce is applied to the IGBT of the upper arm 2u (2v, 2w) in an abnormal state.

[0044] Here, the auxiliary drive circuit 12 is configured to include inverting amplifiers 13a and 13b connected in two stages. The inverting amplifiers 13a and 13b are each configured, for example, as a P-MOS and an N-MOS connected in series. The inverting amplifier of the first stage, 13a, amplifies inversely a voltage generated across the voltage divider resistor RB by a current flowing through the clamping diode ZD into the voltage divider resistors RA and RB. Furthermore, the inverting amplifier of the second stage, 13b, inversely amplifies an output voltage of the inverting amplifier of the first stage, 13a, and generates a control voltage to switch on the upper arm IGBT 2u (2V, 2W) in a saturated operating region. The control voltage is generated, for example, as a voltage approximately equal to a supply voltage VB applied to the auxiliary drive circuit 12.

[0045] The control voltage output in this way by the auxiliary control circuit 12 is divided by an internal impedance of the control circuit of the upper arm 6u (6v, 6w), in particular an equivalent internal impedance (gate resistance) RG of the main control circuit 11 and an output resistance R1 of the auxiliary control circuit 12, and applied to a gate of the IGBT of the upper arm 2u (2v, 2w). The IGBT of the upper arm 2u (2v, 2w) is switched on in a saturated operating region by the control voltage applied in this way, and the energy of the electromotive force applied to the IGBT of the upper arm 2u (2v, 2w) flows through the IGBT of the upper arm 2u (2v, 2w). As a result, the energy of the electromotive force is consumed as heat energy in the IGBT of the upper arm 2u (2v, 2w), which is why an overvoltage breakdown of the IGBT of the upper arm 2u (2v, 2w) can be prevented.

[0046] A diode positioned between the output resistor R1 and an output end of the main driver circuit 11 serves to prevent the output voltage (control voltage) of the auxiliary driver circuit 12 from being applied to the output end of the main driver circuit 11 during normal operation. Thanks to this diode, the control voltage of the main driver circuit 11 is applied to the IGBT of the upper arm 2u (2V, 2W) during normal operation without being affected by the output voltage of the auxiliary driver circuit 12.

[0047] According to the upper arm control circuit 6u (6v, 6w), in which the auxiliary control circuit 12, which outputs a control voltage according to the current flowing through the clamping diode ZD, is connected in parallel with the main control circuit 11 as previously described, the energy of the electromotive force applied to the upper arm IGBT 2u (2v, 2w) can be effectively dissipated as heat energy by the upper arm IGBT 2u (2v, 2w), which is switched on in a saturated operating range. Therefore, even if the operation of the main control circuit 11 stops due to an abnormal situation that results in an electromotive force being applied to the upper arm IGBT 2u (2v, 2w), an overvoltage breakdown of the upper arm IGBT 2u (2v, 2w) caused by the energy of the electromotive force can be effectively prevented.

[0048] Furthermore, according to the configuration described above, the presence of the auxiliary control circuit 12 does not impede the function of the main control circuit 11. Therefore, the IGBT of the upper arm 2u (2v, 2w) can be switched on and off by the control voltage supplied by the main control circuit 11 during normal operation, and thus the switching loss in the IGBT of the upper arm 2u (2v, 2w) does not increase.

[0049] Furthermore, the clamping diode ZD simply clamps the voltage of the energy of the electromotive force applied to the IGBT of the upper arm 2u (2V, 2W) and allows the amount of energy whose voltage was clamped to flow as a breakdown current Ir into the voltage divider resistors RA and RB. Consequently, the breakdown current Ir flowing through the clamping diode ZD can be reduced. Therefore, the chip area of ​​the clamping diode ZD can be reduced while keeping its clamping resistance low. As a result, for example, the clamping diode ZD can also be integrated into the drive circuit of the upper arm 6u (6V, 6W).

[0050] Furthermore, there is no need for the main drive circuit 11 to generate a gate voltage required to switch on the upper arm IGBT 2u (2v, 2w) in a saturated operating state under abnormal conditions, as was previously the case, nor is there a need to increase the internal impedance (gate resistance) RG of the main drive circuit 11. Consequently, there is nothing to increase the switching loss in the upper arm IGBT 2u (2v, 2w) when the inverter 10 is operating normally. Therefore, switching at or above 10 kHz, which is a common switching frequency in the inverter 10, is easily ensured.

[0051] The Fig. Figure 2 shows a schematic configuration of the three-phase motor drive inverter 10, which is configured to include the U-phase, V-phase, and W-phase drive circuits of the upper arm 6u, 6v, and 6w, each comprising the auxiliary drive circuit 12, and the drive circuit of the lower arm 7. In particular, the inverter 10 is configured such that the clamping diode ZD is connected to a collector of each of the IGBTs of the upper arm 2u, 2v, and 2w, and the current is caused to flow via the clamping diodes ZD into the Fig. 1 voltage divider resistors RA and RB shown, which are embedded in the upper arm control circuit 6u (6v, 6w).

[0052] The U-phase, V-phase, and W-phase drive circuits of the upper arm 6u, 6v, and 6w, and the drive circuit of the lower arm 7, are integrated with the semiconductor switching element SW, which is formed from the IGBTs of the upper arm 2u, 2v, and 2w and the IGBTs of the lower arm 3u, 3v, and 3w connected as totem poles, and with the three clamping diodes ZD, thereby forming a semiconductor module 20. This semiconductor module 20 is referred to as an intelligent power module (IPM). Because this type of semiconductor module 20 is such that the breakdown current Ir of the clamping diode ZD is low and the clamping resistance is low, it is sufficient to mount a clamping diode ZD with a small chip area. Consequently, the semiconductor module 20 can be implemented as a compact semiconductor module.

[0053] Using the semiconductor module 20 configured in this way, the inverter 10, which, for example, drives a three-phase motor M, can be easily implemented. At the same time, overvoltage breakdown of the IGBTs of the upper arm 2u, 2v, and 2w can be reliably prevented if an abnormal condition, such as a short circuit, occurs in the output wiring of the inverter 10. Therefore, the semiconductor module 20 offers a large number of practical advantages.

[0054] The invention is not limited to the embodiment described above. A description has been given here in which the inverter 10, as an example, controls a three-phase motor M. However, the semiconductor module 20 can, of course, also be implemented as the inverter 10 itself, which comprises the semiconductor switching element SW for a single phase or two phases and their control circuits. Furthermore, the clamping diode ZD can, of course, also be integrated into any of the control circuits of the upper arm 6u, 6v, and 6w. It is also sufficient that the voltage clamped by the clamping diode ZD and the ratios of the voltages divided by the voltage divider resistors RA and RB are defined according to the specifications of the inverter 10, in particular the operating specifications of the semiconductor switching element SW.Furthermore, the invention can be modified in various ways without deviating from the scope of protection of the invention.

Claims

[1] Inverter control device (1), comprising: a main control circuit (11) which applies a control voltage to a semiconductor switching element (SW) provided in an inverter output stage and controls a current delivery to a load, thereby switching the semiconductor switching element (SW) on and off, wherein the semiconductor switching element (SW) is switched off when a control unit of the inverter control device detects an overcurrent; a clamping diode (ZD) that clamps a voltage of an electromotive counterforce applied to the semiconductor switching element (SW) when the operation of the main control circuit (11) stops; a voltage divider resistor (RA, RB) that resistively divides and detects a voltage proportional to a current flowing through the clamping diode (ZD) when the electromotive force voltage is clamped; and an auxiliary control circuit (12) which generates a control voltage according to the voltage detected by the voltage divider resistor (RA, RB), wherein the control voltage switches on the semiconductor switching element (SW) in an abnormal condition of the inverter, wherein when the auxiliary control circuit (12) generates the control voltage which switches on the semiconductor switching element (SW), a control voltage is applied to the semiconductor switching element (SW) as a drive voltage, wherein the drive voltage is divided via an internal impedance (RG) of the main control circuit (11) and via an output resistance (R1) of the auxiliary control circuit (12). [2] Inverter control device (1) according to claim 1, wherein the semiconductor switching element (SW) is an IGBT, The main control circuit (11) applies the drive voltage to a gate of the IGBT, thereby switching the IGBT on and off, and the auxiliary control circuit (12) applies the control voltage to the gate of the IGBT, thereby switching the IGBT on in a saturated operating range. [3] Inverter control device (1) according to claim 2, wherein the clamping diode (ZD) is formed from a Zener diode having a cathode-anode breakdown voltage that is lower than a collector-emitter breakdown voltage of the IGBT. [4] Inverter control device (1) according to claim 2, wherein the main control circuit (11) applies the control voltage to the gate of the IGBT via a gate resistor when the inverter is operating normally, and The auxiliary control circuit (12) applies the control voltage to the gate of the IGBT via an output resistor when the inverter is in an abnormal state. [5] Inverter control device (1) according to claim 1, wherein the auxiliary control circuit (12) is provided in parallel with the main control circuit (11). [6] Inverter control device (1) according to claim 1, wherein The semiconductor element is an upper arm IGBT (2u, 2v, 2w) and a lower arm IGBT (3u, 3v, 3w) connected as totem poles and alternately driven. the main control circuit (11) is formed from a control circuit of the upper arm (6u, 6v, 6w) which switches the IGBT of the upper arm on and off, and a control circuit (7) of the lower arm (3u, 3v, 3w) which switches the IGBT of the lower arm on and off, and the auxiliary control circuit (12) is provided in the control circuit of the upper arm (6u, 6v, 6w). [7] Inverter control device (1) according to claim 6, wherein the auxiliary control circuit (12) protects the IGBT of the upper arm (2u, 2v, 2w) from an electromotive counterforce applied to the IGBT of the upper arm when the IGBT of the upper arm (2u, 2v, 2w) is switched off and the IGBT of the lower arm (3u, 3v, 3w) is in an off state. [8] Inverter control device (1) according to any one of claims 1 to 7, wherein the auxiliary control circuit (12) includes two inverting amplifiers (13a, 13b) connected in two stages. [9] Semiconductor module (20) comprising a semiconductor switching element (SW) provided in an inverter output stage (10) which controls the current delivery to a consumer, and the inverter control device (1) according to any one of claims 1 to 5 which switches the semiconductor switching element (SW) on and off. [10] Semiconductor module (20) wherein two or three phases of a semiconductor switching element (SW) provided in an inverter output stage (10) and controlling the current delivery to a load (M), and the inverter control device (1) according to any one of claims 1 to 5, which switches the semiconductor switching element on and off, are integrally connected in parallel. [11] Semiconductor module (20) according to claim 9, wherein several of the inverter control devices (1) which are provided in parallel for two or three phases, individually switch on and off several of the semiconductor switching elements SW which are provided in parallel with a predetermined phase difference.

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