HYBRID POWER MEASURING SYSTEM

Abstract

A hybrid current measurement system comprising: a switch coupled to a phase output and to a high-voltage bus; an amplifier coupled to the switch and configured to detect current information from the switch; a current detector coupled between the switch and the voltage bus, the current detector being configured to determine a detected current for the switch; and a current calibrator configured to calibrate the current information for the switch based on the detected current.

Description

FIELD OF TECHNOLOGYThis disclosure relates to three-phase electric motor drives, and more particularly to three-phase electric motor drives that implement advanced control methods.PRIOR ARTIndustries such as the processing industry, healthcare and military, including consumer goods and automobiles, use and rely on electric motors. Electric motors can be found in flying drones, electric cars, and data centers.The three-phase brushless DC motor is the primary electric motor used in many of these applications because it generally has a higher torque, speed (speed), and efficiency. Further, these electric motors are capable of ensuring precise control when used with advanced control methods such as sensorless field-oriented control and direct torque control.These advanced control methods require precise monitoring of the motor parameters, including current essential for both field-oriented control and direct torque control. Precise current sensing circuits tend to increase the complexity, cost, and space requirements of the circuit.As an extension of the electronics industry, three-phase brushless DC motors and their control circuits are being subjected to an ever increasing commercial competitive pressure requiring miniaturization and low cost. Ensuring smaller, less expensive control circuits is an important strategy described in the development plans for the next generation products.Challenges for the realization of next generation products include the large, costly, and complicated current measurement systems required to utilize advanced control techniques. Illustratively, the sensorless field-oriented control, for example, requires precise current readings during each pulse width modulated signal cycle for each phase or branch of an inverter.One prior system for multi-resistor current measurement implemented power resistors or shunt resistors between a low-side drive transistor and ground for each phase of the inverter. The power resistors are bulky, expensive, and inefficient, but can provide accurate current sensing for the use of advanced control techniques.By way of illustration, a multi-resistor current measurement system for a three-phase inverter using a sensorless field-oriented regulation method would implement, for example, three power resistors. Each of these power resistors increases the number of components, the cost of the components, the complexity of the design, and the space requirement of the power stage. Each power resistor also required a current-sense amplifier (DC), an analog-to-digital converter, and associated circuitry, which further increased the cost, complexity, and number of components.Another earlier development was the single shunt current measurement system in which a single current measurement resistor is coupled between all three phases of the output stage and ground. The single shunt resistor provides cost savings, a simpler board layout, and higher efficiency. However, it is unable to provide the precise readings required for advanced engine control.If a single shunt resistor is used to determine the current across all three phases of a three phase inverter, the current senses must be performed during two separate and narrow sampling windows within each pulse width modulation cycle. These two sampling windows are further narrowed because the rising and falling edges of the power transistors require time, the current sense amplifiers used for the sampling have a minimum settling time, and many DC motors have a dead zone or nonlinearity in response to the input voltages.The single shunt current measurement system has these problems both at high pulse width modulation indices and at low pulse width modulation indices. Since the prior single shunt current measurement system requires multiple samples per pulse width modulated signal cycle, the components used must run many times faster than the pulse width modulated signal frequency and at the same time be more accurate, since the constraints on the sampling time window are much more severe than, for example, in the case of the current measurement system with multiple resistors.The single shunt current measurement system cannot provide the precise current readings that are critical to advanced motor control, requires restrictions on the maximum frequency of the pulse width modulated signal that can be used, and is limited to high current, power sensitive and limited space applications such as power tools, fans, and fans.Solutions have long been sought, but these earlier developments have not taught or suggested complete solutions and solutions to these problems have long been debred by those skilled in the art. Thus, there remains a significant need for systems and methods that reduce cost, number of components, space requirements, and complexity of power stages through advanced engine control techniques.US 2016 / 0 325 777 A1 describes a motor control device and an electric power steering device with it and a vehicle.US 2021 / 0 351 728 A1 describes a position observer for electric machines.BRIEF DESCRIPTION OF THE DRAWINGSThe hybrid current measurement system is illustrated in the figures of the accompanying drawings, which are intended to be exemplary and not limiting, in which like reference numerals refer to like components, and in which: FIG. 1 is a schematic view of the hybrid current measurement system in a first embodiment. FIG. 2 is a graph of a three-phase sinusoidal drive signal. FIG. 3 is a timing diagram for a single pulse width modulated signal cycle of the low side switches of FIG. 1 at time 3- 3 of FIG. 2. FIG. 4 is a graph of on-resistance versus temperature for one of the switches of FIG. 1. FIG. 5 is a graph of on-resistance versus current for one of the switches of FIG. 1. FIG. 6 is a control flow for operating the hybrid current measurement system of FIG. 1. FIG. 7 is a schematic view of the hybrid current measurement system in a second embodiment. FIG. 8 is a timing diagram for a single pulse width modulated signal cycle for both the high side switches and the low side switches of FIG. 7. FIG. 9 is a flow diagram for operating the hybrid current measurement system of FIG. 7. FIG. 10 is a flow diagram for manufacturing an embodiment of the hybrid current measurement system.DETAILED DESCRIPTIONIn the following description, reference is made to the accompanying drawings, which form a part of this document and in which is shown by way of illustration embodiments in which the hybrid current measurement system may be employed. It should be understood that other embodiments may be utilized and structural changes may be made without departing from the scope of the hybrid current measurement system.When features, aspects, or embodiments of the hybrid current measurement system are described in terms of steps of a process, operation, control flow, or flow chart, the steps may be combined, performed in a different order, omitted, or provided with additional steps without departing from the hybrid current measurement system described herein.The hybrid current measurement system is described in detail such that those skilled in the art can make and use the hybrid current detection system, and numerous specific details are provided to provide a thorough understanding of the hybrid current measurement system; however, it will be apparent that the hybrid current measurement system can also be used without these specific details.To avoid obscuring the hybrid current measurement system, some known system designs and methods are not disclosed in detail. These known system designs and methods include, for example, the field-oriented control without the transmitter, the space vector modulation and the pulse width modulation.Likewise, the drawings showing embodiments of the system are semi-diagrammatic and not to scale, and in particular some of the dimensions are for clarity of illustration and are greatly exaggerated in the FIGURES. In this document, the term system is defined as an apparatus or method depending on the context in which it is used.As used herein, the term "coupling" is defined as a direct or indirect electrical connection between elements. Similarly, "directly coupled" or "direct coupling" means direct contact between elements or electrical connection through wires without intervening electrical circuits or components.Referring now to FIG. 1, which shows a schematic view of the hybrid current measurement system 100 in a first embodiment. Hybrid current measurement system 100 is described, for illustrative purposes only, with reference to a power stage for driving a brushless three-phase motor. The hybrid current measurement system 100 may be adapted for other applications using pulse width modulated inverters.The power stage shown includes a regulator 102 for generating pulse width modulated signals at a regulator output 104. The regulator output 104 is connected to a three-phase inverter 106 and regulates the operation of switches therein. The pulse-width-modulated signals can be, for example, the pulse-width-modulated signals for the low-side switches shown in FIG. 3.The pulse width modulated signals drive the switches within the three-phase inverter 106 and generate the three-phase sinusoidal drive signal 202 of FIG. 2 The three-phase sinusoidal drive signal 202 is the drive signal generated, for example, to drive a three-phase brushless DC motor (not shown).In particular, the switches within the three-phase inverter 106 may be transistors and are specifically depicted as NPN field effect transistors with doped junctions, each having a gate 108, a drain 110, and a source 112. Although the transistors are depicted as NPN FETs, the hybrid current measurement system 100 may be used regardless of the type of switches used.The switches may be NFETs, complementary PFET+NFETs, eGaN FETs, or other switches. The regulator output 104 may be an individual connection from the regulator 102 to the gate 108 for each switch within the three-phase inverter 106.The three-phase inverter 106 may include six switches arranged in three phases each having two switches. That is, the three-phase inverter 106 is shown as having: a U-phase 114 having a U-output 116 and switches S 1 and S 2; a V-phase 118 having a V-output 120 and switches S 3 and S 4; and a W-phase 122 having a W-output 124 and switches S 5 and S 6.The U output 116, the V output 120, and the W output 124 are phase outputs and will be described with reference to standardized terminal markings and rotational directions for the positive electrical phase sequence U-V-W of three-phase electric motors. Each of the phase outputs, including the U output 116, the V output 120, and the W output 124, may be an input for one phase of a three-phase brushless DC motor.The U output 116 may correspond to a U current or IU, the V output 120 may correspond to a V current or IV, and the W output 124 may correspond to a W current or IW. In detail, S 1, S 3, and S 5 may be considered high side switches, the drains 110 of which are coupled to a high voltage rail (English), for example a source voltage 126. The source voltage 126 may be a positive DC voltage or Vm, the maximum voltage of the three-phase sinusoidal drive signal 202. U output 116 is shown coupled to both source 112 of S 1 and drain 110 of S 2. V output 120 is shown coupled to both source 112 of S 3 and drain 110 of S 4. W output 124 is shown coupled to both source 112 of S 5 and drain 110 of S 6.Similar to the high-side switches, S 2, S 4, and S 6 may be considered low-side switches, the sources 112 of which are coupled via a shunt resistor 130 to a low voltage rail (low voltage rail) such as ground 128. The high voltage rail and the low voltage rail are referred to as source voltage 126 and ground 128, for clarity. It is contemplated that the source voltage 126 may be any high voltage rail with a higher voltage than the low voltage rail, while the ground 128 may be any low voltage rail with a lower voltage than the high voltage rail.Voltage drops across the low-side switches and the shunt resistor 130 may be detected and monitored by current sense amplifiers, each having a positive current sense input 132, a negative current sense input 134, and a voltage output 136. The current sense amplifier may be optimized for monitoring the current of inductive loads such as DC motors and solenoids, where common mode voltages may become negative due to inductive kickback, reverse battery conditions, or transient events.For example, for illustrative purposes, a first current sense amplifier 138 may be coupled across the shunt resistor 130 to sense a voltage drop across the shunt resistor 130 that is proportional to the current through the shunt resistor 130. The current through the shunt resistor 130 may be calculated and determined by dividing the sensed voltage by the known value of the shunt resistor 130.The positive current sense input 132 of the first current sense amplifier 138 may be coupled between the sources 112 of S 2, S 4, and S 6 and the shunt resistor 130. The negative current sense input 134 of the first current sense amplifier 138 may be coupled between the shunt resistor 130 and the ground 128.While current sensing between the low-side switches and ground is depicted with a current detector including the current sense amplifier and shunt resistor 130, other current detectors are conceivable for other applications including Hall-based current detectors or anisotropic magneto-resistance current detectors. Current detectors are electrical components having a structure that allows current flow during detection registration, and other types of current detectors may be used without departing from the scope of the hybrid current measurement system 100.A second current sense amplifier 140 may be coupled across S 2 to sense a voltage drop across S 2. The positive current sense input 132 of the second current sense amplifier 140 may be coupled to the drain 110 of the S 2 and the negative current sense input 134 of the second current sense amplifier 140 may be coupled to the source 112 of the S 2.A third current sense amplifier 142 may be coupled across S 4 to sense a voltage drop across S 4. The positive current sense input 132 of the third current sense amplifier 142 may be coupled to the drain 110 of the S 4 and the negative current sense input 134 of the third current sense amplifier 142 may be coupled to the source 112 of the S 4.A fourth current sense amplifier 144 may be coupled across S 6 to sense a voltage drop across S 6. The positive current sense input 132 of the fourth current sense amplifier 144 may be coupled to the drain 110 of the S 6 and the negative current sense input 134 of the fourth current sense amplifier 144 may be coupled to the source 112 of the S 6.The second current sense amplifier 140, the third current sense amplifier 142, and the fourth current sense amplifier 144 may be any amplifiers, including voltage amplifiers or operational amplifiers, and are shown as current sense amplifiers for clarity only. The second current sense amplifier 140, the third current sense amplifier 142 and the fourth current sense amplifier 144 are to be understood here as current detectors.The voltage outputs 136 for each of the current sense amplifiers may be coupled to an analog-to-digital converter. For example, the voltage output 136 of the first current sense amplifier 138 may be coupled to a first analog-to-digital converter 152. The first analog-to-digital converter 152 may have a trigger for the calibration voltage 154.The calibration voltage trigger 154 may be a time trigger for the first analog-to-digital converter 152 to read the first current sense amplifier 138. The calibration voltage trigger 154 may be located at the center of T 1 of FIG. 3. Alternatively, it is also contemplated that the calibration voltage trigger 154 may also be located in the middle of T 2 of FIG. 3.The voltage output 136 of the second current sense amplifier 140 may be coupled to a second analog-to-digital converter 156 in a U-phase detection layer 158. The second analog-to-digital converter 156 may have a trigger for the U-phase voltage 160. The U-phase voltage trigger 160 may be set to trigger in the middle of T 3 as shown in FIG. 3.The voltage output 136 of the third current sense amplifier 142 may be coupled to a third analog-to-digital converter in a V-phase detection layer 166 and the voltage output 136 of the fourth current sense amplifier 144 may be coupled to a fourth analog-to-digital converter in a W-phase detection layer 168, which are depicted with dashed arrows. It is contemplated that the V-phase voltage trigger for the third analog-to-digital converter and the W-phase voltage trigger of the fourth analog-to-digital converter may also be located in the middle of T 3.The analog-to-digital converters may have digital outputs 170 that provide current information (current information) to the regulator 102. It goes without saying that the regulator 102 has its own structural elements as a current calibrator insofar as it must be coupled to the current measurement amplifiers via communication lines. The digital outputs 170 of the U-phase detection layer 158, the V-phase detection layer 166, and the W-phase detection layer 168 may be input to the regulator 102 as current information. The controller 102 may be a current calibrator to calibrate the current information using the digital output 170 of the first analog-to-digital converter 152.The controller 102 may be a processor or may include logic gates for processing the digital outputs 170. The controller 102 may also include circuitry for calculating position and speed using advanced control techniques. Alternatively, a remote processor could be used with the advanced control methods if they are too resource intensive for the controller 102.It has been found that if only one low-side switch is active during the detection of the first current sense amplifier 138, this detection can be used to calibrate the active switch. For example, as depicted in FIG. 3, S 6 is the only active low side switch in T 1, and sampling of the first current sense amplifier 138 may be used to calibrate the reading of the fourth current sense amplifier 144, for the temperature deviation of S 6, as shown in FIG. 4, and for the on-resistance (RON) of S 6, as shown in FIG. 5.It has been found that the shunt resistor 130 and the first current sense amplifier 138 can provide a very accurate reading during the pulse width modulated signal cycle. Since only a single read value is required during a complete cycle, the first current sense amplifier 138 and the first analog-to-digital converter 152 can be manufactured with lower requirements. That is, slower and less expensive designs may be used for the first current sense amplifier 138 and the first analog-to-digital converter 152.Similarly, it has been found that the hybrid current measurement system 100 enables the second current sense amplifier 140, the third current sense amplifier 142, and the fourth current sense amplifier 144, and the associated analog-to-digital converters, to sample only once during each pulse width modulated signal cycle. This also allows the use of a slower and cheaper design.Illustratively, a three phase brushless DC motor, for example, having eight poles and operating at 8,000 revolutions per minute, would require a drive signal of one kilohertz. The frequency of the pulse width modulated signal is normally at least twenty-four times higher than the sinusoidal frequency. Because the hybrid current measurement system 100 requires only a single sample during the pulse width modulated signal cycle, each of the analog-to-digital converters and current measurement amplifiers could run at twenty-four kilohertz.The hybrid current measurement system 100 can provide a cost reduction compared to a triple shunt solution by eliminating two shunts, which further reduces the space requirement and increases the efficiency due to fewer losses at the sensing resistors. Hybrid current measurement system 100 also provides a simpler layout, allows the use of power modules without individual feedback to ground, and is less sensitive to phase-shunt measurement mismatch due to asymmetry in the layout of the circuit board.To clarify, the high-side switches may pull up the output of their phase when activated with a high pulse width modulated signal, and when the low-side switches are activated, they may pull the output of their phase to ground, thus converting the pulse width modulated signals to the three-phase sinusoidal drive signal 202.For example, S 1 may be activated to draw U output 116 toward source voltage 126, while S 2 may be activated to draw U output 116 toward ground 128. To illustrate, the pulse width modulated signals may occur, for example, at least twenty-four times per cycle of the U-phase portion of the three-phase sinusoidal drive signal 202.Referring now to FIG. 2, a graph of a three-phase sinusoidal drive signal 202 is shown. The three-phase sinusoidal drive signal 202 may be, for example, the output of the three-phase inverter 106 of FIG. 1.Each phase of the three-phase sinusoidal drive signal 202 may be a drive input to a single phase of a three-phase electric motor. The three-phase sinusoidal drive signal 202 is shown with time along a horizontal axis and voltage along a vertical axis.The sinusoidal drive signal is advantageous for most available motors compared to other signals and is therefore used as an illustrative, non-limiting example. The regulator 102 of FIG. 1 may provide pulse width modulated signals to each switch of the three-phase inverter 106, and each phase generates a phase of the three-phase sinusoidal drive signal 202.That is, the U phase 114 of FIG. 1 may generate a U phase 204, the V phase 118 of FIG. 1 may generate a V phase 206, and the W phase 122 of FIG. 1 may generate a W phase 208.The three phase sinusoidal drive signal 202 is shown over a single electrical cycle with six sectors marked I-VI. Each of the six sectors is driven by at least four pulse-width modulated signal cycles; the present example starts from eight pulse-width modulated signal cycles per sector.Each sector is sixty degrees out of phase. A single phase is shown to have a "saddle-shaped" shape to allow full use of the supply, while the peak-to-peak voltages and currents are sinusoidal.Referring now to FIG. 3, which shows a timing diagram for a single pulse width modulated signal cycle of the low side switches 302 of FIG. 1 at time 3- 3 of FIG. 2. The pulse width modulated signal cycle 302 may provide a pulse width modulated S2 signal 304 driving S2 of FIG. 1, a pulse width modulated S4 signal 306 driving S4 of FIG. 1, and a pulse width modulated S6 signal 308 driving S6 of FIG. 1. The pulse width modulated signal cycle 302 is shown over time along a horizontal axis and the voltage along a vertical axis.The high side switches of FIG. 1 are not shown and are driven by pulse width modulated signals that are an inversion of the low side switch on the same phase. For convenience of description, this switching pattern is illustrated without limiting the scope of application of the hybrid current measurement system 100. The single pulse width modulated signal cycle 302 may be divided into states T 0, T 1, T 2, and T 3.In T 0, all low-side switches S 2, S 4 and S 6 are switched off. In T 1, the switch S 6 is active, in T 2, the switches S 6 and S 4 are active, and in T 3, all low-side switches are active. It has been found that when only a single low-side switch is active, a current through that switch can be calculated with a voltage sensed in T 1 with the first current sense amplifier 138 of FIG. 1 divided by the known value of the shunt resistor 130 of FIG. 1.In T 1, the current is sensed by only a single active low-side switch with the shunt resistor 130. This is to be understood as a "single active low-side switch" for the single pulse-width-modulated signal cycle 302. Here, the only active low-side switch would be S6.Only a single detection in T1 over the pulse width modulated signal cycle is not sufficient to use advanced control techniques, but has proven to be an excellent calibration detection for the non-linear voltage of the single active low-side switch. Here, the hybrid current measurement system 100 of FIG. 1 is shown with a low-side calibration detection 310 in the middle of T 1, which corresponds to a detection of the current through S 6 in sector I of the electrical cycle of FIG. 2.T 1 means that each of the low-side switches S 2, S 4, and S 6 is a single active low-side switch at least once during each electrical cycle. Each electrical cycle should have at least twenty-four pulse width modulated signal cycles, in the present example of FIG. 2, it includes forty-eight pulse width modulated signal cycles.The hybrid current measurement system 100 may perform low side switch detection 312 in the middle of T 3, meaning that S 2, S 4, and S 6 are all active. The low-side switch detection 312 can read the voltage via S 2 with the second current sense amplifier 140 from FIG. 1, via S 4 with the third current sense amplifier 142 from FIG. 1 and via S 6 with the fourth current sense amplifier 144 from FIG. 1, wherein these read values should be understood here as current information (current information).The detection of the voltage at S 2, S 4, and S 6 may be performed in parallel with the low side switch detection 312. The outputs of the current measurement amplifiers can be converted into a digital signal by means of analog-to-digital converters.Note that the voltage sensed at low side switch detector 312 may be used as inputs for advanced control methods such as sensorless field-oriented control or direct torque control. However, the current read at the low side switch sense 312 is subject to the temperature non-linearities of FIG. 4 and the RON current non-linearities of FIG. 5.After low side switch detection 312, hybrid current measurement system 100 may execute a remaining duration calculation step 314 of single pulse width modulated signal cycle 302. During the calculation step 314, the current information may be calibrated. In a method under consideration, calibrating the current information may include calculating a calibration weight for the single active low-side switch of T 1.The calibration calculation may begin by determining the sensed current through the single active low-side switch that was sensed during the low-side calibration sensing 310. The detected current can then be compared with an estimated current.The estimated current is calculated based on the voltage sensed across the single active low-side switch during low-side switch sensing 312. Note that during low side switch detection 312 in T 3, all low side switches are active and the reference to the single active low side switch refers only to the switch that was active during T 1.The sensed current is compared to the estimated current, the difference being a calibration weight. The estimated current plus the calibration weight may be the calibrated current. The two switches that are not calibrated during the pulse width modulated signal cycle use a calibration weight from a previous cycle.Thus, the calibrated currents for each of the low-side switches are updated about once in every third pulse width modulated signal cycle, although there are methods to increase the number of switches to two per pulse width modulated signal cycle.The calibrated current for each of the switches may be used as inputs for advanced control methods. The position and speed control of these advanced control methods can be calculated for each phase or branch based on the calibrated current for each switch, for example, by sensorless field-oriented control.In addition to calculating the calibrated current, the resistance of the single low-side active switch may be calculated by dividing the voltage detected during T 3 with the low-side switch detection 312 across the single low-side active switch by the current calculated for the single low-side active switch from the detection during T 1 with the low-side calibration detection 310. The resistance may be correlated to the temperature of the single active low-side switch using the relationship shown in FIG. 4, for example.The resistance of the single active low-side switch may also be correlated to the current, for example, the RON relationship shown in FIG. 5. It has been determined that the resistance for each of the switches may be monitored and any overtemperature may be detected and reported with overtemperature limits or used to generate an error code.The hybrid current measurement system 100 may perform an execution step 316 at the beginning of the single pulse width modulated signal cycle 302 in T 0. During execution step 316, the controller 102 of FIG. 1 may shift the pulse width modulated signals for each of the six position and speed switches based on the calibrated current information determined during the previous pulse width modulated signal cycle. Further, during execution step 316, the controller 102 may update the calibration weight for the single active low side switch of the previous pulse width modulated signal cycle.It is contemplated that hybrid current measurement system 100 may also perform further calibration measurements during the pulse width modulated signal cycle. To illustrate, the hybrid current measurement system 100 could also sense the current through the shunt resistor 130 during calibration sensing with two switches 320.The two-switch calibration detection 320 may be performed during T 2 and may provide additional current detection for a second determinable low-side switch. The current sensed by the shunt resistor 130 in T 2 may be the current of the W output 124 of FIG. 1 or IW of FIG. 1, as S 5 of FIG. 1 would be active for the W output 124. IW can be used to calculate the current through S6 with the Kirchhoff rule which specifies:That is, the current sensed at T 2 with the first current sense amplifier 138 may be used to calibrate the low side switch current sensing at S 6 with the fourth current sense amplifier 144 performed during T 3 and sensing -IW. The two-switch calibration detection 320 may increase the number of switches from one switch calibrated in each pulse width modulated signal cycle to two switches.While the hybrid current measurement system 100 has been described with reference to the controller 102 calibrating the current information with current weights and providing a calibrated current, the hybrid current measurement system 100 may also be implemented with other methods. Other methods for calibrating current information may include using analog-to-digital converters as current calibrators to calibrate the current information from the low-side switches, and thereafter the calculations for using the calibrated current information are done in software or hardware within the advanced control methods.Other methods could also include calculating a calibration error, which could be the difference between a previous calibration weight and a new calibration weight, and which can then be used to calibrate the current information by, for example, adjusting a gain of the current sense amplifier based on the calibration error. The current calibrator for calculating the calibration error could be an amplifier or the controller 102.Still other methods could include calibrating the current information for both the low-side switches and the high-side switches in digital form, for example, in the regulator 102, or in analog form, for example, using a digital-to-analog converter or error amplifier. The regulator 102, the digital-to-analog converter, the error amplifiers, or a combination thereof may therefore be used as a current calibrator.It has been found that these methods have gained importance because they can each take advantage of the calibration readings of the shunt resistor 130, in combination with less uncalibrated current information across the switches, to calibrate the current information that can be used with advanced control methods of three-phase motors.Referring now to Figure 4, there is shown a plot of drain-source resistance in the active state versus temperature for one of the switches of Figure 1. The relationship is represented with the preset fifty amp values through the switch and ten volts across the gate and source of the switch.On the vertical axis, the resistance may be represented as being sensed in 1 milli-ohm per pitch, starting at zero ohms. The horizontal axis may represent temperature in steps of forty degrees Celsius starting at minus sixty degrees.A first trace may be an illustration of a typical trace 402 showing a typical resistance-temperature relationship for one of the switches. A second trace may be an illustration of a maximum trace 404 that shows a maximum resistance-temperature relationship for one of the switches.It has been determined that the temperature of the switches may be estimated once the resistance has been calculated, for example during the calculation step 314 in FIG. 3. The temperatures may be monitored for inconsistencies or trigger an alarm if the temperature rises above a certain threshold.Referring now to FIG. 5, a graph of drain-source resistance in the active state with respect to current for one of the switches of FIG. 1 is shown. The relationship is represented with the preset values of twenty five degrees Celsius for the operating temperature of the switch.The hybrid current measurement system 100 can compensate not only for the temperature of the switch, but also for the RON current deviation. In particular, the RON value of power FET may vary significantly from batch to batch. The actual RON value may deviate from the nominal RON value. The hybrid current measurement system 100 compensates for this error by calibrating for a precise shunt resistance (or other accurate sensing element).On the vertical axis, the resistance may be represented as being sensed in 1 milli-ohm per pitch, starting at zero ohms. The horizontal axis may represent current in steps of fifty amps starting at zero amps.The relationship may be represented with a five volt trace 502, a five and a half volt trace 504, a six volt trace 506, a seven volt trace 508, and a ten volt trace 510. In particular, the current detected by the shunt resistor 130 of FIG. 1 in the middle of T 1 during the low side calibration detection 310 of FIG. 3 is the current of the single active low side switch and may be compared to an estimated current of the single active low side switch based on the voltage detection during T 3 of the low side switch detection 312 of FIG. 3. The difference may be a calibration weight for the single active low-side switch. The traces represent the relationship between current and resistance and depend on the voltage across the switch and the temperature of the switch.Referring now to FIG. 6, a control flow for the operation of the hybrid current measurement system 100 of FIG. 1 is shown. The hybrid current measurement system 100 may begin executing an initialization step 602. During the initialization step 602, the calibration weights 604 may be initialized for each of the switches. Continuing with the example of FIG. 1, three calibration weights may be initialized, one for each of the U-phase 114, one for the V-phase 118, and one for the W-phase 122 of FIG. 1.The operation of the hybrid current measurement system 100 is described for purposes of description with respect to the time states of FIG. 3, but is not limited thereby. After the initialization step 602, the hybrid current measurement system 100 may perform the low-side calibration detection 310 in the middle of T 1 in FIG. 3. During low-side calibration detection 310, a single active low-side switching voltage 612 across shunt resistor 130 of FIG. 1 may be detected with first current sense amplifier 138 of FIG. 1.After the low-side calibration detection 310, the hybrid current measurement system 100 may perform the two-switch calibration detection 320 in the middle of T 2 of FIG. 3. During the two-switch calibration detection 320, the current detected by the shunt resistor 130 in T 2 may be the current of the W output 124 of FIG. 1 or IW of FIG. 1, as S 5 of FIG. 1 would be active for the W output 124. The IW may be used to calculate the current through S 6 of FIG. 1 with the Kirchhoff rule of Equation 1.The hybrid current measurement system 100 may next perform the low-side switch detection 312 in the middle of T 3 in FIG. 3. During low side switch detection 312, the voltages at each of the low side switches are detected including an S2 voltage 624, an S4 voltage 626, and an S6 voltage 628.After the middle of T 3 and during the second half of the pulse width modulated signal cycle, the hybrid current measurement system 100 may perform the calculation step 314. The calculation step 314 may include many smaller calculations.To illustrate, for example, the calculating step 314 may include a step of calculating the estimated current 630. During the estimated current computing step 630, the voltages detected during the low side switch detection 312 may be used to compute the estimated currents 632 for each of the three low side switches by dividing the voltages detected during the low side switch detection 312 by an average resistance or a known resistance.Further, during the calculating step 314, the hybrid current measurement system 100 may execute a step of calculating the sensed current 634. During the step of calculating the sensed current 634, the voltage of the single low-side active switch 612 may be divided by the value of the shunt resistor 130 to obtain a sensed current of the single low-side active switch 636.An additional sensed current may be calculated. To illustrate, the two-switch calibration detection 320 is performed during T 2, since S 5 is active for the W output 124 in T 2, the current detected by the shunt resistor 130 in T 2 may be the current of the W output 124 or IW. The IW may be used to calculate the current through S 6 of FIG. 1 with the Kirchhoff rule of Equation 1.The current through the second determinable low-side switch may be a Kirchhoff current derived from the two-switch calibration detection 320 in T2or a current 642 derived from the low-side switch.Once the step of calculating the sensed current 634 is completed, the hybrid current measurement system 100 may execute a step of calculating the calibration weights 644. The calibration weight calculating step 644 may compare the sensed currents determined in the sensed current calculating step 634 with the estimated currents 632 determined in the estimated current calculating step 630.In particular, the difference between the sensed current 636 of the single active low-side switch and the estimated current 632 for that switch may be the calibration weight 604 for that switch. Similarly, the differences between the low side switch derived current 642 can be compared to the estimated current 632 for the respective switch. In the case of FIG. 3, the corresponding switch would be S 6.The calibration weights 604 for the three switches may be updated. Once the step of calculating the calibration weights 644 is complete, the hybrid current measurement system 100 may execute a step of calculating the calibrated current 646.The calibrated current calculation step 646 may sum the estimated current 632 for each switch with the calibration weight 604 for each switch to produce a calibrated current 648 for each switch. Note that unless a calibration weight 604 is calculated for each switch, a previous or initial calibration weight 604 may be summed with the estimated current 632 to determine the calibrated current 648.It has been found that using the estimated currents 632 allows each switch on the low side to be measured at each pulse width modulated signal cycle, providing excellent information input for advanced control methods. The calibration of one or two switches per pulse width modulated signal cycle increases the accuracy of the advanced control method.Once the calibrated currents 648 have been determined, the calibrated currents 648 may be used as inputs in a step of an advanced control method 650. During the step of an advanced control process 650, the calibrated current 648 for each of the switches may be used as input for advanced control processes such as the sensorless field-oriented control and the direct torque control. The position and speed control of these advanced control methods may be calculated for each phase or branch based on the calibrated current 648 for each switch.After the calculating step 314 is performed, the hybrid current measurement system 100 may perform the executing step 316. During execution step 316, the controller 102 of FIG. 1 may shift the pulse width modulated signals for each of the six position and speed control switches based on the calibrated currents 648 determined during the previous pulse width modulated signal cycle. Further, during execution step 316, the controller 102 may update the calibration weights 604 for the single active low-side switch and the second determinable low-side switch of the previous pulse width modulated signal cycle.Execution step 316 may be performed at the beginning of T 0. Both the calculating step 314 and the executing step 316 may be performed by a processor in the controller 102. Alternatively, some of the more complex computations may also be performed by a remote processor communicating with the controller 102.It is contemplated that the calculating step 314 may further include a calculated temperature value for each of the switches determined in the step of calculating the sensed current 634. The temperature value may be the voltages sensed during the low side switch sensing 312 divided by a corresponding one of the sensed currents determined in the sensed current calculating step 634. That is, for example, voltage 628 of S 6 could be divided by the sensed current 636 of the single active low side switch to determine the resistance of S 6 corresponding to temperature, as shown in FIG. 4.The resistance for each of the switches may be determined once the current has been sensed with the first current sense amplifier 138 for the respective switch. A resistance threshold may be used to generate a fault when the resistance of the switch exceeds the resistance threshold.Referring now to FIG. 7, which shows a schematic view of the hybrid current measurement system 700 in a second embodiment. Hybrid current measurement system 700 is described for illustrative purposes only with respect to a power stage of a brushless three-phase motor drive, and could also be adapted to other applications using pulse width modulated inverters, such as those used with solenoid valves.The power stage is shown with a regulator 702 for generating pulse width modulated signals at a regulator output 704 connected to switches in a three phase inverter 706. The pulse-width-modulated signals can be, for example, the pulse-width-modulated signals for the low-side switches shown in FIG. 3, but also pulse-width-modulated signals for the high-side switches.The pulse width modulated signals drive switches within the three-phase inverter 706 and generate the three-phase sinusoidal drive signal 202 of FIG. 2 The three-phase sinusoidal drive signal is the drive signal 202 generated, for example, to drive a three-phase brushless DC motor (not shown).In particular, the switches within the three-phase inverter 706 may be transistors and are specifically depicted as NPN field effect transistors with doped junctions, each having a gate 708, a drain 710, and a source 712. Although the transistors are depicted as NPN FETs, the hybrid current measurement system 100 may be used regardless of the type of switches used.The switches may be, for illustration purposes, NFETs, complementary PFET+NFETs, eGaN FETs, or other switches. The regulator output 704 may be an individual connection from the regulator 702 to the gate 708 for each switch within the three-phase inverter 706.The three-phase inverter 706 may include six switches arranged in three phases with two switches each. That is, three-phase inverter 706 is shown to have: a U-phase 714 having a U-output 716 and switches S 1 and S 2; a V-phase 718 having a V-output 720 and switches S 3 and S 4; and a W-phase 722 having a W-output 724 and switches S 5 and S 6.The U output 716, the V output 720, and the W output 724 are phase outputs and will be described with reference to standardized terminal markings and rotational directions for the positive electrical phase sequence U-V-W of three-phase electric motors. Each of the phase outputs, including the U output 716, the V output 720, and the W output 724, may be an input for a phase or winding of a three-phase brushless DC motor.In particular, S 1, S 3, and S 5 may be considered high side switches, the drains 710 of which are coupled to a high voltage rail, such as a source voltage 726, via a high side current detector 725. The source voltage 726 may be a positive DC voltage or Vm that is the maximum voltage of the three-phase sinusoidal drive signal 202. The high-side current detector 725 is depicted as a general component block that representatively shows a current sense amplifier connected via a shunt resistor, a Hall-based current detector, or an anisotropic magneto-resistance current detector.U output 716 is shown coupled to both source 712 of S 1 and drain 710 of S 2. V output 720 is shown coupled to both source 712 of S 3 and drain 710 of S 4. W output 724 is shown coupled to both source 712 of S 5 and drain 710 of S 6.S2, S4, and S6 may be considered low side switches, the sources 712 of which are coupled to a low voltage rail, such as ground 728, via the low side current detector. The high voltage rail and the low voltage rail are referred to as source voltage 726 and ground 728 for clarity. It should be appreciated that the source voltage 726 may be any high voltage rail with a higher voltage than the low voltage rail, while the ground 728 may be any low voltage rail with a lower voltage than the high voltage rail.Voltage drops across the low-side switches, high-side switches, and shunt resistor 730 may be detected and monitored by current sense amplifiers, each having a positive current sense input 732, a negative current sense input 734, and a voltage output 736. The current sense amplifier may be optimized for monitoring the current of inductive loads such as DC motors and solenoids, where common mode voltages may become negative due to inductive kickback, reverse battery conditions, or transient events.To illustrate, for example, a first current sense amplifier 738 may be coupled across the shunt resistor 730 to sense a voltage drop across the shunt resistor 730 that is proportional to the current through the shunt resistor 730. The current through shunt resistor 730 may be calculated and determined by dividing the sensed voltage by the known value of shunt resistor 730.The positive current sense input 732 of the first current sense amplifier 738 may be coupled between the sources 712 of S 2, S 4, and S 6 and the shunt resistor 730. The negative current sense input 734 of the first current sense amplifier 738 may be coupled between the shunt resistor 730 and ground 728.While reading the current values between the switches of the three-phase inverter 706 and ground is depicted with the low-side current detector having the first current sense amplifier 738 and the shunt resistor 730, other current detectors are also conceivable for other applications, for example Hall-based current detectors or anisotropic magneto-resistance current detectors. Current detectors are electrical components having a structure that allows current flow during sensing, and other types of current detectors may be used without departing from the scope of the hybrid current measurement system 700.A second current sense amplifier 740 may be coupled across S 2 to sense a voltage drop across S 2. The positive current sense input 732 of the second current sense amplifier 740 may be coupled to the drain 710 of the S 2 and the negative current sense input 734 of the second current sense amplifier 740 may be coupled to the source 712 of the S 2.A third current sense amplifier 742 may be coupled across S 4 to sense a voltage drop across S 4. The positive current sense input 732 of the third current sense amplifier 742 may be coupled to the drain 710 of the S 4 and the negative current sense input 734 of the third current sense amplifier 742 may be coupled to the source 712 of the S 4.A fourth current sense amplifier 744 may be coupled across S 6 to sense a voltage drop across S 6. The positive current sense input 732 of the fourth current sense amplifier 744 may be coupled to the drain 710 of the S 6, and the negative current sense input 734 of the fourth current sense amplifier 744 may be coupled to the source 712 of the S 6.A fifth current sense amplifier 746 may be coupled across S 5 to sense a voltage drop across S 5. The positive current sense input 732 of the fifth current sense amplifier 746 may be coupled to the drain 710 of the S 5, and the negative current sense input 734 of the fifth current sense amplifier 746 may be coupled to the source 712 of the S 5.A sixth current sense amplifier 748 may be coupled across S 3 to sense a voltage drop across S 3. The positive current sense input 732 of the sixth current sense amplifier 748 may be coupled to the drain 710 of the S 3 and the negative current sense input 734 of the sixth current sense amplifier 748 may be coupled to the source 712 of the S 3.A seventh current sense amplifier 750 may be coupled across S 1 to sense a voltage drop across S 1. The positive current sense input 732 of the seventh current sense amplifier 750 may be coupled to the drain 710 of the S 1 and the negative current sense input 734 of the seventh current sense amplifier 750 may be coupled to the source 712 of the S 1.The second current sense amplifier 740, the third current sense amplifier 742, the fourth current sense amplifier 744, the fifth current sense amplifier 746, the sixth current sense amplifier 748, and the seventh current sense amplifier 750 may be any amplifiers, such as voltage amplifiers or operational amplifiers, and are disclosed as current sense amplifiers for clarity only. The second current sense amplifier 740, the third current sense amplifier 742, the fourth current sense amplifier 744, the fifth current sense amplifier 746, the sixth current sense amplifier 748 and the seventh current sense amplifier 750 are to be understood here as current detectors.The voltage outputs 736 for each of the current sense amplifiers may be coupled to an analog-to-digital converter. For example, the voltage output 736 of the first current sense amplifier 738 may be coupled to a first analog-to-digital converter 752. The first analog-to-digital converter 752 may have a trigger for the calibration voltage 754.The calibration voltage trigger 754 may be a time trigger for the first analog-to-digital converter 752 to read the first current sense amplifier 738. The calibration voltage trigger 754 may be located at the center of T 1 of FIG. 3. Alternatively, it is also contemplated that the calibration voltage trigger 754 may also be located in the middle of T 2 of FIG. 3.The high-side current detector 725 could be similarly coupled to the regulator 702 via an analog-to-digital converter having a calibration voltage trigger. It has been found that the sensed current for the switches may be determined with either the high side current detector 725 or a low side current detector including the first current sense amplifier 738 and the shunt resistor 730.That is, the sensed current may be determined for each of the phases by either the high side current detector 725 or the low side current detector using identical methods, and all methods disclosed herein with respect to current sensing with the low side current detector may also be used with the high side current detector. Functionally, either the high-side current detector or the low-side current detector may be used to determine the sensed current, and both may be triggered at the same times to read the same phases of the three-phase sinusoidal drive signal 202.The voltage output 736 of the second current sense amplifier 740 may be coupled to a low-side U-phase analog-to-digital converter 756 in a U-phase detection layer 758. The low-side U-phase analog-to-digital converter 756 may have a low-side trigger for the U-phase voltage 760. The low side trigger for the U-phase voltage 760 may be set to trigger in the middle of T 3 as shown in FIG. 3.The voltage output 736 of the seventh current sense amplifier 750 may be coupled to a high side U-phase analog-to-digital converter 762 in the U-phase detection layer 758. The high-side U-phase analog-to-digital converter 762 may have a high-side trigger for the U-phase voltage 764. The high side trigger for the U-phase voltage 764 may be set to trigger in the middle of T 0 with all high side switches turned on.The voltage output 736 of the third current sense amplifier 742 may be coupled to a low-side V-phase analog-to-digital converter in a V-phase detection layer 766. The voltage output 736 of the sixth current sense amplifier 748 may be coupled to a high-side V-phase analog-to-digital converter in the V-phase detection layer 766.The voltage output 736 of the fourth current sense amplifier 744 may be coupled to a low-side W-phase analog-to-digital converter in a W-phase detection layer 768. The voltage output 736 of the fifth current sense amplifier 746 may be coupled to a high-side W-phase analog-to-digital converter in the W-phase detection layer 768.The connections between the voltage outputs 736 of the third current sense amplifier 742 to the low-side V-phase analog-to-digital converter, between the voltage outputs 736 of the fourth current sense amplifier 744 to the low-side W-phase analog-to-digital converter, between the voltage outputs 736 of the fifth current sense amplifier 746 to the high-side W-phase analog-to-digital converter and between the voltage outputs 736 of the sixth current sense amplifier 748 to the high-side V-phase analog-to-digital converter are depicted with dashed arrows.It is contemplated that the low-side V-phase voltage trigger for the low-side V-phase analog-to-digital converter and the low-side W-phase voltage trigger of the low-side W-phase analog-to-digital converter may also be in the middle of T 3. It is further contemplated that the high side V-phase voltage trigger for the high side V-phase analog-to-digital converter and the high side W-phase voltage trigger of the high side W-phase analog-to-digital converter may be at the center of T 0.It is contemplated that an alternative topology to the high-side and low-side analog-to-digital converters may be a single analog-to-digital converter for each phase along with a multiplexer. Fewer components may be used in this alternative topology.The analog-to-digital converters may have digital outputs 770. The digital outputs 770 of the U-phase detection layer 758, the V-phase detection layer 766 and the W-phase detection layer 768 can be input to the regulator 702 as current information. The controller 702 may be a current calibrator to calibrate the current information using the digital output 770 of the first analog-to-digital converter 752 as described with reference to the calculating step 824 of FIGS. 8 and 9.The controller 702 may be a processor or may include logic gates for processing the digital outputs 770. The controller 702 may also include circuitry for calculating position and speed using advanced control techniques. Alternatively, a remote processor could be used with the advanced control methods if they are too resource intensive for the controller 702.It has been found that if only one low-side switch or high-side switch is active during sensing at shunt resistor 730, that sensing may be used to calibrate the corresponding active switch. For example, as depicted in FIG. 8, S 1 is the only high side switch that is active in T 2, and sampling of the first current sense amplifier 738 may be used to calibrate the reading of the seventh current sense amplifier 750 for the temperature deviation of S 1, as shown in FIG. 4, and for the RON of S 1, as shown in FIG. 5.It has been found that the low-side current detector comprising the shunt resistor 730 and the first current sense amplifier 738 or the high-side current detector 725 can provide a very accurate reading during the pulse width modulated signal cycle. Since only a single read value is needed during a complete cycle, the first current sense amplifier 738 and the first analog-to-digital converter 752 can be manufactured with lower requirements. That is, slower and cheaper designs may be used for the first current sense amplifier 738, the first analog-to-digital converter 752, and the components of the high-side current detector 725.Similarly, it has been found that the hybrid current measurement system 700 enables the second current sense amplifier 740, the third current sense amplifier 742, and the fourth current sense amplifier 744 and the associated analog-to-digital converters during each pulse width modulated signal cycle to sample only once. This also allows the use of a slower and cheaper design.Illustratively, a three phase brushless DC motor, for example, having eight poles and operating at 8,000 revolutions per minute, would require a drive signal of one kilohertz. The frequency of the pulse width modulated signal is normally at least twenty-four times higher than the sinusoidal frequency. Because the hybrid current measurement system 700 requires only a single sample during the pulse width modulated signal cycle, each of the analog-to-digital converters and current measurement amplifiers could run at twenty-four kilohertz.Thus, hybrid current measurement system 700 may eliminate the limitations of the sampling window, as both T0 and T3 are wide enough for sampling. Further, advanced control methods such as sensorless field-oriented control and direct torque control may be used because the voltage across all high-side switches is obtained during T 0 and all low-side switches during T 3.It has been determined that the voltage senses at the high-side switches and the low-side switches, along with an estimate of the current based on the sensed voltage, are sufficient to enable and apply the advanced control methods. The relationship between the detected voltages and the actual currents will vary due to the relationships shown in FIGS. 4 and 5, but this may be compensated.An additional advantage of the second embodiment of FIG. 8 compared to the first embodiment of FIG. 1 is that the three phase current information can be recorded twice per electrical cycle (center of T 0 and center of T 3), thus doubling the theoretical bandwidth of the current control loop. The calibration may be done in the middle of T3 for the low side switches and in the middle of T0 for the high side switches.Another advantage of the hybrid current measurement system 700 is that the resistance of the switches, along with their temperature, can be monitored for deviations or conditions above a preset warning or replacement limit.Referring now to FIG. 8, which shows a timing diagram for a single pulse width modulated signal cycle 802 for both the high side switches and the low side switches of FIG. 7. The pulse width modulated signal cycle 802 may provide: an S1 pulse width modulated signal 804 driving S1 of FIG. 7, an S2 pulse width modulated signal 806 driving S2 of FIG. 7, an S3 pulse width modulated signal 808 driving S3 of FIG. 7, a pulse width modulated S4 signal 810 driving S4 of FIG. 7, a pulse width modulated S5 signal 812 driving S5 of FIG. 7, and a pulse width modulated S6 signal 814 driving S6 of FIG. 7. The pulse width modulated signal cycle 802 is represented over time along a horizontal axis and the voltage along a vertical axis.The S1 pulse width modulated signal 804, the S3 pulse width modulated signal 808, and the S5 pulse width modulated signal 812 represent the high side switches and, as is typically the case, are presented inverted to the S2 pulse width modulated signal 806, the S4 pulse width modulated signal 810, and the S6 pulse width modulated signal 814, which represent the low side switches. For convenience of description, this switching pattern is illustrated, but does not limit the scope of application of the hybrid current measurement system 700.The single pulse width modulated signal cycle 802 may be divided into states T 0, T 1, T 2, and T 3. In T 0, all low-side switches S 2, S 4 and S 6 are inactive, while all high-side switches S 1, S 3 and S 5 are active. In T1, switch S6 is active, while S5 is inactive. In T2, switches S6 and S4 are active, while S5 and S3 are inactive. In T 3, all low-side switches are active, while all high-side switches are inactive.It has been found that during T 1 when only one low side switch is active and during T 2 when only one high side switch is active, a current through the corresponding active switch may be calculated with voltage sensing across shunt resistor 730 of FIG. 7. The current detection and detection in T 1 for one of the low-side switches may be based on the low-side current detector having a voltage detected in T 1 with the first current sense amplifier 738 of FIG. 7 divided by the known value of the shunt resistor 730 or a reading of the high-side current detector 725 of FIG. 7.Similarly, the current detection and detection in T 2 for one of the high side switches may be based on the low side current detector having a voltage detected in T 2 with the first current sense amplifier 738 divided by the known value of the shunt resistor 730 or a reading of the high side current detector 725. Illustratively, here, a low-side calibration detection 816 may be triggered in the middle of T 1 to calculate a current through S 6, the corresponding active switch. Further, in the middle of T2, a high side calibration detection 818 may be triggered to calculate a current through S1.Each pulse width modulated signal cycle may include a state in which a corresponding switch on the high side and the low side may be individually activated relative to other switches on their side. Over an entire electrical cycle, which may include far over twenty-four pulse width modulated signal cycles, each of the switches is sensed and calibrated.The low-side calibration detection 816 in the middle of T 1 corresponds to each of the low-side switches S 2, S 4, and S 6 at least once during a single electrical cycle that should have at least twenty-four pulse width modulated signal cycles. Similarly, the high side calibration detection 818 at the center of T 2 corresponds to each of the high side switches S 1, S 3, and S 5 at least once during a single electrical cycle. The current detections in T 1 and T 2 may be performed by the low-side current detector, the high-side current detector 725, or a combination of both.In T 1, the current through only one of the low-side switches is detected with the low-side current detector or with the high-side current detector 725. This is to be understood as the "corresponding switch" for the single pulse-width modulated signal cycle 802 on the low side. Here, the corresponding switch S 6 would be for low-side sensing. Note that in T1, both S1 and S3 are active on the high side.In T 2, the current through only one of the high-side switches is detected with the low-side current detector or with the high-side current detector 725. This is to be understood as the "corresponding switch" for the single pulse-width modulated signal cycle 802 on the high side.Here, the corresponding switch S 1 would be high-side sensing. Note that in T2, both S4 and S6 are active on the low side. Only a single detection in T1 for the low side and T2 for the high side over the pulse width modulated signal cycle is not sufficient to use advanced control techniques, but has proven to be excellent calibration detection for the non-linear voltage of the respective switches.The hybrid current measurement system 700 may perform high side switch sensing 820 at the center T 0 and low side switch sensing 822 at the center T 3. The high side switch sense 820 may result in all high side switches S 1, S 3, and S 5 being activated. The low-side switch sense 822 may correspond to the current information of all low-side switches S 2, S 4, and S 6 that are inactive.The low-side switch sense 822 may read the voltage via S 2 with the second current sense amplifier 740 of FIG. 7, via S 4 with the third current sense amplifier 742 of FIG. 7, and via S 6 with the fourth current sense amplifier 744 of FIG. 7. The high-side switch sense 820 may read the voltage across S 1 with the seventh current sense amplifier 750 of FIG. 7, across S 3 with the sixth current sense amplifier 748 of FIG. 7, and across S 5 with the fifth current sense amplifier 746 of FIG. 7.It should be appreciated that the low side switch sense 822 of the voltages at S 2, S 4, and S 6 and the high side switch sense 820 of the voltages at S 1, S 3, and S 5 collect current information for each of the low side switches and the high side switches. The current information collected during low side switch detection 822 and high side switch detection 820 is uncalibrated.The low side switch detection 822 of the voltage at S 2, S 4, and S 6 and the high side switch detection 820 of the voltages at S 1, S 3, and S 5 may be performed in parallel. Note that the voltage detected in low side switch detection 822 and high side switch detection 820 may be used as inputs for advanced control methods such as sensorless field-oriented control or direct torque control.The voltage sensed at low-side switch sense 822 and high-side switch sense 820 may be used to estimate the currents through each of the switches, however, these current estimates are subject to the temperature non-linearities of FIG. 4 and the current non-linearities due to the RON of FIG. 5. Although the current estimates are not accurate, they are close enough to use the advanced control methods and can be further calibrated using the low-side calibration sense 816 and the high-side calibration sense 818.During the second half of the pulse width modulated signal cycle 802, the hybrid current measurement system 700 may perform a calculation step 824. During the calculation step 824, the uncalibrated current information determined during the low-side switch detection 822 and the high-side switch detection 820 may be calibrated. In one of the contemplated methods, calibrating the current information may include calculating calibration weights for the single active low-side switch of T 1 and the single active high-side switch of T 2.The calibration calculation may begin by determining the sensed current through the single active low-side switch that was sensed during the low-side calibration sensing 816 and determining the sensed current through the single active high-side switch during the high-side calibration sensing 818. Then, the detected currents are compared with the estimated currents for each of the switches.The estimated current is calculated based on the voltage detected during low side switch detection 822 for the single active low side switch. Similarly, the estimated current is calculated based on the voltage detected across the single active high-side switch during high-side switch detection 820.Note that during low side switch detection 822 in T 3, all low side switches are active and the reference to the single active low side switch refers only to the switch that was active during T 1.The sensed current is compared to the estimated current, the difference being a calibration weight. The estimated current plus the calibration weight may be the calibrated current. The two high-side switches and the two low-side switches that are not calibrated during the pulse width modulated signal cycle use a calibration weight from a previous cycle.Thus, the calibrated currents for each of the low-side switches and each of the high-side switches are updated about once in every third pulse-width modulated signal cycle, although there are methods to increase the number of switches per cycle.The calibrated current for each of the switches may be used as input for advanced control methods such as sensorless field-oriented control and direct torque control. The position and speed control of these advanced control methods may be calculated for each phase or branch based on the calibrated current for each switch.In particular, the current through the shunt resistor 730 of the low-side current detector or through the high-side current detector 725 may be determined during T 1 or T 2. This current in T 1 or T 2 corresponds to the current through the individual active low-side switch or the individual active high-side switch.In addition to calculating the calibrated current, the resistance of the single low-side active switch may be calculated by dividing the voltage detected during T 3 with low-side switch detection 822 across the single low-side active switch by the current calculated for the single low-side active switch from the detection during T 1 with low-side calibration detection 816. Similarly, the resistance of the single high-side active switch may be calculated by dividing the voltage detected during T 0 with high-side switch detection 820 across the single high-side active switch by the current calculated for the single high-side active switch using the detection during T 2 with high-side calibration detection 818.The resistances may be correlated to the temperature of the single low-side active switch and the single high-side active switch using, for example, the relationship shown in FIG. 4. The resistance of the single active low-side switch and the single active high-side switch may also be correlated to the current, for example, the RON relationship shown in FIG. 5. It has been determined that the resistance for each of the switches may be monitored and any overtemperature may be detected and reported with overtemperature limits or used to generate an error code.The hybrid current measurement system 700 may perform an execution step 826 at the beginning of the single pulse width modulated signal cycle 802, which is at the beginning of T 0. During execution step 826, controller 702 of FIG. 7 may shift the pulse width modulated signals for each of the six position and speed control switches based on the readings of the previous pulse width modulated signal cycle. Further, during execution step 826, the controller 702 may update the calibration weight for the single active low-side switch and the single active high-side switch of the previous pulse width modulated signal cycle.It is contemplated that hybrid current measurement system 700 may also perform further calibration measurements during the pulse width modulated signal cycle. For illustration, for example, hybrid current measurement system 700 could also sense the current through low-side current detector shunt resistor 730 or high-side current detector 725 during additional low-side or high-side calibration sensing. It is further contemplated that during low-side calibration detection 816, a inferred high-side switching current may be calculated using the Kirchhoff rule.During high-side calibration detection 818, a derived low-side switch may be derived in addition to the single active low-side switch. The current through the derived low-side switch can be determined with the aid of the Kirchhoff rule. By determining the current through the inferred high-side switch and the inferred low-side switch, four of the six switches may be calibrated once per pulse width modulated signal cycle.Although the hybrid current measurement system 700 has been described with reference to the controller 702, which calibrates the current information with current weights and provides a calibrated current using the controller 702 as a current calibrator, the hybrid current measurement system 700 may be implemented using other methods. Other methods for calibrating current information may include using analog-to-digital converters as current calibrators for calibrating current information from the low-side switches and high-side switches, and thereafter, the calculations for using the calibrated current information are done in software or hardware within the advanced control methods.Other methods could also include calculating a calibration error, which could be the difference between a previous calibration weight and a new calibration weight, and which can then be used to calibrate the current information by, for example, adjusting a gain of the current sense amplifier based on the calibration error. The current calibrator for calculating the calibration error could be an amplifier or the controller 702.Still other methods could include calibrating the current information for both the low-side switches and the high-side switches in digital form, for example, in the regulator 702, or in analog form, for example, using a digital-to-analog converter or error amplifier. The regulator 702, the digital-to-analog converter, the error amplifiers, or a combination thereof may therefore be used as a current calibrator.It has been recognized that these methods have gained importance because they can each take advantage of the calibration readings of the shunt resistor 130, in combination with less uncalibrated current information across the switches, to calibrate the current information that can be used with advanced control methods of three-phase motors.Referring now to FIG. 9, a flowchart for operating an embodiment of the hybrid current measurement system 700 of FIG. 7 is shown. The hybrid current measurement system 700 may begin executing an initialization step 902. During the initialization step 902, the calibration weights 904 for each of the switches may be initialized. Continuing with the example of FIG. 7, six calibration weights may be initialized, two for each of the U-phase 714, two for each of the V-phase 718, and two for each of the W-phase 722 of FIG. 7.Operation of the hybrid current measurement system 700 may next proceed with the high side switch sense 820 in the middle of T 0 of FIG. 8. During high side switch detection 820, the voltages at each of the high side switches are detected, including an S1 voltage 906, an S3 voltage 908, and an S5 voltage 910.After high side switch detection 820, hybrid current measurement system 700 may perform low side calibration detection 816 in the middle of T 1 of FIG. 8. During low-side calibration detection 816, a single active low-side switching voltage 912 may be detected with the low-side current detector via shunt resistor 730 of FIG. 7 with first current sense amplifier 738 of FIG. 7, or detected with high-side current detector 725 of FIG. 7.After the low-side calibration detection 816, the hybrid current measurement system 700 may perform the high-side calibration detection 818 in the middle of T 2 of FIG. 8. During high side calibration detection 818, a single active high side switching voltage 914 may be detected with the low side current detector via shunt resistor 730 with first current sense amplifier 738 or with high side current detector 725.The hybrid current measurement system 700 may next perform the low side switch detection 822 in the middle of T 3 of FIG. 8. During low-side switch detection 822, the voltages at each of the low-side switches are detected including an S2 voltage 924, an S4 voltage 926, and an S6 voltage 928.After the middle of T 3 and during the second half of the pulse width modulated signal cycle, the hybrid current measurement system 700 may perform the calculation step 824. The calculation step 824 may include many smaller calculations.To illustrate, for example, the calculating step 824 may include a step of calculating the estimated current 930. During the estimated current calculation step 930, the voltages detected during the low side switch detection 822 and the high side switch detection 820 may be used to calculate the estimated currents 932 for each of the three high side switches and the three low side switches.Further, during the calculating step 824, the hybrid current measurement system 700 may execute a step to calculate the sensed current 934. During the step of calculating the sensed current 934, the sensed current may be determined with either the low side current detector or the high side current detector 725.To illustrate, for example, when the low-side current detector is used, the voltage of the single low-side active switch 912 may be divided by the value of the shunt resistor 730 to obtain a sensed current of the single low-side active switch 936. Similarly, using the low-side current detector, the voltage of the single high-side active switch 914 may be divided by the value of the shunt resistor 730 to obtain a sensed current of the single high-side active switch 938.Two additional Kirchhoff-derived currents may be calculated, including a Kirchhoff-derived current for a second determinable low-side switch and a second determinable high-side switch. To illustrate, if the low-side calibration detection 816 is made during T 1, the W output 124 of FIG. 1 is active and is detected via the shunt resistor 130 of FIG. 1. IW of FIG. 1 from the W output 124 may be used to determine a high side switch derived current 940 using the Kirchhoff rule of equation 1. Here, the current 940 derived from the high side switch may correspond to a second determinable high side switch, and in this example, the high side switch of the W output 124 or S 5 of FIG. 1. Similarly, when the high side calibration detection 818 is performed during T 2, S 1 of FIG. 1 is active and the shunt resistor 130 detects the current IU of FIG..1, The current may be used to calculate a low side switch derived current 942 using the Kirchhoff rule of Equation 1. Here, the derived low-side switch current 942 may correspond to a second determinable low-side switch, and in this example, the low-side switch of the U output 116 of FIG. 1 or S 2 of FIG. 1.Once the step of calculating the sensed current 934 is completed, the hybrid current measurement system 700 may execute a step of calculating the calibration weights 944. The calibration weight calculating step 944 may compare the sensed currents determined in the sensed current calculating step 934 with the estimated currents 932 determined in the estimated current calculating step 930.In particular, the difference between the sensed current 936 of the single active low-side switch and the estimated current 932 for that switch may be the calibration weight 904 for that switch. Similarly, the differences between the sensed current 938 of the single active high-side switch, the inferred current 942 of the low-side switch, and the inferred current 940 of the high-side switch may be compared to the estimated current 932 for the respective switch.The calibration weights 904 for the four switches may be updated. Once the step of calculating the calibration weights 944 is complete, the hybrid current measurement system 700 may execute a step of calculating the calibrated current 946.The calibrated current calculation step 946 may calibrate the voltage detected across the switch by adding the estimated current 932 for each switch to the calibration weight 904 for each switch to produce a calibrated current 948 for each switch. Note that unless a calibration weight 904 is calculated for each switch, a previous or initial calibration weight 904 may be summed with the estimated current 932 to determine the calibrated current 948.It has been found that using the estimated currents 932 allows each switch on the high side and on the low side to be detected in each pulse width modulated signal cycle, which provides excellent information input for advanced control methods. Calibration of one, two or four switches per pulse width modulated signal cycle increases the accuracy of the advanced control method.Once the calibrated currents 948 have been determined, the calibrated currents 948 may be used as inputs in a step of an advanced control method 950. During the step of an advanced control process 950, the calibrated current 948 for each of the switches may be used as input for advanced control processes such as the sensorless field oriented control and the direct torque control. The position and speed control of these advanced control methods may be calculated for each phase or branch based on the calibrated current 948 for each switch.After the calculating step 824 is performed, the hybrid current measurement system 700 may perform the executing step 826. During execution step 826, the controller 702 of FIG. 7 may shift the pulse width modulated signals for each of the six position and speed control switches based on the calibrated currents 948 determined during the previous pulse width modulated signal cycle. Further, during execution step 826, the controller 702 may update the calibration weights 904 for the single active low-side switch, the single active high-side switch, the second determinable low-side switch, and the second determinable high-side switch of the previous pulse width modulated signal cycle.Execution step 826 may be performed at the beginning of T 0. Both the calculating step 824 and the executing step 826 may be performed by a processor in the controller 92. Alternatively, some of the more complex computations may also be performed by a remote processor communicating with the controller 92. To the middle of T0, the hybrid current measurement system 700 again performs high side switch detection 820.It is contemplated that the calculating step 824 may further calculate a temperature value for each of the switches determined in the detected current calculating step 934. The temperature value may be the voltages detected during high side switch detection 820 and low side switch detection 822 divided by a corresponding one of the detected currents determined in the detected current calculation step 934. That is, the voltage 928 of S 6 could be divided by, for example, the sensed current 936 of the single active low side switch to determine the resistance of S 6 corresponding to the temperature, as shown in FIG. 4.The resistance for each of the switches may be determined once the current has been detected with the low-side current detector or the high-side current detector 725 for the respective switch. A resistance threshold may be used to generate a fault when the resistance of the switch exceeds the resistance threshold.Referring now to FIG. 10, which shows a flow diagram for fabricating an embodiment of the hybrid current measurement system. The method may include: coupling a high-side switch to a first phase output and coupled to a high-voltage rail in a package 1002; coupling a low-side switch to a second phase output and to a low-voltage rail in a package 1004; coupling an amplifier to the low-side switch, the amplifier configured to detect first current information as a first voltage drop across the low-side switch in a package 1006; coupling a second amplifier coupled to the high-side switch, the second amplifier configured to detect second current information as a second voltage drop across the high-side switch in a package 1008; coupling a current detector between the low side switch and the low voltage rail, wherein the current detector is configured to determine a first sensed current for the low side switch and the current detector is configured to determine a second sensed current of the high side switch in a housing 1010; and forming a regulator as a current calibrator to calibrate the first current information for the low side switch based on the first sensed current and to calibrate the second current information for the high side switch based on the second sensed current in a housing 1012.Thus, it has been found that the hybrid current measurement system provides important and heretofore unknown and unavailable solutions, capabilities, and functional aspects that allow the current information determined in each pulse width modulated signal cycle to be utilized with advanced control techniques while simultaneously calibrating the current information with the sensed current reading using only a single shunt resistor or the like.The resulting designs are simple, cost effective, uncomplicated, highly versatile, accurate, sensitive, and effective, and can be implemented by adapting known components for rapid, efficient and economical manufacture, application, and use, as fewer current resistances are needed. Furthermore, the space requirement of the system is lower and the monitoring hardware does not have to be extremely fast.Although the hybrid current sensor system has been described in connection with a particular optimal mode, it will be appreciated that many alternatives, modifications and variations will be apparent to those skilled in the art in light of the foregoing description. Accordingly, it is intended to embrace all such alternatives, modifications and variations that fall within the scope of the appended claims. All matter illustrated herein or shown in the accompanying drawings is to be understood in an illustrative and non-limiting sense.Within the scope of the present disclosure, an aspect is directed to a hybrid current measurement system and method, which may include: a switch coupled to a phase output and to a voltage rail; an amplifier coupled to the switch, the amplifier configured to detect current information of the switch; a current detector coupled between the switch and the voltage rail, the current detector configured to determine a sensed current for the switch; and a current calibrator configured to calibrate the current information for the switch based on the sensed current.

Claims

A hybrid current measurement system, comprising: a switch coupled to a phase output and coupled to a high voltage rail; an amplifier coupled to the switch and configured to detect current information of the switch; a current detector coupled between the switch and the voltage rail, the current detector configured to determine a sensed current for the switch; and a current calibrator configured to calibrate the current information for the switch based on the sensed current.The system of claim 1, wherein: the current detector is a current sense amplifier coupled to a shunt resistor.The system of any preceding claim, wherein: the current calibrator is configured to calibrate the current information for an on-resistance of the switch.The system of any preceding claim, wherein: the current calibrator is configured to calibrate the current information for a temperature of the switch.The system of any preceding claim, wherein: an analog-to-digital converter coupled to the amplifier, and wherein the analog-to-digital converter has a digital output coupled to the current calibrator.A hybrid current measurement system, comprising: a high-side switch coupled to a phase output and coupled to a high voltage rail; a low-side switch coupled to a phase output and coupled to a low voltage rail; an amplifier coupled to the low-side switch and configured to detect first current information of the low-side switch; a current detector configured to determine a sensed current for the low-side switch; and a current calibrator configured to calibrate the first current information for the low-side switch based on the sensed current.The system of claim 6, further comprising: a second amplifier coupled to the high side switch, wherein the second amplifier is configured to detect second current information of the high side switch; and wherein: the current calibrator is configured to calibrate the second current information for the high side switch based on the detected current.The system of claim 6 or 7, wherein: the current detector is configured to detect the sensed current during a T1 state of a pulse width modulated signal cycle.The system of any of claims 6 to 8, further comprising: a controller configured to determine a temperature of the low-side switch based on the first current information of the low-side switch and the sensed current.The system of any of claims 6 to 9, wherein: the current calibrator is configured to determine a Kirchhoff-derived current for the high side switch based on the sensed current.A method of manufacturing a hybrid current measurement system, comprising: coupling a switch to a phase output and to a voltage rail; coupling an amplifier to the switch, the amplifier configured to detect a first current information of the switch; forming a current detector to determine a sensed current for the switch; and forming a current calibrator to calibrate the first current information for the switch based on the sensed current.The method of claim 11, wherein: coupling the current detector comprises coupling a current sense amplifier and a shunt resistor between the switch and the voltage rail.The method of claim 11 or 12, wherein: forming the current calibrator comprises forming the current calibrator to calibrate the first current information for an on-resistance of the switch.The method of any of claims 11 to 13, wherein: forming the current calibrator comprises forming the current calibrator to calibrate the first current information for a temperature of the switch.The method of any of claims 11 to 14, further comprising: coupling an analog-to-digital converter to the amplifier and the current calibrator, the analog-to-digital converter having a digital output coupled to the current calibrator.The method of any of claims 11 to 15, wherein: coupling the switch to the voltage rail includes coupling a low-side switch to a low-voltage rail; and further comprising: coupling a high-side switch to the phase output and to a high-voltage rail.The method of claim 16, further comprising: coupling a second amplifier to the high side switch, wherein the second amplifier is configured to detect second current information of the high side switch; and wherein: forming the current calibrator includes forming the current calibrator to calibrate the second current information of the high side switch based on the detected current.The method of claim 16 or 17, wherein: coupling the current detector includes coupling the current detector configured to detect the sensed current during a T1 state of a pulse width modulated signal cycle.The method of any of claims 16 to 18, further comprising: forming a controller to determine a temperature of the low-side switch based on the first current information of the low-side switch and the sensed current.The method of any of claims 16 to 19, wherein: forming the current calibrator includes forming the current calibrator to determine a Kirchhoff-derived current for the high side switch based on the sensed current.

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