Supply voltage generation for a resistive sensor element

Abstract

A circuit arrangement for operating a resistive sensor element is described. According to one embodiment, the circuit arrangement comprises a first supply terminal and a second supply terminal, between which a supply voltage is applied during operation, as well as a sensor circuit with at least one resistive sensor element. The sensor circuit has a first circuit node and a second circuit node for applying a sensor voltage, the first circuit node being connected to the first supply terminal. The circuit arrangement further comprises a sensor supply circuit configured to electrically couple a charged capacitor to the second circuit node during a measurement time interval such that the sensor voltage between the first circuit node and the second circuit node is greater than the supply voltage.

Description

TECHNICAL FIELD

[0001] The present description relates to the field of sensor technology, in particular the generation of a supply voltage for one or more resistive sensor elements. BACKGROUND

[0002] There are a variety of sensors with resistive sensor elements whose electrical resistance depends directly or indirectly on the physical quantity to be measured. One example of this is TC sensors (thermal conductivity sensors) for measuring the thermal conductivity of gases. With this type of sensor, a resistor (sensor element) is heated by applying a voltage. The resistor is exposed to an analysis gas. Due to the voltage applied to the sensor element, a sensor current flows, which results in a certain power loss in the sensor element. Due to this power loss, the sensor element (e.g. a resistor) heats up to a certain temperature above the ambient temperature. At this temperature, the total heat loss of the resistor corresponds to the electrical energy consumed. The heat loss depends to a large extent on the thermal conductivity of the analysis gas.The higher the thermal conductivity of the gas, the greater the cooling effect and the lower the temperature (and the greater the electrical conductivity) of the sensor element at a constant voltage. The greater the electrical conductivity and resistance of the sensor element, the higher the current. The current is therefore a measure of the thermal conductivity of the analysis gas.

[0003] The sensitivity of such a TC sensor increases with the supply voltage. There is an approximately cubic dependence of the sensor's output signal on the applied supply voltage. The sensor's supply voltage is not applied continuously, but rather at regular intervals (periodically or from time to time) to reduce average power consumption and increase the sensor's service life.

[0004] In many applications, the operation of a sensor is controlled by an integrated control circuit, for example, a microcontroller. Such a control circuit—like many other integrated circuits—can operate with a relatively low supply voltage, for example, 3.3 volts. However, this voltage may be too low for resistive sensor elements, especially the TC sensors mentioned above, to achieve sufficient sensor sensitivity.

[0005] DE 10 2011 002 937 A1 discloses a particle sensor comprising an integrated amplifier circuit with a voltage converter for generating an amplified reference voltage.

[0006] US 2015 / 0 316 586 A1 discloses a sensor system comprising one or more charge pumps for providing a high voltage.

[0007] DE 102017 106 249 A1 describes the testing of a capacitive sensor, whereby a bias voltage is applied to the sensor via a high-impedance component. A path for applying the test signal excludes the high-impedance component.

[0008] The inventors have set themselves the task of improving existing concepts for the operation of resistive sensor elements, in particular with regard to the generation of the sensor supply voltage. SUMMARY

[0009] The above-mentioned object is achieved by the circuit arrangement according to claim 1 and by the method according to claim 11. Various embodiments and further developments are the subject of the dependent patent claims.

[0010] The following describes a circuit arrangement for operating a resistive sensor element. According to one exemplary embodiment, the circuit arrangement comprises a first supply terminal and a second supply terminal, between which a supply voltage is applied during operation, and a sensor circuit with at least one resistive sensor element. The sensor circuit has a first circuit node and a second circuit node for applying a sensor voltage, wherein the first circuit node is connected to the first supply terminal. The circuit arrangement further comprises a sensor supply circuit which is designed to electrically couple a charged capacitor to the second circuit node during a measuring time interval such that the sensor voltage between the first circuit node and the second circuit node is greater than the supply voltage.During the measuring time interval, the charged capacitor is connected between the second supply terminal and the second circuit node of the sensor circuit in such a way that the sensor voltage essentially corresponds to the sum of the supply voltage and the capacitor voltage across the precharged capacitor.

[0011] Furthermore, a method for operating a sensor circuit is described, which has at least one resistive sensor element, wherein the sensor circuit has a first circuit node and a second circuit node for applying a sensor voltage. According to one embodiment, the method comprises charging a capacitor by coupling it to a voltage supply that provides a supply voltage between a first supply terminal and a second supply terminal, wherein the first circuit node of the sensor circuit is connected to the first supply terminal. The method further comprises coupling the charged capacitor to the second circuit node of the sensor circuit during a measuring time interval such that the resulting sensor voltage between the first circuit node and the second circuit node of the sensor circuit is greater than the supply voltage.During the measuring time interval, the charged capacitor is connected between the second supply terminal and the second circuit node of the sensor circuit in such a way that the sensor voltage essentially corresponds to the sum of the supply voltage and the capacitor voltage across the precharged capacitor. BRIEF DESCRIPTION OF THE ILLUSTRATIONS

[0012] Below, exemplary embodiments are explained in more detail using illustrations. The illustrations are not necessarily to scale, and the exemplary embodiments are not limited to the aspects shown. Rather, emphasis is placed on illustrating the principles underlying the exemplary embodiments. Regarding the illustrations: Fig. 1 illustrates the structure of a sensor element of a TC sensor using micromechanical structures. Fig. Figure 2 shows a bridge circuit (resistance bridge) that can be used in a TC sensor. Fig. Figure 3 is a diagram illustrating the dependence of sensor sensitivity on the sensor voltage of the sensor. Fig. 4 shows, as a first embodiment, a circuit arrangement with a supply circuit for generating a sensor voltage for a sensor with resistive sensor elements. Fig. 5 contains several timing diagrams to illustrate the function of the circuit arrangement of Fig. 4. Fig. 6 shows another embodiment of the supply formwork from Fig. 4 in more detail. Fig. 7 shows an alternative (equivalent) implementation of the example from Fig. 4. Fig. 8 and Fig. 9 show extensions / modifications of the example from Fig. 4, where in addition to the bridge output voltage, the sensor voltage across the bridge is also measured. Fig. 10 is a diagram illustrating a measurement process. DETAILED DESCRIPTION

[0013] Fig. Figure 1 shows a micromechanical structure of a resistive sensor element of a TC sensor. The micromechanical structure can be integrated into a silicon chip, which in turn is arranged in a chip package. Various methods for manufacturing integrated microelectromechanical systems (MEMS) are known per se and will therefore not be discussed further here.

[0014] According to the Fig. In the example shown in Figure 1, the micromechanical structure 10 forms a frame, which may, for example, have a rectangular shape. The shape of the frame, however, is not important. An electrical conductor 20, which may, for example, be designed as a strip conductor, runs across the frame (like a "bridge" from one side of the frame to an opposite side). The conductor 20 may be made of metal (e.g., aluminum) or polysilicon. In the case of a metallic conductor, it may be arranged on a silicon web that connects two opposite sides of the frame 10.

[0015] The micromechanical structure 10 forms a cavity in which the analysis gas mentioned above can be located. In practice, the analysis gas is often a mixture of several gaseous substances, such as air. Fig. 1 shows a voltage source Q S , which determines the sensor voltage V SThe resulting sensor current i S depends on the resistance R S (electrical conductivity σ S =R S -1 ) of the sensor element. The power P converted into heat in the sensor element S is equal to V S 2 ·σ S = i S 2 ·R S . In thermal equilibrium (ie at constant temperature of the conductor 20), the heat removed per unit of time (cooling power) is equal to the electrical power V S 2 ·σ S The cooling capacity increases with the thermal conductivity κ of the gas being analyzed. Therefore, the temperature T S of the conductor 20 should be smaller, the greater the thermal conductivity κ of the analysis gas. A lower temperature T S of the conductor 20 (because the temperature coefficient of resistance R S is positive) a greater electrical conductivity σ Sas a result, i.e. σ S = σ S (κ) is a function of the thermal conductivity κ. The sensor current i S therefore depends on the thermal conductivity κ, because i S = V S ·σ S (κ). A higher thermal conductivity κ results in a lower temperature T S and consequently in a higher electrical conductivity σ S and a higher sensor current i S .

[0016] As with many other resistive sensors, the resistive sensor elements can be Fig. 1 to form a bridge circuit (also called a Wheatstone bridge). An example is shown in Fig. 2. Accordingly, the bridge circuit comprises two voltage dividers connected between a first bridge supply node (where the supply voltage V SUPPLY is applied) and a second bridge supply node (to e.g. a reference or ground potential V REFTwo of the four bridge resistors are sensor elements according to Fig. 1 and with R S1 and R S2 The other two resistors are reference resistors R REF1 and R REF2 . Under the assumptions R REF1 = R REF2 = R and R S1 = R S2 = R+ΔR and with the approximation 2R+ΔR ≈ 2R (because ΔR << R) the bridge output voltage V OUT = (V SUPPLY -V REF )·ΔR / 2, where ΔR is the deviation of the sensor resistance from the known, nominal resistance value R. In many known sensor circuits, the reference potential V REF equal to the ground potential, namely 0 volts.

[0017] The diagram from Fig. 3 illustrates the dependence of the bridge output voltage V OUT from the bridge supply voltage V SUPPLY (assuming V REF=0V). The two curves shown relate to different gaseous mixtures, labeled "Gas A" and "Gas B." It can be seen that the bridge output voltage V OUT increases disproportionately with the bridge supply voltage V SUPPLY because at a higher bridge supply voltage V SUPPLY also the current through the sensor elements and consequently their temperature T S increases. When the voltage V SUPPLY by two thirds (e.g. from 3V by about 67% to 5V) the bridge output voltage V OUT by significantly more than 67%, in the examples shown by several hundred percent.

[0018] In addition to the sensor elements, sensors usually also include circuits for controlling the sensor elements and for (post-)processing the sensor signal (e.g. the bridge output voltage V OUT). These tasks can be performed by control circuits, which may, for example, include a microcontroller. The control circuit (the microcontroller) may, for example, have an analog-to-digital converter designed to measure the output voltage of a sensor element or the bridge output voltage V OUT a resistance bridge (cf. Fig. 2) to digitize. Modern digital circuits such as microcontrollers operate with relatively low supply voltages such as 3.3 volts. As explained above with reference to Fig. However, as explained in section 3, a higher supply voltage is advantageous for resistive sensors (where the sensor elements are arranged, for example, in a resistance bridge) and especially for TC sensors.

[0019] For these reasons, two different supply voltages are required in conventional sensor designs. For example, a 6V supply voltage applied to the resistor bridge is used to generate a 3.3V supply voltage for the microcontroller using a linear voltage regulator, which results in undesirable losses in the voltage regulator. Instead of a voltage regulator, a DC / DC switching converter can also be used, for example, to generate a 3.3V supply voltage from a 6V supply voltage or vice versa. However, such DC / DC switching converters, such as boost converters or buck converters, are complex and require an inductor that cannot be integrated into a chip.

[0020] The Fig. The circuit arrangement shown in Figure 4 comprises a resistance bridge 30, a control circuit 40 (microcontroller) with an analog-to-digital converter for digitizing the bridge output voltage V OUT and a supply circuit configured to temporarily (during a measurement time interval) supply the bridge circuit 30 with a supply voltage V SUPPLY -V REF which is greater than the supply voltage V SUPPLY , which also supplies the control circuit 40.

[0021] In the Fig. In the example shown in Figure 4, the supply voltage is 3 volts (V SUPPLY =3V). The supply voltage V SUPPLY is supplied to the control circuit 40 and the first bridge supply node N1. The second bridge supply node N2 is at a reference voltage V N2 =V REF, which in the examples described here is not constant, but is provided by a supply circuit 50. The operation of the supply circuit is controlled by the control circuit 40 (the microcontroller), wherein the control circuit 40 is designed to temporarily (during a measurement time interval) generate a voltage V N2 to the bridge supply node N2, which is below the operating voltage range of 0-3V. That is, the voltage V N2 is negative during the measuring time interval and the bridge supply voltage V SUPPLY -V N2 is therefore greater than the supply voltage V SUPPLY . The operation of the supply circuit 50 is explained below using the timing diagrams from Fig. 5 is explained in more detail.

[0022] The supply circuit 50 receives a logic signal V as a control signal IO, which is generated by the microcontroller 40. Conventional microcontrollers have digital outputs to which a logic signal can be output. Some microcontrollers have configurable connections (so-called General Purpose IO (GPIO) terminals) that can be configured as both digital inputs and digital outputs. Such GPIO connections can also be used to output the logic signal V IO The logic signal V IO can only assume two different levels, namely a low level which is approximately equal to the ground potential V GND =0V, and a high level that corresponds approximately to the supply voltage V SUPPLYIn the example shown, the supply circuit 50 has a capacitor C1, which is connected on the one hand to a digital output of the microcontroller 40 (e.g., by means of a resistor R1 for current limiting) and on the other hand to the circuit node N3. This node N3 can in turn be connected to the ground node GND by means of a diode D1, whereby the diode D1 can also be replaced by another component such as a controllable electronic switch (e.g., a transistor or a more complex network). The node N3 is connected to the second bridge supply node N2 by means of a further electronic switch T1 (a MOS transistor in the example shown). The transistor T1 can also be replaced by other types of electronic switches or more complex switching networks.

[0023] The microcontroller can determine the mentioned measuring time interval by the level of the logic signal V IOAs shown in diagram (a) of the Fig. 5, the measuring time interval (from time t1 to time t2) is defined by a low level (approx. 0V) of the logic signal V IO Outside the measurement time interval, the logic signal has a high level (approx. 3V in the example shown). As long as the logic signal V IO has a high level, a current i C from the digital output GPIO of the microcontroller 40 through the resistor R1, the capacitor C1, and the diode D1 to the ground node GND and charge the capacitor C1. The charging process ends when the capacitor voltage Vc has reached a steady-state value, which essentially corresponds to the high level of the logic signal minus the forward voltage V F (approx. 0.7V) of the diode D1 (ie V C ≈ V SUPPLY -V F ). Immediately before time t1, the capacitor voltage Vc is practically constant and the voltage V N3is approximately equal to the voltage drop V F across the diode D1 (V N3 = V F ≈ 0.7V, see diagram (b) of the Fig. 5). When the microcontroller 40 detects the level of the logic signal V IO at time t1 (ie at the beginning of a measuring time interval) from high to low, then the voltage V N3 of node N3 abruptly from V F to a negative value of approximately -V SUPPLY +V F , i.e. approximately -2.3 volts in the current example, and the diode D1 is blocked. At the same time (also at time t1), the electronic switch / transistor T1 turns on and connects the node N3 to the bridge supply node N2, resulting in a voltage V SUPPLY- V N3 at the bridge circuit 30, which is greater than the supply voltage V SUPPLY (initially approx. 2 V SUPPLY -V F ). The voltage drop across transistor T1 is neglected for simplicity.

[0024] During the measurement time interval (between t1 and t2), a current flows through the resistor bridge 30 while the capacitor C1 is discharged. Consequently, the voltage V N3 during the measurement time interval and the bridge supply voltage V SUPPLY- V N3 decreases accordingly (see diagram (c) of the Fig. 5). Depending on the capacitance of capacitor C1, the voltage change ΔV during the measurement time interval can be smaller or larger. At the end of the measurement time interval (at time t2), microcontroller 40 sets the level of the logic signal V IO from Low to High and the capacitor is recharged relatively quickly (in Fig. 5 (not shown in detail). The diode D1 becomes conductive again, the electronic switch (transistor T1) switches off and the voltage V SUPPLY- V N2across the bridge circuit 30 drops to zero. During the measurement time interval, the microcontroller 40 can monitor the course of the bridge output voltage V OUT scan and digitize (see also Fig. 10). From the corresponding digital values, the microcontroller 40 can determine the desired information (e.g., the thermal conductivity of the analysis gas in the case of a TC sensor). For this purpose, the microcontroller can use calibration data and / or characteristic curves stored in a memory.

[0025] The Fig. The circuit shown in Figure 4 is merely an example. In general, the embodiments described here relate to a circuit arrangement for operating a sensor. In a general example, the circuit arrangement is controlled by means of a supply voltage V SUPPLYFor this purpose, the circuit arrangement has a first supply terminal and a second supply terminal GND between which, during operation, the supply voltage V SUPPLY The circuit arrangement further comprises a sensor circuit 30 with at least one resistive sensor element (cf. Fig. 2, resistors R S1 , R S2 ). The sensor circuit 30 (e.g. a Wheatstone bridge) is supplied with a sensor voltage V S The sensor circuit 30 therefore has a first circuit node N1 and a second circuit node N2, between which the sensor voltage V S The first circuit node N1 of the sensor circuit is connected to the first supply terminal (see Fig. 4). The circuit arrangement further comprises a sensor supply circuit 50. This is designed to electrically couple a precharged capacitor C1 to the second circuit node N2 during a measurement (during a measurement time interval) in such a way (in the illustrated examples by means of the transistor T1) that the sensor voltage V S (which is applied between the first circuit node N1 and the second circuit node N2) is greater than the supply voltage V SUPPLY . That is, the capacitor voltage Vc of the charged capacitor C1 is equal to the supply voltage V SUPPLY superimposed (added to it). The sensor voltage VS is thus - at least temporarily - approximately equal to V SUPPLY +V C .

[0026] Fig. Figure 6 shows another example of the supply circuit 50 and also illustrates the operation of the microcontroller's digital output GPIO in more detail. Fig. 6, the microcontroller 40 includes a driver that generates the logic signal V IO at the digital output GPIO. In the example shown, the driver comprises two electronic switches, a high-side switch and a low-side switch. The high-side switch couples the GPIO output (e.g., an output pin of the microcontroller 40) to the supply voltage V SUPPLY , and the low-side switch couples the GPIO output to the ground potential V GND at the second supply node GND. The high-side switch is controlled by the Boolean signal CTL, and the low-side switch is controlled by the inverse signal CTL. In practice, the driver's output stage can be implemented, for example, as a CMOS inverter. The CTL signal can be represented, for example, by a bit of a register that a processor of the microcontroller 40 can set (to "1") and reset (to "0"). That is, the signal V IOhas either a high level of approximately V SUPPLY or a low level approximately equal to V GND corresponds.

[0027] Furthermore, the example from Fig. 6 of the supply circuit 50 from Fig. 4, where diode D1 has been replaced by a second transistor T2, whose control electrode (gate) is also connected to the digital output GIPO. The control electrode of transistor T1 is connected to the ground node GND. If no measurement is taking place, the microcontroller 40 outputs a high level as logic signal V IO off. This is why transistor T2 turns on and transistor T1 turns off (because the gate-source voltage is approximately zero). The voltage drop across the series circuit R1-C1 is therefore approximately equal to the high level of the signal V IO (ie V IO ≈V SUPPLY ) and the capacitor C1 is charged until the capacitor voltage Vc equals the high level of the logic signal VIO corresponds (ie V C ≈V IO ≈V SUPPLY ). Resistor R1 is optional and serves only to limit the current.

[0028] During a measurement (in the measurement time interval), the microcontroller 40 outputs the logic signal V IO a low level (around V GND =0V). As a result, transistor T2 turns off and the electrical potential in node N3 drops from approximately 0V to a value of approximately V N3 = -Vc. In the example from Fig. 4, diode D1 is blocking due to the negative potential at node N3. The negative electrical potential V N3 in node N3 also leads to switching on of transistor T1 (positive gate-source voltage at T1), which is why the second circuit node N2 of the sensor circuit 30 (bridge circuit) also has the electrical potential V N2 = V N3 = -Vc, which means that the sensor voltage V Sacross the resistance bridge 30 approximately the sum V SUPPLY + Vc corresponds to (V S = V SUPPLY - V N2 = V SUPPLY - (-Vc) = V SUPPLY + Vc). During the measurement, the sensor voltage V S i.e. greater than the supply voltage V SUPPLY , which, as explained at the beginning (cf. Fig. 3), which can result in higher sensitivity. During the measurement, the capacitor C1 discharges and the capacitor voltage Vc decreases. The sensor voltage V S (Bridge voltage) is - as in diagram (c) the Fig. 5 - not constant during a measurement.

[0029] Fig. 7 shows an alternative (but functionally equivalent) implementation of the example from Fig. 4. It is also said that the circuit Fig. 7 is complementary to the circuit of Fig. 4. Sensor circuit 30 and supply circuit 50 are reversed, so to speak, with the polarity of diodes being changed, logic signals inverted, and n-channel transistors being replaced by p-channel transistors (and vice versa). The function of the circuit arrangement remains unchanged. To charge capacitor C1, the logic signal V IO now has a low level. Diode D1 is therefore conductive, the (p-channel) transistor T1 is off, and current can flow from the voltage supply through capacitor C1 and resistor R1 to the digital output GPIO, charging the capacitor. For a measurement, the microcontroller 40 outputs the logic signal V IO a high level. This increases the electrical potential V N3 in node N3 via the supply voltage V SUPPLY raised beyond (V N3 =V SUPPLY +V C ). As a result, the diode D1 is blocked and the transistor T1 is switched on, which is why the electrical potential VN2 at node N2 increases (V N2 ≈V N3 =V SUPPLY +V C ). The resistance bridge 30 “sees” a sensor voltage V S =V N2 -V N1 =V N2 which is greater than the supply voltage V SUPPLY (analogous to the examples from the Fig. 4 and Fig. 6).

[0030] Fig. 8 and Fig. 9 show extensions / modifications of the example from Fig. 4, where in addition to the bridge output voltage V OUT also indirectly the sensor voltage V S above the bridge. In the Fig. In the example shown in Figure 7, a voltage divider is connected in parallel to the sensor element 30 (Wheatstone bridge), which divides the resistors R A and R B The voltage across the resistor R A is equal to V S ·R A / (R A +R B ). If both resistances are equal (R A =R B), then the voltage across the resistor R A equal to V S / 2. This voltage can be fed to a differential amplifier 32 (e.g. with a gain of one) and the output signal of the amplifier 32 can be fed to the analog input of an analog-to-digital converter contained in the microcontroller 40, which converts the voltage V S ·R A / (R A +R B ) digitized (sampled and quantized). The corresponding digital values ​​represent the sensor voltage V S during the measurement time interval. Incidentally, the example from Fig. 8 same as the example from Fig. 4 and it will refer to the corresponding description above.

[0031] The example from Fig. 9 is very similar to the example from Fig. 7, but instead of the voltage divider (resistors R A and R B ) a part of the resistance bridge is used. Under the assumptions R S1=R+ΔR and R REF1 =R and the approximation ΔR / R<<1 (ΔR is negligibly small compared to R) the differential amplifier 32 “looks” like in the example of the Fig. 7 half the sensor voltage V S / 2, if the amplifier 32 receives the voltage across the resistor R S1 supplied (see Fig. 7 in conjunction with Fig. 2). Furthermore, the example from Fig. 9 same as the example from Fig. 8.

[0032] Fig. 10 shows a timing diagram of a measurement time interval (analogous to Fig. 5) The upper diagram shows the sensor voltage V S (V S =V SUPPLY -V N2 in the example from Fig. 4) and the lower diagram the corresponding bridge output voltage V OUT as well as the corresponding digital values ​​(samples) that the control circuit 40 (e.g., microcontroller) generates by means of an analog-to-digital converter. Since the capacitor C1 (see, e.g., Fig. 4 or Fig. 7) during a measurement (which in the example from Fig. 10 10ms) discharges, the sensor voltage V S and consequently also the bridge output voltage V OUT not constant. In a measuring time interval (measuring cycle), a large number of individual measured values ​​V OUT at different sensor voltages V S From the (digitized) measured values, the microcontroller 40 can determine the desired information (e.g., the thermal conductivity of the analysis gas in the case of a TC sensor). For this purpose, the microcontroller can use calibration data and / or characteristic curves stored in a memory and / or digitally post-process the measured values. The acquisition of a large number of measured values ​​at different sensor voltages V S also allows the compensation of cross-sensitivities with regard to other physical parameters (e.g. humidity) which affect the output voltage V OUTcan influence.

[0033] In the illustrated embodiments, the control circuit 40 is a microcontroller with at least one processor, which is designed to execute software instructions stored in a memory in order to perform the functions described here, such as controlling the measurement time intervals (measurement cycles) by outputting the signal V IO at a digital output and controlling the operation of the analog-to-digital converter (see e.g. Fig. 4). Many microcontrollers have one or more analog-to-digital converters integrated. However, separate analog-to-digital converters can also be used. In the examples from Fig. 8 and Fig. 9, instead of two analog-to-digital converters, a multiplexer and a single analog-to-digital converter can also be used, with two analog measured values ​​being digitized sequentially by means of the multiplexer and the analog-to-digital converter. However, the control circuit 40 does not necessarily have to include a processor for executing software instructions, but can also include a one-time programmable (OTP) or hard-wired digital circuit that provides essentially the same function.

[0034] The functionality of the embodiments described above implements a method for operating a sensor circuit that has at least one resistive sensor element and a first circuit node and a second circuit node for applying a sensor voltage. According to the examples described here, the method includes charging a capacitor (see, for example, Fig. 4 or Fig. 6, capacitor C1) by coupling it to a voltage supply which provides a supply voltage between a first supply terminal and a second supply terminal (cf. Fig. 4, supply terminals SUP and GND), wherein the first circuit node of the sensor circuit is connected to the first supply terminal. The method further comprises coupling the charged capacitor to the second circuit node of the sensor circuit during a measurement time interval such that the resulting sensor voltage between the first circuit node and the second circuit node of the sensor circuit is greater than the supply voltage (cf. Fig. 5, V S is greater than V SUPPLY because V N3 becomes negative).

[0035] According to one embodiment, an output voltage of the sensor circuit can be measured during the measuring time interval (e.g. by means of an analog-to-digital converter, see e.g. Fig. 4 and 7-9). The measurement time interval can be indicated by a logic signal generated, for example, by a control circuit (e.g., a microcontroller). According to the embodiments depicted here, the at least one resistive sensor element can be part of a Wheatstone bridge. However, the concepts described here are not dependent on a resistive bridge and can also be applied to other resistive sensors. In some embodiments, the sensor voltage can also be measured during the measurement time interval, whereby the capacitor discharges during the measurement time interval and consequently the sensor voltage decreases.

[0036] By digitizing (sampling and quantizing) both the output voltage of the sensor circuit and the sensor voltage supplying the sensor circuit during the measurement time interval, a wide range of measured values ​​can be acquired at different sensor voltages. Especially with TC sensors, this enables digital post-processing of the measured values ​​to compensate for cross-sensitivities, such as the cross-sensitivity of the sensor element to moisture.

Claims

[1] A circuit arrangement comprising: a first supply terminal and a second supply terminal (GND), between which a supply voltage (V SUPPLY ) is present; a sensor circuit (30) with at least one resistive sensor element (R S1 , R S2 ), wherein the sensor circuit (30) has a first circuit node (N1) and a second circuit node (N2) for applying a sensor voltage (V S ), and wherein the first circuit node (N1) is connected to the first supply terminal; and a sensor supply circuit (50) configured to electrically couple a charged capacitor (C1) to the second circuit node (N2) during a measurement time interval such that the sensor voltage (V S ) between the first circuit node (N1) and the second circuit node (N2) is greater than the supply voltage (V SUPPLY ), wherein the charged capacitor (C1) is connected between the second supply terminal (GND) and the second circuit node (N2) of the sensor circuit (30) during the measuring time interval such that the sensor voltage (V S ) essentially the sum of supply voltage (V SUPPLY ) and the capacitor voltage (Vc) across the precharged capacitor (C1). [2] The circuit arrangement according to claim 1, wherein the sensor supply circuit (50) comprises an electronic switch (T1) configured to switch the charged capacitor (C1) on the basis of a logic signal (V IO ), which indicates the measurement time interval, to the second circuit node (N2). [3] The circuit arrangement according to claim 2, wherein the sensor supply circuit (50) comprises a further electronic switch (D1, T1) which is designed to separate the charged capacitor (C1) from the second supply terminal (GND) during the measuring time interval. [4] The circuit arrangement according to one of claims 1 to 3, wherein the at least one resistive sensor element (R S1 , R S2 ) of the sensor circuit (30) is part of a Wheatstone bridge which is connected to the sensor voltage (V S ) is supplied. [5] The circuit arrangement according to one of claims 1 to 3, wherein the at least one resistive sensor element (R S1 , R S2 ) is a microelectromechanical system (MEMS) integrated into a chip, whose electrical resistance depends on the thermal conductivity of an analysis gas surrounding the MEMS. [6] The circuit arrangement according to one of claims 1 to 5, further comprising: a control circuit (40) comprising at least one analog-digital converter which is designed to digitize an output voltage of the sensor circuit (30) during the measuring time interval. [7] The circuit arrangement according to claim 6, wherein the control circuit (40) is further configured to, during the measuring time interval, generate an output voltage of the sensor circuit (30) which is proportional to the sensor voltage (V S ) proportional voltage (V S / 2) to digitize. [8] The circuit arrangement according to any one of claims 2 and 3 to 5, as far as dependent on claim 2, further comprising: a control circuit (40) which is designed to output the logic signal (V IO ), whereby the level of the logic signal (V IO ) shows the measurement time interval. [9] The circuit arrangement according to claim 7 or 8, wherein the control circuit is connected to the supply voltage (V SUPPLY ) is supplied. [10] A method for operating a sensor circuit comprising at least one resistive sensor element (R S1 , R S2 ), wherein the sensor circuit (30) has a first circuit node (N1) and a second circuit node (N2) for applying a sensor voltage (V S ); the method comprises: Charging a capacitor (C1) by coupling the capacitor (C1) to a voltage supply having a supply voltage (V SUPPLY ), wherein the first circuit node (N1) of the sensor circuit (30) is connected to the first supply terminal; and Coupling the charged capacitor (C1) to the second circuit node (N2) of the sensor circuit (30) during a measuring time interval such that the resulting sensor voltage (V S ) between the first circuit node (N1) and the second circuit node (N2) is greater than the supply voltage (V SUPPLY ); wherein the charged capacitor (C1) is connected between the second supply terminal (GND) and the second circuit node (N2) of the sensor circuit (30) during the measuring time interval such that the sensor voltage (V S ) essentially the sum of supply voltage (V SUPPLY ) and the capacitor voltage (Vc) across the precharged capacitor (C1). [11] The method according to claim 10, further comprising: Measuring an output voltage (V OUT ) of the sensor circuit during the measuring time interval. [12] The method benefited from claim 11, which further comprises: Generating a logic signal (V IO ), which indicates the measurement time interval. [13] The method according to any one of claims 10 to 12, wherein the at least one resistive sensor element (R S1 , R S2 ) is part of a Wheatstone bridge. [14] The method according to any one of claims 10 to 13, further comprising: Measuring the sensor voltage (V S ) during the measuring time interval, whereby the capacitor (C1) discharges during the measuring time interval and consequently the sensor voltage (V S ) reduced.

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