Bridgeless hybrid limited-slip converter and operation

Abstract

A device, such as a power converter as discussed herein, may comprise the following features: a first winding; a first bidirectional switching circuit arrangement; and a second bidirectional switching circuit arrangement arranged in series with the first bidirectional switching circuit arrangement. A combination of the first bidirectional switching circuit arrangement and the second bidirectional switching circuit arrangement may be configured to control the magnitude of a current through the first winding to produce an output voltage.

Description

[0001] As the name suggests, a conventional power converter transforms a received input voltage into an output voltage. One type of conventional power converter receives an alternating current (AC) voltage and converts it into a corresponding direct current (DC) output voltage.

[0002] The object of the present invention is to create a device and a method with improved properties.

[0003] This problem is solved by a device according to claim 1 and a method according to claim 18.

[0004] A device, such as a power supply or other suitable entity as discussed herein, may be configured to include the following features: a first winding; a first bidirectional switching circuit arrangement; and a second bidirectional switching circuit arrangement arranged in series with the first bidirectional switching circuit arrangement. A combination of the first bidirectional switching circuit arrangement and the second bidirectional switching circuit arrangement may be configured to control the magnitude of a current through the first winding to produce an output voltage.

[0005] As further discussed herein, the device may also include a capacitor arranged in a series circuit path that includes the first winding. The series circuit path can be effective in promoting resonance of the current through / to the first winding. In particular, the series circuit path that includes the capacitor arranged in series with the first winding can be a resonant circuit.

[0006] It should also be noted that the first winding can be arranged in a transformer. A second winding of the transformer can be magnetically coupled to the first winding. In such a case, the flow of the (first) current through the first winding induces a (second) current to flow through the second winding. The transformer windings support the conversion of a received input voltage into an output voltage, as well as the output voltage from the second winding.

[0007] Furthermore, as discussed herein, the first bidirectional switching arrangement can be effective in blocking the passage of current / voltage in both a first direction and a second direction through the first bidirectional switching arrangement; the second bidirectional switching arrangement can be effective in blocking the passage of current / voltage in both a first direction and a second direction through the second bidirectional switching arrangement.

[0008] According to further examples, as discussed herein, the first bidirectional switching circuit arrangement and the second bidirectional switching circuit arrangement can be implemented in any suitable way. In one example, the first bidirectional switching circuit arrangement can be a first GaN (gallium nitride) switch; the second bidirectional switching circuit arrangement can be a second GaN (gallium nitride) switch.

[0009] Furthermore, the first bidirectional switch circuit arrangement can comprise a first switch arranged in series with a second switch; the second bidirectional switch circuit arrangement can comprise a third switch arranged in series with a fourth switch; a combination of the first switch and the second switch can be arranged in a first series circuit path between a first node and a second node of the power converter; a combination of the third switch and the fourth switch is arranged in a second series circuit path between the second node and a third node of the power converter.

[0010] Additionally, as discussed herein, the device can be configured to include a control effective to: i) during a first mode in which an input voltage between the first node and the third node is positive, activate both the first switch and the third switch to ON states, and ii) alternately switch the second switch and the fourth switch to an ON state to control the magnitude of the current through the first winding. The control can further be effective to: during a second mode in which the input voltage between the first node and the third node is negative, activate both the second switch and the fourth switch to ON states, and ii) alternately switch the first switch and the third switch to an ON state to control the magnitude of the current through the first winding.

[0011] Furthermore, the device, such as a circuit arrangement as discussed herein, can be configured to include a transformer comprising a first winding and a second winding. The second winding can be magnetically coupled to the first winding within the transformer. In such a case, a current flowing through the first winding effectively generates the output voltage, which is then output by the second winding. The control can be configured to regulate the magnitude of the output voltage by switching between operation in the first mode and the second mode, and by implementing switch control in each of the first and second modes.

[0012] Further examples, as discussed herein, include a configuration in which the first bidirectional switching circuit arrangement may be a first dual-gate switch comprising a first drain node, a first source node, a first gate node, and a second gate node. The second switching circuit arrangement may be a second dual-gate switch comprising a second drain node, a second source node, a third gate node, and a fourth gate node.

[0013] Another example, as discussed herein, includes a controller. The controller can be configured to: during a first mode in which an input voltage between the first node and the third node is positive: i) apply ON control signals to the first gate node and the third gate node, and ii) toggle between applying an ON control signal to the second gate node and the fourth gate node to control the magnitude of the current through the first winding. Additionally or alternatively, the controller can further be configured to: during a second mode in which an input voltage between the first node and the third node is negative: i) apply ON control signals to the second gate node and the fourth gate node, and ii) toggle between applying an ON control signal to the first gate node and the third gate node to control the magnitude of the current through the first winding.

[0014] In another example, the control is effective in regulating the output voltage by switching between operation in the first mode and the second mode.

[0015] Further examples, as discussed herein, include the device further comprising: a first capacitor arranged in parallel to a combination of the first bidirectional switch circuit arrangement and the first winding; and a second capacitor arranged in parallel to a combination of the second bidirectional switch circuit arrangement and the first winding.

[0016] The device, as discussed herein, may further comprise a first capacitor. The first bidirectional switching circuit arrangement may be directly connected to the second bidirectional switching circuit arrangement via a first node, the first capacitor being arranged in series between the first node and the first winding. The device may further comprise a control effective for controlling the first bidirectional switching circuit arrangement and the second bidirectional switching circuit arrangement based on a feedback signal generated by a second node directly connecting the first capacitor and the first winding.

[0017] Other examples, as discussed herein, include a method comprising: controlling the operation of the first bidirectional switching circuit arrangement; controlling the operation of the second bidirectional switching circuit arrangement, which is arranged in series with the first bidirectional switching circuit arrangement; and wherein the controlled operation of the first bidirectional switching circuit arrangement and the second bidirectional switching circuit arrangement controls a magnitude of a current through the first winding to produce an output voltage.

[0018] As previously discussed, the first bidirectional switch circuit arrangement can be configured to include a first switch in series with a second switch; the second bidirectional switch circuit arrangement can be configured to include a third switch in series with a fourth switch. A combination of the first and second switches can be arranged in a first series path between a first node and a second node. A combination of the third and fourth switches can be arranged in a second series path between a second node and a third node.In such a case, the method may further include, via a control, the following: during a first mode in which an input voltage between the first node and the third node is positive: i) activating both the first switch and the third switch to ON states, and ii) alternating between activating the second switch and the fourth switch to an ON state in order to control the magnitude of the current through the first winding. For example, in the first mode, when the first switch is ON, the second switch turns off; and when the first switch is off, the second switch is ON.

[0019] Further examples, as discussed herein, concerning control during a second mode in which the input voltage between the first and third nodes is negative, include: i) activating both the second and fourth switches to ON states, and ii) alternating between activating the first and third switches to an ON state to control the magnitude of the current through the first winding. For example, in the second mode, when the first switch is ON, the second switch is OFF; and when the first switch is OFF, the second switch is ON.

[0020] These and other more specific examples are revealed in more detail below.

[0021] Other examples, as discussed herein, include a device having the following features: a first transformer winding; a sensing circuit arrangement effective in sensing a first energy supplied to the first transformer winding by an input voltage; and a switching circuit arrangement effective in applying a power factor correction associated with converting the input voltage into an output voltage derived from an output of a second transformer magnetically coupled to the first transformer winding, the applied power factor correction being effective in controlling a flow of the first energy from the input voltage to the first transformer winding.

[0022] The output voltage can supply a second energy to a respective load. The device can further include a control system effective in: i) monitoring a quantity of the first energy via feedback from the sensing circuit arrangement, and ii) applying power factor correction via controlled operation of the switching circuit arrangement, such that an average quantity of the first energy supplied by the input voltage to the first transformer winding can be substantially equal to an average quantity of the second energy supplied by the second transformer winding to the respective load.

[0023] The detection circuit arrangement can further comprise a first capacitor. The device can also include a series circuit path comprising the first capacitor, which is coupled in series with the first transformer winding. The device can further include a control mechanism that effectively controls the flow of a resonant current through the series circuit path via a control of the switching circuit arrangement, the flow of the resonant current being controlled based on the applied power factor correction.

[0024] In further examples, as discussed herein, the sensing circuit arrangement may be configured to include a capacitor effective in sensing a quantity of the first energy supplied by the input voltage through the switching circuit arrangement to the first transformer winding. The output voltage may be derived from the output of the second transformer winding, which may be effective in supplying a second energy to a respective load.The device may further include a control that is effective in: i) monitoring feedback received from the sensing circuit arrangement, wherein the feedback indicates the magnitude of the first energy, and ii) controlling the operation of the switching circuit arrangement via power factor correction, such that an average magnitude of the first energy over several control cycles may be substantially equal to an average magnitude of the second energy over the several control cycles.

[0025] In yet other examples, the detection circuit arrangement, as discussed herein, can be configured to include a series path comprising a first capacitor arranged in series with a first switch, the series path being arranged in parallel with the first transformer winding.

[0026] In another example, the sensing circuit arrangement can include a sensing capacitor that is effective in storing a voltage value that represents an integral of a current supplied by the input voltage through the first transformer winding.

[0027] The device, as discussed herein, may further comprise the following features: a signal generator circuit effective to produce a threshold signal based at least in part on a magnitude of the output voltage with respect to a set reference voltage; and a control effective to control the switching circuit arrangement and the flow of the first energy from the input voltage to the first transformer winding based on the threshold signal in order to apply the power factor correction.

[0028] Another device, as discussed herein, includes a control system.The control can be configured as follows: via a control of a switchgear arrangement, controlling a flow of first energy received from an input voltage through a first transformer winding magnetically coupled to a second transformer winding, wherein the controlled flow of first energy through the first transformer winding to the second transformer winding is effective in generating an output voltage based on second energy supplied from an output of the second transformer winding to a load; receiving feedback specifying a magnitude of first energy; and adjusting the operation of the switchgear arrangement over time based on at least the feedback and a magnitude of output voltage, wherein the adjusted operation of the switchgear arrangement is effective in adjusting a magnitude of first energy supplied to the first transformer winding.

[0029] Further examples, as discussed herein, include a method having the following features: via a switching circuit arrangement, controlling a flow of a first energy received from an input voltage through a first transformer winding, the first transformer winding being magnetically coupled to a second transformer winding, the controlled flow of the first energy through the first transformer winding to the second transformer winding generating an output voltage based on a second energy supplied from an output of the second transformer winding to a load; receiving feedback specifying a magnitude of the first energy;and applying a power factor correction via a control of the switching circuit arrangement over time based on at least the received feedback and a magnitude of the output voltage, wherein the control of the switching circuit arrangement includes a setting of a magnitude of the first energy supplied to the first transformer winding.

[0030] Controlling the flow of the first energy can involve controlling a resonant current supplied by the input voltage through the first transformer winding.

[0031] Furthermore, the control of the switch circuit arrangement can be configured over time to essentially balance an average size of the first energy and an average size of the second energy.

[0032] As further discussed herein, the control of the switch circuit arrangement over time can include the following steps: Setting the operation of the switching circuit arrangement over time based on one or more of: i) the feedback specifying the magnitude of the initial energy, ii) the magnitude of the output voltage, iii) a magnitude of the input voltage, iv) a capacitance associated with a sensing circuit arrangement generating the feedback, and iv) a switching period controlling the switching circuit arrangement.

[0033] Furthermore, the control of the switch circuit arrangement can include the following steps: generating a threshold signal; comparing the feedback, which indicates the magnitude of the initial energy, with the threshold signal; and terminating the flow of an initial current through the first transformer winding based on this comparison. The magnitude of the threshold signal can vary over time based on the magnitude of the input voltage.

[0034] Furthermore, the method, as discussed herein, may include the following steps: generating the threshold signal based on a combination of: i) the magnitude of the output voltage with respect to a set reference voltage, ii) the magnitude of the input voltage, and iii) a capacitance associated with a sensing circuit arrangement that generates the feedback.

[0035] In another example, the threshold signal is the threshold signal TS. Generating the threshold signal TS can involve the following steps: setting the threshold signal TS = OSV - [(tsw1 / (K * C)) * Vin] - cmp, where OSV is an offset value, where tsw1 is a measure of one period of controlling the switching circuit arrangement, where C is a capacitance associated with a sensing circuit arrangement that generates the feedback, where K is a value based on an error voltage derived from comparing the magnitude of the output voltage with a target reference voltage, where Vin is the magnitude of the input voltage, and where cmp is an optional compensation factor against currents induced by the input voltage AC line in the sensing circuit arrangement.

[0036] Furthermore, as discussed herein, the method can include receiving feedback from the sensing circuit arrangement. The feedback can be generated by the sensing circuit arrangement based on the integration of a first current supplied by the input voltage through the first transformer winding. The first transformer can be arranged in a resonant circuit, with the first current being a resonant current flowing through the first transformer winding.

[0037] It should also be noted that, although examples such as those discussed herein are applicable to controlling the operation of a power converter, the concepts disclosed herein may be advantageously applied to any other suitable topologies.

[0038] Additionally, it should be noted that, although each of the various features, techniques, configurations, etc., may be discussed at different points in this disclosure, it is intended, where appropriate, that each of the concepts may optionally be implemented independently or in combination with one another. Accordingly, the one or more inventions presented herein, as described herein, may be implemented and viewed in many different ways.

[0039] It should also be noted that this preliminary discussion of examples herein intentionally does not list every example and / or every incrementally novel aspect of the present disclosure or the claimed invention(s). Instead, this brief description presents only general examples and corresponding points of novelty over conventional techniques.

[0040] Preferred embodiments of the present invention are discussed in more detail below with reference to the accompanying drawings. Fig. Figure 1 is an example diagram illustrating a configuration of a power converter as discussed herein. Fig. Figure 2A is a diagram illustrating various exemplary implementations of a bidirectional switching circuit as discussed herein. Fig. Figure 2B is a diagram illustrating an exemplary implementation of a bidirectional switching circuit as discussed herein. Fig. Figure 3 is an exemplary detailed diagram illustrating a power converter circuit that implements multiple instances of a bidirectional switching circuit to convert an input voltage into an output voltage, as discussed herein. Fig. Figure 4 is an exemplary timing diagram illustrating the control of a respective bidirectional switching circuit arrangement in a power converter to convert an input voltage into an output voltage, as discussed herein. Fig. Figure 5 is an exemplary method for operating a respective power converter circuit comprising several instances of a bidirectional switching circuit arrangement to convert an input voltage into an output voltage, as discussed herein. Fig. Figure 6 is an exemplary circuit diagram illustrating a power converter as discussed herein. Fig. Figure 7 is an exemplary circuit diagram illustrating an implementation of a power converter and corresponding control as discussed herein. Fig. Figure 8 is an exemplary diagram illustrating a derivation of a control procedure using a generated threshold level, as discussed herein. Fig. Figure 9 is a timing diagram illustrating the control of a respective power converter over several AC input voltage cycles, as discussed herein. Fig. Figure 10 is an exemplary timing diagram illustrating the control of a respective power converter as discussed herein. Fig. Figure 11 is an exemplary diagram illustrating the control of a respective power converter as discussed herein. Fig. Figure 12 is an exemplary circuit diagram illustrating power flow measurement and control as discussed herein. Fig. Figure 13 is an exemplary diagram illustrating a detection circuit configured to monitor a quantity of current (first power) through a given primary transformer winding, as discussed herein. Fig. Figure 14 is an exemplary diagram illustrating a detection circuit configured to monitor a quantity of current (first power) through a given primary transformer winding, as discussed herein. Fig. Figure 15 is an exemplary diagram illustrating a detection circuit configured to monitor a quantity of current (first power) through a given primary transformer winding, as discussed herein. Fig. Figure 16 is an exemplary method for controlling a respective resonant / blocking power converter to provide a power factor correction as discussed herein.

[0041] The foregoing and other tasks, features, and advantages of examples herein will become apparent from the more detailed descriptions herein, as illustrated in the accompanying drawings, in which the same reference numerals in the various views refer to the same parts. The drawings are not necessarily to scale, the emphasis instead being placed on illustrating the examples, principles, concepts, etc.

[0042] With reference to the drawings, it is now Fig. 1 A diagram illustrating a power supply comprising a bridgeless hybrid blocking system using a bidirectional circuit arrangement as shown in the examples herein.

[0043] As in Fig. As shown in Figure 1, the power converter 100 comprises a power source 120, a bidirectional switching circuit arrangement 131, a bidirectional switching circuit arrangement 132, a transformer 161, a capacitor C1, a capacitor C2, a capacitor C3, a capacitor C4, a diode D1 and a load 118.

[0044] The bidirectional switch circuit arrangement 131 comprises a switch Q11 and a switch Q12. The bidirectional switch circuit arrangement 132 comprises a switch Q21 and a switch Q22.

[0045] The transformer 161 comprises a primary winding 161-1 and a secondary winding 161-2. The secondary winding 161-2 is magnetically coupled (inductively coupled) to the primary winding 161-1. The primary winding 161-1 is connected between nodes N3 and N6. The secondary winding 161-2 is connected between nodes N7 and N8.

[0046] The power source 120, which provides the input voltage Vin (such as an AC voltage or an input voltage 121), is connected between nodes N1 and N2. Accordingly, the voltage across nodes N1 and N2 is Vin. The capacitor C1 (such as a standard EMI capacitor or an electromagnetic interference capacitor) is connected in parallel with the power source 120 between nodes N1 and N2 to filter high-frequency currents and to store a small charge associated with the input voltage Vin.

[0047] It should be noted that the bidirectional switch circuit arrangement 131 or 132 can be configured in any suitable way. A non-restrictive example of an implementation of the bidirectional switch circuit arrangement associated with the power transformer 100 is given in Fig. 2 shown. The bidirectional switch circuit arrangement shown in Fig. Figure 2 is discussed in more detail below.

[0048] With renewed reference to Fig. Note that the source node S of switch Q11 is connected to node N1. The drain node D of switch Q11 is connected to node N4 and the corresponding drain node D of switch Q12. The source node S of switch Q12 is connected to node N3 and the corresponding source node S of switch Q21. The drain node D of switch Q21 is connected to the drain node D of switch Q22. The source node S of switch Q22 is connected to node N2.

[0049] It should be noted that the bidirectional switch circuit arrangement, as discussed herein, can be configured in any suitable way. For example, a first option No. 1 of the bidirectional switch circuit arrangement, as discussed herein, is in Fig. 2A shown, such as that it has a common drain connection; a second option No. 2 of the bidirectional switch circuit arrangement, as discussed herein, is shown in Fig. 2A shows, for example, that it has a common source connection; and a BDS switch option No. 3 is in Fig. 2A shown.

[0050] It should be noted that Fig. 1 illustrates an implementation of the first option No. 1; Fig. Figure 3 illustrates an implementation of option 3. Another option, number 4, of the bidirectional switching circuit arrangement (such as a so-called IGBT or insulated-gate bipolar transistor) is described below. Fig. 2B discussed.

[0051] Even further away is, as in Fig. Figure 1 shows capacitor C2 connected between nodes N1 and N6. Capacitor C3 is connected between nodes N6 and N2.

[0052] Each of the capacitors C2 and C3 can have any suitable capacitance value.

[0053] Accordingly, a combination of the bidirectional switch circuit arrangement 131 and the bidirectional switch circuit arrangement 132 is connected in series between nodes N1 and N2. The series circuit path comprising the bidirectional switch circuit arrangement 131 and the bidirectional switch circuit arrangement 132 is arranged in parallel with the power source 120 and the corresponding capacitor C1.

[0054] It should also be noted that the power converter 100, which is in Fig. As shown in 1, it can be configured to include a controller 140.

[0055] In one example, the controller 140 generates the control signal S11 (i.e., LSN) to drive the gate node G of switch Q11. The control signal S11 is used to switch switch Q11 between an ON state and an OFF state.

[0056] Controller 140 generates the control signal S12 (i.e., HSP) to drive the gate node G of switch Q12. The control signal S12 is used to switch switch Q12 between an ON state and an OFF state.

[0057] Controller 140 generates the control signal S21 (i.e., HSN) to drive the gate node G of switch Q21. The control signal S21 is used to control switch Q21 between an ON state and an OFF state.

[0058] Controller 140 generates the control signal S22 (i.e., LSP) to drive the gate node G of switch Q22. The control signal S22 is used to switch switch Q22 between an ON state and an OFF state.

[0059] As shown in this example and as discussed previously, each instance of the bidirectional switching circuit arrangement, as discussed herein, is configured to selectively block or conduct current regardless of voltage direction.

[0060] For example, the bidirectional switch circuit arrangement 131 can be controlled to prevent the transmission of voltage / current from node N1 to node N3.

[0061] The bidirectional switch circuit arrangement 131 can be controlled to prevent the transmission of voltage / current from node N3 to node N1.

[0062] Similarly, the bidirectional switch circuit arrangement 132 can be controlled to prevent a current flow from node N2 to node N3.

[0063] The bidirectional switching circuit arrangement 132 can be controlled to prevent the current flow from node N3 to node N2.

[0064] It is noted that a current flow 151 (such as a varying magnitude) through the primary winding 161-1 can cause a corresponding current flow 152 through the secondary winding 161-2. The current flow 152 through the secondary winding 161-2 generates the output voltage 123 (i.e., Vout) across the capacitor C4 and the load 118.

[0065] It is further noted that there are two phases: one only for the input current 151 and one for the transmission currents 151 and 152 at the same time.

[0066] Diode D1 prevents current 152 from flowing in a negative direction through secondary winding 161-2. In other words, the secondary current 152 through secondary winding 161-2 flows in one direction from node N7 to node N8. Diode D1 prevents current flow from node N7 through diode D1 to node N9 when diode block D1 is in the energy absorption phase.

[0067] It is noted that the power converter 100 in Fig. Figure 1 is shown as a non-restrictive example. As discussed below, the power converter 140, as discussed herein, can also be implemented as in Fig. 3 shown, or implemented via other configurations.

[0068] Thus, with renewed reference to Fig. 1. The device, as discussed herein, such as a power supply, a power converter, or another suitable entity, may be configured to include the following features: a first winding 161-1; a first bidirectional switching circuit arrangement 131; and a second bidirectional switching circuit arrangement 132 arranged in series with the first bidirectional switching circuit arrangement 131. A combination of the first bidirectional switching circuit arrangement 131 and the second bidirectional switching circuit arrangement 132 may be configured to control the magnitude of a current 151 through the first winding 161-1 of the transformer 161 to produce an output voltage 123 (i.e., Vout).

[0069] Furthermore, the second winding 161-2 of the transformer 161 can be magnetically coupled to the first winding 161-1 of the transformer 161. In such a case, a current flow 151 (i.e., iPRI) through the first winding 161-1 effectively converts a received input voltage Vin and a corresponding current 121 into the output voltage 153 (i.e., Vout) via an output of the output voltage 153 and the corresponding current 152 from the second winding 161-2 to the load 118.

[0070] Furthermore, as discussed above, the first bidirectional switching circuit arrangement 131 can be effective in blocking the passage of voltage in both a first direction and a second direction through the first bidirectional switching circuit arrangement 131; the second bidirectional switching circuit arrangement 132 can be effective in blocking the passage of voltage in both a first direction and a second direction through the second bidirectional switching circuit arrangement 132.

[0071] Furthermore, the first bidirectional switch circuit arrangement 131 can comprise a first switch Q11 arranged in series with a second switch Q12; the second bidirectional switch circuit arrangement 132 can comprise a third switch Q21 arranged in series with a fourth switch Q22.

[0072] A combination of the first switch Q11 and the second switch Q12 can be arranged in a first series path between a node N1 and a node N3; a combination of the third switch Q21 and the fourth switch Q22 can be arranged in a second series path between the node N3 and the node N2.

[0073] Even further encompass, as in Fig. Figure 1 shows examples, as discussed herein, that the power converter 100 further comprises the following features: a capacitor C2 arranged in parallel to a combination of the first bidirectional switching circuit arrangement 131 and the first winding 161-1; and a capacitor C3 arranged in parallel to a combination of the second bidirectional switching circuit arrangement 132 and the first winding 161-1.

[0074] A first circuit path, comprising capacitor C2 and winding 161-1, is a first resonant circuit; a second circuit path, comprising capacitor C3 and winding 161-1, is a second resonant circuit. The first resonant circuit path is arranged in parallel with the second resonant circuit path.

[0075] Furthermore, as discussed previously, the bidirectional switch circuit arrangement, such as the bidirectional switch circuit arrangement 131 and the bidirectional switch circuit arrangement 132, can be implemented in any suitable way. An example of the bidirectional switch circuit arrangement 131 and the bidirectional switch circuit arrangement 132 is shown in Fig. 2 shown.

[0076] Fig. Figure 2B is a diagram illustrating an exemplary implementation of a bidirectional switch as discussed herein.

[0077] In this example, an implementation of the bidirectional switch circuit arrangement 131 (such as the bidirectional switch circuit arrangement 131-1) can include a transistor 211, a transistor 212, a diode D11 and a diode D12.

[0078] Transistor 211 and diode D11 can be connected together at node N4. The combination of transistor 211 and diode D11 can be arranged in series between node N1 and node N3.

[0079] The diode D12 and the transistor 212 can be connected to each other at node N4 and are arranged in series between node N1 and node N3.

[0080] An implementation of the bidirectional switch circuit arrangement 132 (such as the bidirectional switch circuit arrangement 132-1) can comprise a transistor 221, a transistor 222, a diode D21, and a diode D22. The transistor 221 and the diode D21 can be connected together at node N5 and can be arranged in series between node N2 and node N3. The diode D22 and the transistor 222 can be connected together at node N5 and can be arranged in series between node N2 and node N3.

[0081] Another example of an implementation of the first bidirectional switching circuit arrangement 131 and the second bidirectional switching circuit arrangement 132 in an AC-DC power converter is shown in Fig. Figure 3 is shown. It is noted that the output voltage 153 can be a DC voltage.

[0082] Fig. Figure 3 is an exemplary power converter circuit diagram illustrating an implementation of multiple bidirectional dual-gate switches, such as gallium nitride (GaN) switches, in a power converter to convert an input voltage into an output voltage, as discussed herein.

[0083] As discussed previously, the first bidirectional switch circuit arrangement 131 and the second bidirectional switch circuit arrangement 132 can be implemented in any suitable way.

[0084] In one example, the first bidirectional switch circuit arrangement 131 can be implemented as a bidirectional dual-gate switch circuit arrangement 131-2, such as a first GaN (gallium nitride) switch. The second bidirectional switch circuit arrangement 132 can be implemented as a bidirectional dual-gate switch circuit arrangement 132-2, such as a second GaN (gallium nitride) switch.

[0085] As further in Fig. As shown in Figure 3, the first bidirectional switch circuit arrangement 131-2 can be a first dual-gate switch comprising a first drain node D31, a first source node S31, a first gate node GF1 (such as a floating gate) and a second gate node G31.

[0086] The second bidirectional switch circuit arrangement 132-1 can be a second dual-gate switch comprising a drain node D32, a source node S32, a gate node GF2 (such as a floating gate) and a gate node G32.

[0087] The power converter 100-1 also includes the EMI cancellation circuit 301, such as capacitor C31, capacitor C32 (similar to capacitor C1 in Fig. 1) comprising inductor 311 and inductor 312. If desired, controller 140 or another suitable entity provides power factor correction.

[0088] In this example, the power converter 100-1 also includes a capacitor C39, which is arranged in series with the transformer winding 161-1 between node N3 and node N6. The series path can be configured to support current sensing.

[0089] As further shown, the power converter 100-1 can be configured to include the controller 140-1 (which operates similarly to the controller 140) to control the first bidirectional switching circuit arrangement 131-2 and the second bidirectional switching circuit arrangement 132-2 based on a feedback signal 399 (such as a voltage) generated at node N14, which directly connects capacitor C39 and the first winding 161-1.

[0090] It should be noted that the controller can be configured to monitor feedback from any suitable node to control the first bidirectional switch circuit arrangement and the second bidirectional switch circuit arrangement. For example, additional examples, as discussed herein, may include monitoring node N6 or N8 to control the switch circuit arrangement.

[0091] Fig. Figure 4 is an exemplary timing diagram illustrating the control of a respective bidirectional switching circuit arrangement in a power converter to convert an input voltage into an output voltage, as discussed herein.

[0092] With reference to the power converter 100 in Fig. 1 and the time diagram 400 in Fig. 4 activates the controller 140 during a first mode (when the polarity of the input voltage Vin is positive), such as between time T31 and time T41, in which the input voltage Vin between node N1 and node N2 is positive, as detected by the controller 140 or another suitable entity: i) both switch Q11 (by driving the control signal S11 to a logic high between time T31 and time T41) and switch Q21 to ON states (by setting the control signal S21 to a logic high between time T31 and time T41), and ii) alternately switches between activating the second switch Q12 and the fourth switch Q22 to an ON state to control the magnitude of the current 151 through the first winding 161-1.

[0093] In other words, during the first mode, between time T31 and time T41, when controller 140 activates switch Q12 to an ON state, it deactivates switch Q22 to an OFF state. Conversely, when controller 140 deactivates switch Q12 to an OFF state, it activates switch Q22 to an ON state.

[0094] It should be noted that the switching frequency of the turning on and off of switches Q12 and Q22 between time T31 and time T41 is significantly larger than the mains frequency (as defined by the period of the input voltage Vin between time T31 and time T51) which is associated with the input voltage Vin.

[0095] Between time T51 and time T61, between time T71 and time T81, etc., the controller 140 controls the operation of the respective bidirectional switch circuit arrangement 131 and the bidirectional switch circuit arrangement 132 in a similar manner to that discussed above (the first mode) for the duration between time T31 and time T41.

[0096] Between time T21 and time T31, between time T41 and time T51, between time T61 and time T71, etc., the controller 140 operates in a second mode (when the polarity of the input voltage Vin is negative).

[0097] During the second mode, in which the input voltage Vin between node N1 and node N2 is negative, the controller 140: i) activates the bidirectional switching circuit arrangement Q12 and the bidirectional switching circuit arrangement Q22 to ON states between time T41 and time T51, and ii) switches the bidirectional switching circuit arrangement Q11 and the bidirectional switching circuit arrangement Q21 to an ON state between activating them to control the magnitude of the current 151 through the first winding 161-1.

[0098] In other words, during the second mode, between time T41 and time T51, when controller 140 activates switch Q11 to an ON state, it deactivates switch Q21 to an OFF state. Conversely, when controller 140 deactivates switch Q11 to an OFF state, it activates switch Q21 to an ON state.

[0099] It should be noted that the switching frequency of the turning on and off of switches Q11 and Q12 between time T41 and time T51 is significantly larger than the mains frequency (as defined by the period of the input voltage Vin between time T31 and time T51) which is associated with the input voltage Vin.

[0100] Between time T21 and time T31, between time T61 and time T71, etc., the controller 140 controls the operation of the respective bidirectional switch circuit arrangement 131 and the bidirectional switch circuit arrangement 132 in a similar manner to that discussed above (the first mode) for the duration between time T41 and time T51.

[0101] As discussed above, the second winding 161-2 in the transformer 161 can be magnetically coupled to the first winding 161-1. In such a case, a current flow 151 through the first winding 161-1 effectively induces a current flow 152 through the secondary winding 161-2 to generate the output voltage 153, which is then supplied from the second winding 161-2 to the load 118 and the corresponding capacitor C4. The controller 140 can be configured to regulate the magnitude of the output voltage 153 by switching between operation in the first mode and the second mode.

[0102] With reference to the power converter 100-1 in Fig. 3 and the time diagram 400 in Fig. 4. During a first mode (when the polarity of the input voltage Vin is positive), such as between time T31 and time T41, when the input voltage Vin between node N1 and node N2 is positive, as detected by the controller 140-1 or another suitable entity, the controller 140-1: i) generates the control signal S11, which is applied to the gate node GF1, so that it is a logic high between time T31 and time T41, and generates the control signal S21, which is supplied to the gate node G32, so that it is a logic high between time T31 and time T41, and ii) switches between logic high and logic low states between generating the control signal S12 and the control signal S22 in a similar manner to that discussed above.

[0103] Specifically, during the first mode, when the controller 140-1 generates the control signal S12, which is applied to gate node G31 in a logic high state, it also generates the control signal S22, which is applied to gate node GF2 in a logic low state. Conversely, when the controller 140-1 generates the control signal S12, which is applied to gate node G31 in a logic low state, it also generates the control signal S22, which is applied to gate node GF2 in a logic high state.

[0104] It should be noted that the switching frequency of the turning on and off of the gate nodes G31 and GF2 between time T31 and time T41 is significantly larger than the mains frequency (as defined by the period of the input voltage Vin between time T31 and time T51) which is associated with the input voltage Vin.

[0105] Between time T51 and time T61, between time T71 and time T81, etc., the controller 140 controls the operation of the respective bidirectional switch circuit arrangement 131-2 and the bidirectional switch circuit arrangement 132-2 in a similar manner to that discussed above (the first mode) for the duration between time T31 and time T41.

[0106] Between time T21 and time T31, between time T41 and time T51, between time T61 and time T71, etc., the controller 140 operates in a second mode (when the polarity of the input voltage Vin is negative).

[0107] During a second mode (when the polarity of the input voltage Vin is negative), such as between time T41 and time T51, in which the input voltage Vin between node N1 and node N2 is negative, as detected by the controller 140-1 or another suitable entity, the controller 140-1: i) generates the control signal S22, which is applied to the gate node GF2, so that it is a logic high between time T41 and time T51, and generates the control signal S12, which is supplied to the gate node G31, so that it is a logic high between time T41 and time T51, and ii) switches between logic high and logic low states between generating the control signal S11 and the control signal S21.

[0108] Specifically, during the second mode, when the controller 140-1 generates the control signal S21, which is applied to gate node G32 in a logic high state, it also generates the control signal S11, which is applied to gate node GF1 in a logic low state. Conversely, when the controller 140-1 generates the control signal S21, which is applied to gate node G32 in a logic low state, it also generates the control signal S11, which is applied to gate node GF1 in a logic high state.

[0109] It should be noted that the switching frequency of the turning on and off of the gate nodes G32 and GF1 between time T41 and time T51 is significantly larger than the mains frequency (as defined by the period of the input voltage Vin between time T31 and time T51) which is associated with the input voltage Vin.

[0110] Between time T21 and time T31, between time T61 and time T71, etc., the controller 140 controls the operation of the respective bidirectional switching circuit arrangement 131-2 and the bidirectional switching circuit arrangement 132-2 in a manner similar to that discussed above (the first mode) for the duration between time T41 and time T51. In other words, between time T21 and time T31, between time T41 and time T51, between time T61 and time T71, etc., the controller 140 operates in a second mode (when the polarity of the input voltage Vin is negative).

[0111] Fig. Figure 5 is an exemplary method for operating a respective power converter circuit comprising several instances of a bidirectional switching circuit arrangement to convert an input voltage into an output voltage, as discussed herein.

[0112] In this example, the control unit 140 of the power converter 100 controls the operation of the first bidirectional switch circuit arrangement 131 in the processing operation 510.

[0113] In the processing operation 520, the controller 140 controls the operation of the second bidirectional switch circuit arrangement 132, which is arranged in series with the first bidirectional switch circuit arrangement 131.

[0114] In the processing operation 530, the controller 140 can be configured via the controlled operation of the first bidirectional switch circuit arrangement 131 and the second bidirectional switch circuit arrangement 132 to control a magnitude of current 151 through the first winding 161-1 to generate an output voltage Vout which is supplied to the load 118.

[0115] It should be noted again that the techniques described herein are well suited for use in power supply applications. However, it should be noted that the examples given here are not limited to use in such applications and that the techniques discussed herein are also well suited to other applications.

[0116] Fig. Figure 6 is an exemplary circuit diagram illustrating a power converter as discussed herein.

[0117] As discussed previously, it should be noted again that the implementation of the example circuits contained herein may vary depending on the application. For example, the following illustrates Fig. 6. A circuit example that includes bidirectional switches. It should be noted that Fig. 13 to Fig. Figure 15 illustrates a unidirectional circuit option for implementing techniques as discussed herein. Accordingly, the techniques herein can be implemented in both unidirectional and bidirectional circuit options. In this case, the previously defined signals LSN and HSP are combined into a single HS signal that drives a unidirectional transistor. In the same way, LSP and HSN can be combined into a single LS signal.

[0118] Additionally, it should be noted that the control concept, as discussed herein using a generated threshold-signal TS control, can be described as a charge control technique, such as one based on dq or DeltaQ, since the control can be implemented based on a charge difference. For this purpose, the circuit arrangement, as discussed herein, can be configured to generate a shunt capacitor voltage (such as feedback 725) produced by a sensing circuit 145, and the control threshold (such as threshold signal TS) produced by the threshold level generator 715 (such as an integrated circuit or a semiconductor chip), as shown in Fig. 7 shown, to include.

[0119] In this example of Fig. 6 The power converter 100-6 comprises a power source 120 (input voltage source), a bidirectional switching circuit arrangement 131, a bidirectional switching circuit arrangement 132, a transformer 161, a capacitor C1, a capacitor C2, a capacitor C3, a capacitor C4, a diode D1 and a load 118.

[0120] The bidirectional switch circuit arrangement 131 comprises a switch Q11 and a switch Q12. The bidirectional switch circuit arrangement 132 comprises a switch Q21 and a switch Q22.

[0121] It should also be noted that circuit arrangement 131 can include a single field-effect transistor switch instead of multiple switches Q11 and Q12; circuit arrangement 132 can include a single field-effect transistor switch instead of multiple switches Q21 and Q22. Thus, it is not necessary to include a bidirectional switch circuit arrangement.

[0122] Transformer 161 comprises a primary winding 161-1 and a secondary winding 161-2. The secondary winding 161-2 is magnetically coupled (inductively coupled) to the primary winding 161-1. In this example, a combination of the primary winding 161-1 and the capacitor C19 is connected in series between nodes N3 and N6. The secondary winding 161-2 is connected between nodes N7 and N8.

[0123] The power source 120, which provides the input voltage Vin (such as an AC voltage or an input voltage Vin or an input current 121), is connected between nodes N1 and N2. Accordingly, the voltage across nodes N1 and N2 is Vin.

[0124] It should also be noted that the power converter 100-6, which is in Fig. As shown in Figure 6, it can be configured to include a controller 140 that generates the respective control signals.

[0125] In one example, the controller 140 generates the control signal S11 (i.e., LSN) to drive the gate node G of switch Q11. The control signal S11 is used to switch switch Q11 between an ON state and an OFF state.

[0126] Controller 140 generates the control signal S12 (i.e., HSP) to drive the gate node G of switch Q12. The control signal S12 is used to switch switch Q12 between an ON state and an OFF state.

[0127] Controller 140 generates the control signal S21 (i.e., HSN) to drive the gate node G of switch Q21. The control signal S21 is used to control switch Q21 between an ON state and an OFF state.

[0128] Controller 140 generates the control signal S22 (i.e., LSP) to drive the gate node G of switch Q22. The control signal S22 is used to switch switch Q22 between an ON state and an OFF state.

[0129] It is noted that a current flow 151 (such as a varying magnitude) through the primary winding 161-1 causes a corresponding current flow 152 through the secondary winding 161-2. The current flow 152 through the secondary winding 161-2 generates the output voltage 123 (i.e., Vout) across the capacitor C4 and the load 118.

[0130] Diode D1 prevents current 152 from flowing in a negative direction through secondary winding 161-2. In other words, the secondary current 152 through secondary winding 161-2 flows in one direction from node N7 to node N8. Diode D1 prevents current flow from node N7 through diode D1 to node N9.

[0131] It is noted that the power converter 100-6 in Fig. 6 is shown as a non-restrictive example.

[0132] In this example, the power supply 100-6 includes the detection circuit arrangement 145, such as the capacitor C19, the resistor R1, the capacitor C21 and the switch 325.

[0133] The sensing circuit 145 is configured to detect an initial power supplied by the input voltage Vin through the first transformer winding 161-1 to a second transformer winding 161-2. As discussed above, the second transformer winding 161-2 is magnetically coupled to the first transformer winding 161-1.

[0134] As further discussed herein, the controller 140 and the switch circuit arrangement (Q11, Q12, Q21 and Q22) together provide a power factor correction which is associated with a conversion of the input voltage Vin into an output voltage Vout, which is output from an output of the second transformer winding 161-2 to the load 118.

[0135] The power factor correction, as discussed herein, can be configured to control a flow of first power (and a corresponding magnitude) from the input voltage Vin through the first transformer winding 161-1.

[0136] Additional details of the implementation of the detection circuit arrangement 145 and the corresponding control of the respective switches to provide power factor correction are discussed in the following drawings and the corresponding descriptive text.

[0137] Fig. Figure 7 is an exemplary circuit diagram illustrating an implementation of a power converter and corresponding control as discussed herein.

[0138] In this example, each of the power converters, as discussed herein, comprises a respective switch circuit arrangement 710 (such as a switch Q11, a switch Q12, etc.) to control a respective magnitude of current (and thus power P1) supplied by the input voltage Vin to the transformer winding 161-1.

[0139] As discussed above, transformer winding 161-2 is magnetically coupled to transformer winding 161-1 to receive power P1 from transformer winding 161-1. In other words, transformer 161 facilitates the transfer of power P1 from transformer winding 161-1 to transformer winding 161-2. The energy P1 received by the secondary transformer winding 161-2 is used as a basis to generate the output voltage that delivers power P2 to load 118. Techniques employed here include power factor correction and significant equalization of the average power P1 and the average power P2.

[0140] Furthermore, it should be noted in this example that the controller 140 or another suitable entity can be configured to include a circuit arrangement to control the transmission of current 151 through the corresponding winding 161-1 based on the use of a respective control signal (threshold signal TS) generated by the threshold level generator 715. The current 151 may or may not flow through the sensing circuit arrangement 145.

[0141] As further shown, it should be noted that the circuit arrangement, as discussed herein, can be configured to include one or more of an error voltage signal generator 735, a filter function 738, a threshold level generator 715 and a comparator 720.

[0142] As shown in this example, the fault voltage signal generator 735 generates the respective fault signal 731 (such as a fault voltage, fault current, etc.) based on a difference between a value of the output voltage Vout and the target reference voltage 729. In one example, the target reference voltage 729 represents the target value at which the controller 140 regulates the value of the output voltage Vout via an implementation of the control signal (threshold signal TS) generated by the generator 715.

[0143] Furthermore, it should be noted that the filter function 738, such as a PID controller (where P = Proportional, I = Integral, D = Differential), a PI controller or another suitable entity, converts the received error signal 731 into the signal K (such as a filtered error signal) which is supplied to the threshold level generator 715.

[0144] The threshold level generator 715 receives the signal K, the capacitance C, which is assigned to the detection circuit 145, the switching period Tsw, which is assigned to the operation of the respective switches 710, and the input voltage Vin.

[0145] In one example, the controller 140 and / or a corresponding circuit arrangement assigned to the controller 140 generate the control threshold signal TS as follows: Setting the threshold signal TS = OSV - [(tsw1 / (K * C)) * Vin] - cmp (note that K can be placed in the denominator or numerator depending on whether a PID is being created). The minus sign can also be positive depending on the acquisition configuration. In other words, if the acquisition reference is not AC_HB but instead capacitor C19 - (see Fig. 6). If OSV is a selected offset value, it's important to note that OSV can be negative, positive, or zero. In one example, OSV is constant for a single switching cycle (not AC), but can otherwise vary over time. If the switching period Tsw1 is a measure of a period of controlling the switch circuit arrangement around the time of generating the corresponding threshold signal TS, the switching period may be generated based on one or more current or previous switching periods of controlling the switches, or may possibly be based on the current switching period of controlling the switches in the current control cycle. If C is the capacitance associated with the sensing circuit arrangement 145 (there is one of several different configurations associated with the sensing circuit arrangement 145. For one configuration, C is an equivalent capacitance to the shunt capacitor). If K is the feedback (filtered fault voltage value), and If Vin is the sampling instance of the magnitude of the input voltage around the time of generating the corresponding threshold signal TS. If cmp is an optional compensation factor against currents induced by the AC line (input voltage) in the sensing circuit arrangement, then in an example, the line current (@50 / 60 Hz) from the input voltage flows through capacitors C91 and C92, interrupting the sensing network. A portion of this current is considered by the sensing network to be transferred to the secondary side when it only circulates locally. The cmp compensation improves the power factors. The compensation (Cmp) can be proportional to dVin / dt.

[0146] As further shown, the comparator 720 is used to generate the trigger signal 755, which (by the controller 140) is used as a basis to stop the activation of one or more of the switches Q11 and Q12 during a positive cycle when the input voltage is greater than 0, or to stop the activation of one or more of the switches Q21 and Q22 during a negative cycle when the input voltage is less than 0, with both operations limiting the amount of power supplied by the input voltage Vin to the transformer winding 161-1 to provide power factor correction.

[0147] In one example, the comparator 720 generates the respective trip signal 755 in response to the comparator 720 detecting a transition of the feedback 725 (which specifies the power P1, such as based on the integration of the current) that crosses the respective threshold signal TS. As further discussed herein, the trip signal 755 can be used as a basis to terminate the activation of the respective switch circuit arrangement 710, so that the input voltage Vin of the transformer winding 161-1 no longer supplies the corresponding power P1.

[0148] Accordingly, the controller 140 and the corresponding sensing circuit 145, as discussed herein, can be configured to jointly: i) monitor a quantity of the first power (integration of the current 151) via a feedback 725 (signal) from the sensing circuit 145, and ii) provide a power factor correction via a controlled operation of the switching circuit arrangement (Q11 and Q12) such that an average quantity of the first power P1 delivered by the input voltage Vin to the first transformer winding 161-1 over time is substantially equal to an average quantity of the second power P2 delivered by the second transformer winding 161-2 to the respective load 118.

[0149] With renewed reference to Fig. 6. The circuit path between node N3 and node N6 can be configured to include a first capacitor C19, which is coupled in series with the first transformer winding 161-1. The controller 140 is effective in controlling the flow of a resonant current through the series path via a control of the switch circuit arrangement. The controlled flow of the resonant current is effective in providing the power factor correction, as discussed herein.

[0150] Furthermore, the detection circuit arrangement 145 in this example includes a series path comprising a capacitor C21 which is arranged in series with the switch 325, wherein the series path (combination of C21 and switch 325) is arranged in parallel with the capacitor C19.

[0151] Furthermore, as previously discussed, the sensing circuit 145 in this example includes a sensing capacitor C19, which is placed in series with the first transformer winding 161-1 to sensing the initial number of charges (or the quantity or magnitude of charge) supplied to the first transformer winding 161-1 by the input voltage Vin. It should be noted that the shunt capacitor cannot sensing power—it generates an arithmetic sum of the quantity of charges flowing through it (i.e., the capacitor integrates based on the current through it). The sensing capacitor C19 is initially set to an initial state, such as 0 volts or another suitable value for a measurement phase of monitoring a quantity of power P1. The output voltage Vout is generated by the secondary stage of the power supply 100-6 based on the power received by the transformer winding 161-2 (i.e.,The second transformer winding 161-2 receives power from the first transformer winding 161-1. The second transformer winding 161-2 uses the received power P1 to generate the output voltage Vout, which is supplied to power the load 118. The load 118 consumes the power P2.

[0152] Examples here include controlling the amount of power input to the first winding 161-1 and transferred to the second winding 161-2 to provide power factor correction. For example, the sensing circuit 145 generates a corresponding feedback signal 725 (i.e., such as the voltage V19 across capacitor C19).The controller 140: i) monitors a voltage level (such as the feedback 725) across capacitor C19 to determine the power P1 supplied to transformer winding 161-1, and ii) controls the operation of the switching circuit arrangement (switches Q11, Q12, Q21, and Q22) such that the average amount of power transferred from winding 161-1 to winding 161-2 over several control cycles is essentially equal to the average amount of power consumed by the respective dynamic load 118 over the same control cycles. In other words, on average, the amount of power supplied to transformer winding 161-1 and transferred to the secondary transformer winding 161-2 should generally be equal to the amount of power supplied by the secondary winding 161-2 and consumed by the load 118.

[0153] As previously discussed, the sensing circuit 145, as a non-restrictive example, includes a sensing capacitor C19 which is effective to store a voltage value (i.e., the feedback 725) which indicates an integral of a current 151 (which also indicates the power) supplied by the first transformer winding 161-1.

[0154] As discussed below, the threshold signal generator circuit 715 generates the threshold signal TS (threshold level) based on one or more parameters, such as the signal K, the capacitance associated with the sensing circuit 145, the switching period Tsw, and the magnitude of the input voltage Vin. The controller 140 controls the switching circuit arrangement based on the threshold signal TS to provide the power factor correction. Details related to the generation of the threshold signal TS are given in Fig. 8 discussed further.

[0155] Fig. Figure 8 is an exemplary diagram illustrating a derivation of a tax procedure as discussed herein.

[0156] In this example, the threshold signal TL (threshold value, such as [Tsw / (K * C)] * Vin) is derived based on several received values, such as the input voltage Vin, the switching period Tsw, the filtered error signal K and the capacitance C associated with the detection circuit 145.

[0157] The connection between average cycle current and resonant capacitor and Delta V, as discussed herein, is useful.

[0158] As discussed above, a comparator 720 (and the corresponding threshold signal TS and feedback 725) can be used to determine the precise time at which iPRI or the input current 121 (or the current 151) and the corresponding first power P1 supplied to the transformer winding 161-1 have reached a certain threshold level. In one example, the average of the current 151 over time is integrated across the respective sensing circuit 145 to detect the first power P1 supplied to the transformer winding 161-1 by the input voltage Vin.

[0159] In one example, the real control is based on the stored charge q, which represents a quantity of the transferred charge; and the charge transferred during a duration is the average current.

[0160] In another example, if the input current iPRI is kept proportional to the input voltage, the input behaves like a PFC (Power Factor Correction).

[0161] This is an example definition of a PFC: If iIPRI = k*Vin, then it is a PFC.

[0162] The only thing left to do is to control Delta V so that it is proportional to Vin. For example, Delta u = Δu = T_sw / KC * V_in fulfills the PFC requirement. Tsw is the switching period.

[0163] In one example, the signal K is a scaling factor representing the system load resistance.

[0164] C is the equivalent capacity of the acquisition network assigned to the acquisition circuit 145.

[0165] For a closed-loop control system, K can be used as a feedback input.

[0166] Any controller (PI, PID, P ...) or filter (such as a low-pass, notch, ...) 738 with low gain at an AC frequency, such as 50 to 100 Hertz, can be used to generate the signal K.

[0167] The input voltage Vin, which is fed into the threshold signal generator 715, can be measured spontaneously (instantaneous voltage) because the input voltage Vin varies over time, whereby the magnitude of the first power stored in the transformer winding 161-1 depends over time on the magnitude of the input voltage Vin (such as a sine wave).

[0168] Tsw can be generated and / or measured using any suitable circuit, such as a so-called bang-bang circuit, which determines a recent magnitude of the switching cycle or the last value, the slope (the actual value of the current cycle), etc.

[0169] Fig. Figure 9 is a timing diagram illustrating the control of a respective power converter over several AC input voltage cycles, as discussed herein.

[0170] As shown in the timing diagram 800, the input voltage Vin is essentially a sine wave (AC signal or alternating current or alternating voltage), with the input voltage Vin having a positive polarity between time T31 and time T41, between time T51 and time T61, and so on. The input voltage Vin has a negative polarity between time T21 and time T31, between time T41 and time T51, and so on.

[0171] The voltage V4 is the voltage at node N6.

[0172] The signal S21 (HSN) controls the operation of switch Q21; the signal S12 (HSP) controls the operation of switch Q12; the signal S11 (LSN) controls the operation of switch Q11; and the signal S22 (LSP) controls the operation of switch Q22.

[0173] When the input voltage Vin has a positive polarity: The controller 140 generates the control signal S11, which is a logic high to activate the corresponding switch Q11. The controller 140 also generates the control signal S21, which is a logic high to activate switch Q21. Via the signals S12 and S22, the controller 140 alternately switches between activating switch Q12 and switch Q22. As discussed herein, switches Q21 and Q22 are controlled to supply a desired amount of power from the input voltage Vin through the transformer winding 161-1 to provide power factor correction.

[0174] If the input voltage has a negative polarity: The controller 140 generates the control signal S12, which is a logic high to activate the corresponding switch Q12. The controller 140 also generates the control signal S22, which is a logic high to activate switch Q22. Via the signals S11 and S21, the controller 140 alternately switches between activating switch Q11 and switch Q21. As discussed herein, switches Q11 and Q21 are controlled to supply a desired amount of power from the input voltage Vin through the transformer winding 161-1.

[0175] As discussed above, the feedback 725 is generated by the detection circuit 145 (monitoring circuit), which monitors a quantity of current 151 (and thus power) supplied to the transformer winding 161-1.

[0176] As further shown and discussed above, the timing diagram 900 also illustrates how the magnitude of the threshold signal TL varies over time to provide power factor correction. For example, as discussed above, the controller 140 receives the feedback signal 725 (shown as an envelope and timing diagram 900) from the sensing circuit 145. By comparing the feedback signal 725 with the threshold signal TL (e.g., a control signal), the controller 140 provides power factor correction based on controlled operation of the switching circuit arrangement.In one example, the controlled operation of the switch circuit arrangement results in an average magnitude of the first power (as specified by the sensing circuit 145) supplied by the input voltage Vin to the first transformer winding 161-1 being essentially equal to an average magnitude of the second power (such as based on an output current, as supplied by the output voltage Vout) output by the second transformer winding 161-2 to the respective load 118.

[0177] Fig. Figure 10 is an exemplary timing diagram illustrating the control of a respective power converter as discussed herein.

[0178] In this example, at time T33 in the timing diagram 1000, the controller 140 or another suitable entity precharges the voltage of the shunt capacitor C19 (associated with the sensing circuit arrangement 145) to a known value. Any suitable precharge value can be used, which may include shorting node N22 to node N3.

[0179] Since the magnitude of the AC input voltage Vin is positive, the controller 140, as shown in timing diagram 1000, activates both switches Q11 and Q21 to an ON state. The controller 140 alternately switches between activating switch Q12 and switch Q22 to provide power factor correction, as discussed herein.

[0180] Between time T32 and time T33, the controller 140 sets switch Q22 to an ON state and switch Q12 to an OFF state. In this case, the energy stored in transformer winding 161-1 is discharged based on the generation of the output voltage Vout. In other words, the energy in transformer 161-1 decreases.

[0181] Shortly after or around time T33, the controller 140 generates the control signal S12 to activate switch Q12 to an ON state. The controller 140 generates the control signal S22 to deactivate switch Q22 to an OFF state. Thus, at or around time T33, energy input (energy into the transformer winding 161-1) begins, increasing the magnitude of the current 151 supplied by the input voltage Vin through capacitor C19 and the transformer winding 161-1.

[0182] It should be noted that the primary current also magnetizes current 167, since transformer 161 is used as an inductor.

[0183] Due to the increased flow of current 151 (resonant current) from the input voltage source 120 through the combination of switch Q11 and switch Q12, capacitor C19, and transformer winding 161-1, starting at time T33, the voltage V19 (feedback 725) across the shunt capacitor C19 drops from its initial pre-charged value of 1 volt or another suitable value. In other words, the magnitude of the feedback signal 725 decreases proportionally to the amount of initial energy stored in the transformer winding 161-1.

[0184] As previously discussed, the detection circuit arrangement 145 implements ( Fig. 7) and the corresponding controller 140 uses the comparator 720 to compare the feedback 725 (or the voltage V19 from capacitor C19) with the threshold signal TS, as generated by the threshold level generator 715. When the comparator 720 detects that the magnitude of the feedback 725 falls below the threshold signal TS (such as -40 mV or another suitable value below the pre-charge voltage) at or about time T34, the comparator 720 generates the trigger signal 755, which notifies the controller 140 of the trigger event (the magnitude of the feedback 725 falls below the threshold signal TS). Responding to this trigger event, as indicated by the trigger signal 755, the controller 140 disables switch Q12 and enables switch Q22 at or about time T34.

[0185] The deactivation of switch Q12 at or around time T34 ends the energy absorption phase (such as energy received from the input voltage Vin to transformer winding 161-1) and begins the energy transfer phase (energy in transformer winding 161-1 being transferred to transformer winding 161-2 to generate a corresponding output voltage Vout).

[0186] Due to the blocking capacitors C2 and C3, most of the current flowing from the shunt capacitor flows back inwards. At the end (around time T36) of the energy transfer cycle (where the cycle lies between time T33 and time T36), the voltage V19 across the shunt capacitor C19 is almost back to its initial charge value of approximately 1 volt.

[0187] In one example, resistor R1 provides drift cancellation with respect to capacitor C19, although any suitable drift capacitor circuit can be used.

[0188] Accordingly, one concept, as discussed herein, is that the voltage V19 (such as the feedback 725) can be used to determine Delta Q, which can be translated into an average of the current 151 and the corresponding initial energy during the recording (such as between time T33 and time T34) and (for a hybrid blocking operation) also into an average current during one period (Tsw), such as between time T33 and time T36. The power converter, as discussed herein, can be configured to match the average current and average voltage during a switching period by a constant (feedback-dependent, no gain at 50 / 60 Hz), resulting in power factor correction.

[0189] Thus, the feedback signal 725 (the high-frequency signal with respect to the frequency of the input voltage Vin), generated by the sensing circuit arrangement 145, is a sensing signal used for DeltaQ control. The threshold signal TS is the threshold against which the signal 725 is compared. When both are identical (crossing), as detected by the comparator 720, the controller 140 stops drawing energy from the input voltage Vin to the transformer winding 161-1 at or around times T34, T37, T38, etc. DeltaQ can be above or below the threshold, particularly during the energy transfer phase.

[0190] It is further noted that the magnitude of the feedback signal 725 can be directly proportional to the magnitude of the energy transferred from the input voltage Vin to the transformer winding 161-1, wherein the magnitude of the feedback signal 725 is configured to increase over time rather than decrease over time. In such an instance, the comparator 720 or another suitable entity can be configured to generate the trigger signal 755 in response to the detection that the magnitude of the feedback signal 725 is greater than the threshold signal TS.

[0191] Accordingly, as previously discussed, the controller 140, as discussed herein, can be configured to continuously adjust the operation of the switching circuit arrangement (i.e., Q11, Q12, Q21, Q22) by generating a threshold signal TS based on at least the magnitude of the output voltage Vout and potentially other parameters (on an as-needed basis, one cycle at a time). The comparator 720 compares the feedback signal 725 (which indicates the magnitude of the first energy transferred to the transformer winding 161-1), generated by the sensing circuit 145, with the threshold signal TS. Based on the comparison and the detection of a tripping event, such as the magnitude of the feedback signal 725 falling below the threshold signal TS, the controller 140 operates the switching circuit arrangement to terminate the flow of the first current 151 through the first transformer winding 161-1.

[0192] In one example, the magnitude of the threshold signal TS varies over time. Another example of this is found particularly in Fig. 11 shown.

[0193] With renewed reference to Fig. 10 and as previously discussed, the threshold level signal generator 715 can be configured to generate the threshold signal TS based on a combination of one or more of: i) a magnitude of the input voltage Vin, ii) a capacitance CSC associated with a sensing circuit arrangement 145 that generates the feedback signal 725, iii) a filtered fault voltage signal K derived based on a difference between a magnitude of the output voltage Vout and a set reference voltage 729 (i.e., the filtered fault voltage signal K is derived by the fault voltage signal generator 785 based on the fault signal 736, which is generated based on a difference between the magnitude of the output voltage Vout and the set reference voltage 729), and iv) the switching period Tsw of operating the respective switching circuit arrangement.

[0194] It should be noted that the switching frequency associated with generating the respective control signals S11, S12, S21, and S22 can vary over time. As discussed previously, the duty cycle associated with activating the respective high-side switch circuit arrangement Q12 can also vary over time based on the magnitude of the threshold signal TS (threshold level, which varies to provide control and power factor correction).

[0195] In one example, as discussed previously, the controller 140, responding to the detection that the magnitude of the feedback signal 725 falls below the threshold level TS, sets the respective switch Q12 to an OFF state, so that the input current iIN from the voltage source 120 no longer flows through the switches and through the transformer winding 161-1 (the magnitude of the current 151 no longer increases). The residual energy stored in the primary winding 161-1 is transferred to the secondary transformer winding 161-2 to generate the output voltage Vout.

[0196] As discussed previously, the threshold signal TS can change slowly with a dependence on the input voltage (Vin) while simultaneously compensating for a change in the period (Tsw).

[0197] In one example, the offset value OSV, which is associated with the threshold signal TS, is used to keep the value TS within the range of the comparator 720. In other words, the offset OSV can be implemented to prevent the threshold signal TS (threshold level) from exceeding the comparator(720) rails (1 V in sim, no effect on the control), where the value K is the feedback input (for example, it could be an equivalent input resistance enforced by the control – changing it so that it alters the average power consumption) and the capacitance C is the gain of the sensing capacitor network in coulombs.

[0198] Fig. Figure 11 is an exemplary timing diagram illustrating the control of a respective power converter as discussed herein.

[0199] The timing diagram 1100 illustrates an example of a sense voltage (Vsence or feedback 725, generated by the sensing circuit 145) and a delta-U control. In this example, the voltage Vsence (e.g., feedback 725) is the sensing voltage feedback generated by a shunt capacitor or a capacitive divider circuit (such as the sensing circuit 145) that monitors a quantity of the current 151 or the energy supplied to or through the primary winding 161-1.

[0200] In one example, at the beginning of the recording (i.e., end of the transmission), such as at time T92, the voltage signal Vsense and the corresponding detection circuit 145 are initialized to a known value.

[0201] It should be noted that a certain drift in the initial condition (precharge) associated with the detection circuit 145 is to be expected. Without resetting the detection circuit 145 and the corresponding capacitors, the U1 value will be lost over time.

[0202] As further shown, a next cycle is started when (such as time T91, time T93, etc.) Vsense (feedback 725) is equal to, crosses, or falls below the threshold control line (threshold signal TS) at voltage U2. For simplicity, as discussed previously, an offset OSV can be added to the control signal (TS). The offset value OSV associated with the threshold signal TS has no effect on Delta U, as the offset is canceled out. As discussed previously, using the offset value OSV eliminates the need for an additional set of comparator voltage rails to operate the comparator 720.

[0203] Thus, the time diagram illustrates 1100 in Fig. 11, how the size of the threshold signal TS varies over time from one cycle to the next, and how the size of the feedback 725 generates the respective trigger signal for starting a new cycle, as previously discussed.

[0204] Fig. Figure 12 is an exemplary circuit diagram illustrating a power converter circuit and a corresponding primary winding power / energy flow sensing system, as discussed herein.

[0205] Depending on the control position, as shown in the power supply 600-12, an additional winding 1110 and capacitor C93 can be implemented to bring the sensing voltage (feedback signal 725 generated by the sensing circuit 145-1) to the same reference as the control 140. This can be useful for bridgeless HBF (hybrid blocking) configurations due to the floating placement of the control 140.

[0206] Fig. Figure 13 is an exemplary diagram illustrating a detection circuit configured to monitor a quantity of current (first energy) through a given primary transformer winding, as discussed herein.

[0207] In this example, the power supply 600-13 is essentially identical to the power supply 600, as discussed previously. However, the controller 140 controls the operation of switches 131 and 132 to transfer the initial energy from the input voltage Vin (from node N1) to the first transformer winding 161-1, which is magnetically coupled to the transformer winding 161-2.

[0208] The sensing circuit 1301 (such as a capacitor divider circuit), implemented in this example to monitor a quantity of energy supplied to the transformer winding 161-1, comprises several capacitors C201 and C202 arranged in series between node N6 and node N2. The combination of capacitors C201 and C202 is arranged in parallel with the resonant capacitor C3.

[0209] Furthermore, in this example, node N210, which couples capacitors C201 and C202, generates the respective feedback signal 725, which indicates a quantity of the first energy P1 that is supplied to the transformer winding 161-1 via the control of switches 131 and 132.

[0210] In an example, as discussed previously, the power supply 600 (and any corresponding instantiations such as 600-1, 600-2, 600-3, etc.) are implemented as blocking power converters that operate in a resonant mode via the controlled switching of the respective switches 131 and 132, as implemented by the controller 140.

[0211] In one example, the controller 140 controls the flow of the first energy P1 through the first winding 161-1 by controlling a resonant current (such as current 151) flowing from the input voltage Vin through the first transformer winding 161-1. As discussed herein, the controlled flow of the resonant current (such as current 151) through the first transformer winding 161-1 provides power factor correction, so that an average magnitude (such as over one or more cycles of the input voltage) of the first energy delivered from the input voltage to the first transformer winding is essentially equal to an average magnitude of the second energy delivered from the second winding to the respective load.

[0212] Thus, charge control by the transformer winding 161-1 and power factor correction, as discussed herein, is an option for implementing any resonant power converter topology to connect the resonant capacitor voltage with the transfer of energy from a transformer winding 161-1 to the transformer winding 161-2.

[0213] During the high-side ON time (such as the activation of switch 131), the voltage difference across the capacitor is proportional to the input current. Δu=U_end−U_start=1 / C∫_(T_intk) 〚I_intk dt〛=I_avg / C

[0214] By controlling DeltaV via the resonant capacitor C3, the average input current is controlled, thus providing the greatest benefit of current-mode control with resistance to reactive / circulating current.

[0215] Fig. Figure 14 is an exemplary diagram illustrating a detection circuit configured to monitor a quantity of current (first energy) through a given primary transformer winding, as discussed herein.

[0216] In this example, the power supply 600-14 is essentially identical to the power supply 600, as discussed previously. However, in this example, the controller 140 controls the operation of switches 131 and 132 to transfer the initial energy from the input voltage Vin to the first transformer winding 161-1, which is magnetically coupled to the transformer winding 161-2.

[0217] The sensing circuit 1401 (such as a shunt capacitor sensing circuit), implemented to monitor a quantity of energy supplied to the transformer winding 161-1, includes a capacitor C211 arranged in series with capacitor C3 between node N6 and node N2. Node N220, which couples capacitors C3 and C211, generates the respective energy feedback signal (such as feedback 725), indicating a quantity of the corresponding initial energy supplied to the transformer winding 161-1 by the input voltage Vin via the control of switches 131 and 132.

[0218] Fig. Figure 15 is an exemplary diagram illustrating a detection circuit configured to monitor a quantity of current (first energy) through a given primary transformer winding, as discussed herein.

[0219] In this example, the power supply 600-15 is essentially identical to the power supply 600, as discussed previously. However, in this example, the controller 140 controls the operation of switches 131 and 132 to transfer the initial energy from the input voltage Vin to the first transformer winding 161-1, which is magnetically coupled to the transformer winding 161-2.

[0220] As shown, the sensing circuit 1501 (such as via a floating reference sensing circuit) is implemented to monitor a quantity of the power supply 600-3 to the transformer winding 161-1.

[0221] Fig. 16 is an exemplary method for controlling a respective resonant power converter to provide a power factor correction as discussed herein.

[0222] In processing operation 1810, the controller 140, via a control of a switching circuit arrangement, such as switches Q11, Q12, Q21, Q22, controls a flow of first energy (such as based on the resonant current 151 or iPRI), which is received from an input voltage Vin and transmitted through a first transformer winding 161-1, which is magnetically coupled to a second transformer winding 161-2. The controlled flow of the first energy (such as the resonant current 151) through the first transformer winding 161-1 to the second transformer winding 161-2 results in the generation of an output voltage Vout based on a second energy (first power converter to the second energy), which is delivered from the second transformer winding 161-2 to a load 118.

[0223] In the processing operation 1820, the controller 140 receives feedback, such as 725 (such as a voltage V19 across capacitor C19 or feedback from any of the different sensing circuit arrangements 145, as discussed herein). In one example, the feedback 725, multiplied by the input voltage, provides a quantity of the first energy transferred through the first windings 161-1 for a duration such as between time T33 and time T34.

[0224] In processing operation 1830, the controller 140 sets the operation of the switching circuit arrangement (switches Q11, Q12, Q21, Q22) over time based on at least the feedback (which specifies the initial energy P1) and a magnitude of the output voltage Vout. The set operation of the switching circuit arrangement, such as the termination of the activation of the high-side switching circuit arrangement Q12, sets a magnitude of the initial energy P1 that is delivered to the first transformer winding 161-1 in each cycle. In particular, the activation of the switching circuit arrangement at time T34, as discussed above, prevents further energy P1 from being delivered to the transformer winding 161-1.

[0225] In an example, as discussed above, the controlled flow of the first energy through the first winding 161-1 involves controlling a resonant current (such as current 151) flowing from the input voltage Vin through the first transformer winding 161-1. The controlled flow of the resonant current through the first transformer winding provides power factor correction associated with the conversion of the input voltage Vin to the output voltage Vout by the blocking power converter configuration, as shown in the power converter 600 and other power converters, as discussed above.

[0226] It should be noted again that the techniques described herein are well suited for use in power supply applications. However, it should be noted that the examples given here are not limited to use in such applications and that the techniques discussed herein are also well suited to other applications.

[0227] Although this invention has been shown and described with particular reference to preferred examples thereof, those skilled in the art will understand that various modifications in form and details can be made to it without departing from the spirit and scope of the present application as defined by the appended claims. Such variations are intended to be covered by the scope of this present application. Thus, the foregoing description of examples of the present application is not intended to be limiting. Rather, any limitations of the invention are presented in the following claims.

Claims

[1] Device having the following features: a first winding (161-1); a first bidirectional switch circuit arrangement (131); a second bidirectional switch circuit arrangement (132) arranged in series with the first bidirectional switch circuit arrangement (131); and a combination of the first bidirectional switching circuit arrangement (131) and the second bidirectional switching circuit arrangement (132) which is effective to control a magnitude of current through the first winding (161-1) to generate an output voltage. [2] Device according to claim 1, further comprising the following feature: a capacitor arranged in a series path containing the first winding (161-1), wherein the series path is effective to assist a resonance of the current through the first winding (161-1). [3] Device according to claim 1 or 2, further comprising the following feature: a transformer (161) comprising the first winding (161-1) and a second winding (161-2), wherein the second winding (161-2) is magnetically coupled to the first winding (161-1) in the transformer (161). [4] Device according to any one of claims 1 to 3, wherein the first bidirectional switching circuit arrangement (131) is effective in blocking the passage of current in both a first direction and a second direction through the first bidirectional switching circuit arrangement (131); and wherein the second bidirectional switching circuit arrangement (132) is effective in blocking the passage of current in both a first direction and a second direction through the second bidirectional switching circuit arrangement (132). [5] Device according to any one of claims 1 to 4, wherein the first bidirectional switching circuit arrangement (131) is a first GaN (gallium nitride) switch; and wherein the second bidirectional switching circuit arrangement (132) is a second GaN (gallium nitride) switch. [6] Device according to any one of claims 1 to 5, wherein the first bidirectional switch circuit arrangement (131) comprises a first switch which is arranged in series with a second switch; wherein the second bidirectional switch circuit arrangement (132) comprises a third switch which is arranged in series with a fourth switch; wherein a combination of the first switch and the second switch is arranged in a first series path between a first node and a second node; and wherein a combination of the third switch and the fourth switch is arranged in a second series path between the second node and a third node. [7] Device according to claim 6, further comprising the following feature: a control (140) that is effective to: during a first mode in which an input voltage between the first node and the third node is positive: i) activate both the first switch and the third switch to ON states, and ii) alternately switch between activating the second switch and the fourth switch to an ON state at a frequency and first duty cycle to control the magnitude of the current through the first winding (161-1) so that the output voltage is regulated to a desired setpoint. [8] Device according to claim 7, wherein the control (140) further acts to: during a second mode in which the input voltage between the first node and the third node is negative: i) activate both the second switch and the fourth switch to ON states, and ii) alternately switch between activating the first switch and the third switch to an ON state at frequency and a second duty cycle to control the magnitude of the current through the first winding (161-1) so that the output voltage is regulated to the desired setpoint. [9] Device according to claim 8, further comprising the following feature: a transformer (161) comprising the first winding (161-1) and a second winding (161-2), wherein the second winding (161-2) is magnetically coupled to the first winding (161-1) in the transformer (161); wherein a current flows through the first winding (161-1) to generate the output voltage, the output voltage being output from the second winding (161-2); and wherein the control (140) is effective to: i) select between the first mode and the second mode based on a polarity of the input voltage, and ii) regulate a magnitude of the output voltage by alternately switching between operation in the first mode and the second mode. [10] Device according to any one of claims 1 to 9, wherein the first bidirectional switching circuit arrangement (131) is a first dual-gate switch comprising a first drain node, a first source node, a first gate node and a second gate node; and wherein the second bidirectional switching circuit arrangement (132) is a second dual-gate switch comprising a second drain node, a second source node, a third gate node and a fourth gate node. [11] Device according to claim 10, further comprising the following feature: a control (140) that is effective to: during a first mode in which an input voltage between the first node and the third node is positive: i) apply ON control signals to the first gate node and the third gate node, and ii) switch between applying an ON control signal to the second gate node and the fourth gate node to control the magnitude of the current through the first winding (161-1). [12] Device according to claim 11, wherein the control (140) is further effective in: during a second mode in which an input voltage between the first node and the third node is negative: i) to apply ON control signals to the second gate node and the fourth gate node, and ii) to switch between applying an ON control signal to the first gate node and the third gate node to control the magnitude of the current through the first winding (161-1). [13] Device according to claim 12, wherein the control (140) is further effective in regulating the magnitude of the output voltage by switching between operation in the first mode and the second mode. [14] Device according to any one of claims 1 to 13, which further comprises the following features: a first capacitor arranged in parallel to a combination of the first bidirectional switch circuit arrangement (131) and the first winding (161-1); and a second capacitor arranged in parallel to a combination of the second bidirectional switch circuit arrangement (132) and the first winding (161-1). [15] Device according to any one of claims 1 to 14, further comprising the following feature: a first capacitor; and wherein the first bidirectional switch circuit arrangement (131) is directly connected to the second bidirectional switch circuit arrangement (132) via a first node, wherein the first capacitor is arranged in series between the first node and the first winding (161-1). [16] Device according to claim 15, further comprising the following feature: a control (140) that is effective to control the first bidirectional switch circuit arrangement (131) and the second bidirectional switch circuit arrangement (132) based on a feedback signal generated at a second node that directly connects the first capacitor and the first winding (161-1). [17] Device according to any one of claims 1 to 16, further comprising the following feature: a first resonant capacitor; a second resonant capacitor; wherein the first winding (161-1) is connected between a first node and a second node, wherein the first node directly couples the first bidirectional switch circuit arrangement (131) and the second bidirectional switch circuit arrangement (132), and wherein the second node directly couples the first resonant capacitor and the second resonant capacitor; and a controller (140) that is effective to control the first bidirectional switching circuit arrangement (131) and the second bidirectional switching circuit arrangement (132) based on a feedback signal received from the second node. [18] Method comprising the following steps: Control of the operation of the first bidirectional switch circuit arrangement (131); Controlling the operation of the second bidirectional switching circuit arrangement (132), which is arranged in series with the first bidirectional switching circuit arrangement (131); and wherein the controlled operation of the first bidirectional switching circuit arrangement (131) and the second bidirectional switching circuit arrangement (132) controls a magnitude of a current through a first winding (161-1) to generate an output voltage. [19] Method according to claim 18, wherein the first bidirectional switch circuit arrangement (131) comprises a first switch arranged in series with a second switch; wherein the second bidirectional switch circuit arrangement (132) comprises a third switch which is arranged in series with a fourth switch; wherein a combination of the first switch and the second switch is arranged in a first series path between a first node and a second node; and wherein a combination of the third switch and the fourth switch is arranged in a second series path between the second node and a third node, wherein the method further comprises the following steps: via a control (140): during a first mode in which an input voltage between the first node and the third node is positive: i) activating both the first switch and the third switch to ON states, and ii) switching between activating the second switch and the fourth switch to an ON state to control the magnitude of the current through the first winding (161-1). [20] The method according to claim 19, further comprising the following step: during a second mode in which the input voltage between the first node and the third node is negative: i) activating both the second switch and the fourth switch to ON states, and ii) switching between activating the first switch and the third switch to an ON state to control the magnitude of the current through the first winding (161-1).

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