Transmitter device, noise predistortion and decorrelation processes in such a transmitter device

Abstract

The invention relates to a multi-channel transmitter device (10) comprising on each channel: - a transmission processing channel adapted to process an input signal and comprising a predistortion block (11) applying predistortion to said input signal as a function of predistortion coefficients, a DAC (13), a first analog filter (14), a power amplifier (16) and a sigma-delta encoder (12) between the predistortion block and the digital-to-analog converter and adapted to perform, in particular, quantization of the predistortion block (11); - a return channel associated with the transmission processing channel and comprising a predistortion coefficient estimation block (19) for estimating, as a function of said input signal and a return signal, predistortion coefficients; said transmitter device (10;50) being characterized in that the predistortion coefficient estimation block (19) is adapted to estimate said predistortion coefficients as a function of at least said input signal and a signal resulting from the subtraction, said return signal, of a signal representing the quantization noise resulting from the quantization performed by the sigma-delta encoder (12). Figure for the abstract: Fig. 6;

Description

Title of the invention: Transmitting device, methods of predistortion and noise decorrelation in such a transmitting device Technical field

[0001] The invention lies in the field of signal transmission devices comprising a power amplifier and performing a digital predistortion operation to compensate for the non-linearity of this power amplifier. Prior art

[0002] Predistortion is a known operation (see for example H. Qian, H. Huang and S. Yao, "A General Adaptive Digital Predistortion Architecture for Stand-Alone RF Power Amplifiers," in IEEE Transactions on Broadcasting, vol. 59, no. 3, pp. 528-538, Sept. 2013), which consists of compensating for the non-linearity of the amplifier by adding to the signal presented at the input of the latter an inverse non-linearity, so that a linear transfer function is obtained for the assembly comprising the predistortion module and the amplifier.

[0003] [Fig.l] represents a block diagram which illustrates an example of a transmitter device architecture with known predistortion. It comprises a transmission processing channel receiving the useful signal in baseband and, which, on each of two processing chains I, Q intended to process one the in-phase signal (I) selectively and the other the quadrature phase signal (Q) selectively, comprises a digital predistortion block followed by a digital-to-analog converter (DAC), an analog filter to suppress noise, a frequency change stage (optional) to transpose the signal onto a carrier (only one processing chain is represented in [Fig.l]). The processing channel finally comprises a power amplifier (PA).

[0004] The digital predistortion block performs a predistortion operation based on predistortion coefficients.

[0005] The transmitter device further comprises a return path which makes it possible to determine the effects of the variation of the gain of the power amplifier as a function of the power of the digital signal supplied at the input of the transmission processing path. A coupler (or any other measuring device) is used at the output of this power amplifier to obtain the signal at the output of the power amplifier. The return path comprises a frequency change stage (optional) to transpose the signal into baseband, and on each of two processing chains intended to process one the in-phase signal (I) selectively and the other the quadrature phase signal (Q) selectively (only one of these chains is shown in [Fig.l]), an analog filter, an analog-to-digital converter (ADC), a predistortion coefficient estimation block, adapted to carry out this estimation as a function of at least the signal at the output of the converter and the useful signal at the input of the transmission processing channel. The updated predistortion coefficients are then delivered to the predistortion block (of chain I, respectively Q), which uses them to carry out the following predistortions.

[0006] The non-linearities of the power amplifier create harmonics of the signal which are represented in [Fig.2] illustrating the spectrum of the signal at the output of the amplifier. The useful signal power, supplied at the input of the processing channel in transmission, is represented as a function of the frequency in hatching. The signal has a bandwidth + / - B with respect to the central frequency. The non-linearity conferred by the amplifier is often modeled by a transfer function of the power amplifier in polynomial form. To simplify the explanation, we assume here that this polynomial is of order r, with for example r=5. This non-linearity of the power amplifier will therefore generate noise in the band (dotted in [Fig.2]) which will limit the performance of the transmitting device.It will also generate intermodulation products of the input signal which will be outside the useful signal band: in the example considered with r=5, from B to 3B for harmonics of order 3 and from 3B to 5B for harmonics of order 5.

[0007] Many algorithms can be used to extract these nonlinearities. One solution is to focus on the signal outside the bandwidth of the input signal (in the case r= 5: in the bands [B,3B] for the 3rd harmonic and [3B,5B] for the 5th harmonic): see for example DG Pham, P. Desgreys, MA Hussein, P. Loumeau and O. Venard, "Impact of subband quantization on DPD correction performance," 2013 IEEE 20th International Conference on Electronics, Circuits, and Systems (ICECS), 2013).

[0008] The advantage is that in these bandwidths, the intermodulation products, resulting from the distortion of the signal, are isolated from the input signal, located at a lower frequency. The analysis of these different bands will also make it possible to dissociate the different coefficients of the non-linear model.

[0009] The solution described here is a conventional approach which has the disadvantage of potentially requiring a DAC with a large number of bits (so as not to degrade the signal-to-noise ratio of the in-band and out-of-band signal). This high number of bits is inconvenient because it involves significant complexity for the DAC (which will generate a large surface area and high consumption) and, in the case where the DAC and the digital circuit for generating the modulated signals are separate, it is necessary to have numerous digital outputs at the output of the digital data modulation circuit and numerous inputs on the circuit containing the DAC. This phenomenon is even more problematic in multi-channel transmitter devices since the number of bits in the DAC must be multiplied by the number of channels to obtain the number of bits to be transferred.

[0010] Il est connu de positionner un codeur sigma-delta en amont du DAC : cf. H. H. Boo, S. W. Chung and J. L. Dawson, "Adaptive Predistortion Using a $\Delta\Sigma$ Modulator for Automatic Inversion of Power Amplifier Nonlinearity," in IEEE Transactions on Circuits and Systems II: Express Briefs, vol. 56, no. 12, pp. 901-905, Dec. 2009.

[0011] This solution makes it possible to reduce the number of bits of the DAC in return for having a high sampling frequency to benefit from a good oversampling factor which is necessary for sigma-delta coding. This constraint is not necessarily a problem in the architecture of the transmitting device since it is usual to use a high sampling frequency in the transmitting DAC so as not to overly constrain the analog filter following this same DAC. In the approach proposed by Boo et al., it is shown that the use of the sigma-delta coder upstream of the DAC also makes it possible to simplify the implementation of the predistortion by avoiding the need to invert the transfer function of the amplifier. It makes it possible to use a simple look-up table to model the non-linear behavior of the DAC to compensate for that of the PA.The main drawback of this architecture is that the quantization noise due to sigma-delta coding, whose power increases as a function of frequency from the end of the signal bandwidth, must be filtered by an analog filter with a low cutoff frequency, which places strong constraints on the analog filter. Moreover, in the example considered, there is no predistortion actually implemented for the amplifier. In the case where one wanted to use predistortion, it would potentially be necessary to increase the bandwidth of the DAC and therefore increase the sampling frequency accordingly to keep a constant oversampling factor.

[0012] [Fig. 3] illustrates the spectrum of the quantization noise brought by a sigma delta coder when it is positioned upstream of the DAC in a transmitter device of the type in [Fig. 1], as mentioned above, but without predistortion operation. The hatched line shows the spectrum of the useful signal, the curve Q1 represents the quantization noise generated by the sigma-delta coder (low in the +B / -B band), the dashed curve FAI represents the analog filtering which follows the DAC, which as in the example given by Boo et al., eliminates the quantization noise outside the +B / -B bandwidth.

[0013] In the example of [Fig.4], predistortion is implemented, the non-linearity noise of the amplifier is shown in dotted lines: the quantization noise Q2 generated by the sigma-delta encoder must be attenuated in a wider bandwidth, so the required sampling frequency will be higher, which is pro- problematic when the input signal bandwidth is large. Summary of the invention

[0014] To this end, according to a first aspect, the present invention describes a transmitter device comprising N channels Vi, i = 1 to N > 1, each of said channels Vi comprising

[0015] - a transmission processing channel adapted to process an input signal and comprising a predistortion block, a digital-to-analog converter, a first analog filter and a power amplifier, said transmitter device comprising a transducer device coupled to said processing path and receiving the input signal once processed by said processing path, the predistortion block being adapted to apply a predistortion to said input signal, according to predistortion coefficients, to compensate for the non-linearities of the power amplifier;

[0016] - a return channel associated with the transmission processing channel, adapted for receiving the signal processed by the processing channel and comprising a second analog filter, an analog-to-digital converter, and a pre-distortion coefficient estimation block, said distortion coefficient estimation block being adapted to estimate, as a function of at least said input signal and a signal, called the return signal, obtained following at least the successive operations, on the received processed signal, of filtering by the second filter and conversion by the analog-to-digital converter, pre-distortion coefficients and to deliver said estimated pre-distortion coefficients to the pre-distortion block;

[0017] in which the processing channel further comprises a sigma-delta coder between the predistortion block and the digital-to-analog converter, adapted to carry out in particular a quantization;

[0018] said transmitting device being characterized in that the predistortion coefficient estimation block is adapted to estimate said predistortion coefficients as a function of at least said input signal and a signal resulting from the subtraction, from said return signal, of a signal representing the quantization noise resulting from the quantization carried out by the sigma-delta coder.

[0019] The invention thus proposes to cancel the quantization noise generated by a sigma delta coder arranged upstream of the DAC in the bands outside the B / -B band of the useful input signal (from B to 5B in the example mentioned above with r= 5 or beyond depending on the predistortion) at the level of the feedback loop. Thus, a moderate sampling frequency of the DAC can be maintained.

[0020] In embodiments, such a device will further comprise at least one of the following features:

[0021] - said signal representing the quantization noise is a function of said quantization noise fication shaped by the NTF(z) transfer function of the sigma-delta encoder noise;

[0022] - the wireless transmitting device is adapted to obtain said signal representing the quantization noise by performing at least one of said operations among:

[0023] - filtering and decimation to adapt the sampling frequency and the signal bandwidth representing quantization noise to the input signal

[0024] - filtering to reproduce at least the effect of the first and second filters;

[0025] - FIR type filtering to reproduce the effect of the first and second filters, the transfer function of said FIR filtering being estimated as a function of the comparison between said return signal and a signal obtained from the quantization noise extracted from the sigma-delta coder;

[0026] N>1;

[0027] - the size of data paths in the pre coefficient estimation block distortion in each channel is oversized, to introduce additional nonlinearities; and / or

[0028] - the transmitting device comprises a control block adapted for, in a phase prior transient, trigger the supply of deterministic signals to the input of the digital-to-analog converters (13), such that given any two distinct channels, the amplitude and / or frequency of these deterministic signals is different between said channels.

[0029] According to another aspect, the invention describes a predistortion method in a transmitter device comprising N channels Vi, i = 1 to N > 1, each of said channels Vi comprising:

[0030] - a transmission processing channel adapted to process an input signal and comprising a predistortion block, a digital-to-analog converter, a first analog filter and a power amplifier, said transmitter device comprising a transducer device coupled to said processing path and receiving the input signal once processed by said processing path, the predistortion block being adapted to apply a predistortion to said input signal, according to predistortion coefficients, to compensate for the non-linearities of the power amplifier;

[0031] - a return channel associated with the transmission processing channel, adapted for receiving the signal processed by the processing channel and comprising a second analog filter, an analog-to-digital converter, and a pre-distortion coefficient estimation block;

[0032] in which the processing path further comprises a sigma-delta encoder between the predistortion block and the digital-to-analog converter, adapted to perform including quantification;

[0033] said method comprising a step of estimating the pre-distortion coefficients by said distortion coefficient estimation block, as a function of at least said input signal and a signal, called return signal, obtained following at least the successive operations, on the received processed signal, of filtering by the second filter and of conversion by the analog-digital converter, said estimated pre-distortion coefficients being delivered to the pre-distortion block;

[0034] said method being characterized in that the predistortion coefficient estimation block (estimates said predistortion coefficients as a function of at least said input signal and a signal resulting from the subtraction, from said return signal, of a signal representing the quantization noise resulting from the quantization carried out by the sigma-delta coder.

[0035] In embodiments, such a method will further comprise at least one of the following features:

[0036] - said signal representing the quantization noise is a function of said quantization noise fication shaped by the NTF(z) transfer function of the sigma-delta encoder noise;

[0037] - said signal representing the quantization noise is obtained by carrying out at least one of the said operations among:

[0038] - filtering and decimation to adapt the sampling frequency and the signal bandwidth representing quantization noise to the input signal;

[0039] - filtering to reproduce at least the effect of the first and second filters;

[0040] - FIR type filtering to reproduce the effect of the first and second filters, the transfer function of said FIR filtering being estimated as a function of the comparison between said return signal and a signal obtained from the quantization noise extracted from the sigma-delta coder.

[0041] According to another aspect, the invention describes a method for decorrelating quantization noise in a transmitting device, according to which a predistortion method according to the invention is implemented on each of the channels Vi, i = 1 to N, with N integer >1.

[0042] In embodiments, this method comprises the following steps:

[0043] - the size of data paths in the pre coefficient estimation block distortion in each channel is oversized, to introduce additional nonlinearities; and / or

[0044] - in a prior transient phase, deterministic signals are provided to the input of the digital-to-analog converters, such that given any two distinct channels, the amplitude and / or frequency of these deterministic signals is different between said channels. Brief description of the drawings

[0045] The invention will be better understood and other characteristics, details and advantages will appear more clearly on reading the following description, given without limitation, and thanks to the appended figures, given by way of example.

[0046] [Fig-1] [Fig.l] is an illustration of an architecture of a transmitter device of prior art performing predistortion processing;

[0047] [Fig.2] [Fig.2] is a schematic representation of the output signal spectrum of the power amplifier in a transmitting device exhibiting distortion;

[0048] [Fig.3] [Fig.3] is a schematic representation of the output signal spectrum of the power amplifier in a transmitter device comprising a sigma-delta encoder and a distortion-free noise shaping filter;

[0049] [Fig.4] [Fig.4] is a schematic representation of the output signal spectrum of the power amplifier in a transmitter device comprising a sigma-delta encoder and a noise shaping filter and performing predistortion processing;

[0050] [Fig.5] [Fig.5] is a schematic representation of the output signal spectrum of the power amplifier in a transmitter device comprising a sigma-delta encoder and a noise shaping filter and performing predistortion processing in one embodiment of the invention;

[0051] [Fig.6] [Fig.6] is an illustration of an architecture of a transmitter device ef performing predistortion processing in one embodiment of the invention;

[0052] [Fig.7] [Fig.7] is an illustration of a block for estimating the coefficients of com thinking in an embodiment of the invention;

[0053] [Fig.8] [Fig.8] illustrates part of a block for estimating the coefficients of com thinking in an embodiment of the invention;

[0054] [Fig.9] [Fig.9] is an illustration of an architecture of an ef transmitter device performing predistortion processing in one embodiment of the invention;

[0055] [Fig. 10] [Fig. 10] is an illustration of the steps of a method in one embodiment of the invention.

[0056] Identical references may be used in different figures when they designate identical or comparable elements. Description of the embodiments

[0057] In [Fig.6], a block diagram of a transmitter device 10 is shown in one embodiment of the invention, in a non-limiting example a wireless telecommunications transmitter device, here radiofrequency (it will be noted in particular that the possible I / Q channels of quadrature signals are not shown for the sake of clarity).

[0058] The transmitter device 10 comprises at least one transmission processing channel and a return channel associated with this transmission processing channel.

[0059] The transmission processing channel comprises, from the input to the output of the transmission processing channel, on each of the two processing chains I, Q (only one chain being represented): a digital predistortion block (DPD) 11, a sigma-delta encoder 12, a digital-to-analog converter (DAC) 13, a low-pass filter 14, a frequency change stage (up-conversion) 15_1. The processing channel then comprises, downstream of the two processing chains I, Q, a power amplifier (PA) 16. It is adapted to obtain at the output of the PA the signal to be transmitted, intended here for an RF antenna.

[0060] The return path associated with this transmission processing path comprises, from its input to its output, a frequency change stage (down-conversion) 15_2, then, on each of two parallel processing chains I, Q (only one chain being shown), a low-pass filter 17, an analog-to-digital converter (ADC) 18. The return path finally comprises a block for estimating the predistortion coefficients 19, receiving as input the outputs of the ADCs 18.

[0061] The device 10 comprises, in the present case, an optional noise shaping block 20, arranged between the sigma-delta coder 12 and the estimation block 19 and to which the quantization noise of the sigma-delta coder 12 is applied.

[0062] The return path is adapted to obtain the signal at the output of the PA 16 and determine the predistortion coefficients defining the non-linearity as a function of said signal at the output of the PA 16.

[0063] Block 15 comprising the frequency changing stages 15_1 and 15_2 is also optional.

[0064] The invention comprises the subtraction of the quantization noise extracted from the sigma_delta coder 12 obtained for a digital signal at the input of the processing channel, named input_bb, from the signal delivered at the output of the PA 16 for this same input signal input_bb, then the non-linearities of the PA 16 as a function of the power are evaluated by the estimation block 19 by comparing the result of this subtraction (i.e. the signal at the output of the PA 16 corresponding to the input signal input_bb minus the quantization noise) and the input signal input_bb.

[0065] By thus canceling the quantization noise in the estimation block 19, the constraints on the transmission processing path in terms of quantization noise and sampling frequency can be reduced.

[0066] The DPD block 11 is adapted to carry out a digital predistortion on a digital signal which is supplied to it at input, intended to compensate for the non-linearities of the PA 16 in digital, according to predistortion coefficients calculated by the estimation block 19, for example by increasing the power of the input signal for the power values ​​for which the PA 16 provides attenuation. Thus, the non-linear effects of the PA 16 will be reduced (less in-band and out-of-band noise) and it will be possible to use the PA 16 with higher output power and therefore improve energy efficiency.

[0067] The techniques implemented are diverse and known to those skilled in the art: for example, predistortion type by LUT (Look-Up Table) or by mathematical models.

[0068] The LUT is a memory table which provides, for an input signal value input_bb, a corresponding output value as a function of predistortion coefficients, which it contains, which are adapted as a function of the distortions generated by the amplifier 16.

[0069] Predistortion by mathematical models is based on the mathematical modeling of non-linear systems to determine the inverse characteristic of the amplifier. For example, models derived from Volterra series are known, such as the polynomial model, the models of the Hammerstein-Wiener family or the NARMA model, etc. Coefficients of a model are thus calculated as a function of the input and output signals of the amplifier, according to an error minimization criterion.

[0070] The sigma-delta coder 12 is here a digital-to-digital sigma-delta coder which receives a bit stream, with a precision of M bits and delivers a bit stream with a precision of T bits, with T< M and M, T integers.

[0071] A sigma-delta 12 encoder is, as is known, a feedback loop device performing encoding on a small number of bits (for example, one or two or less than 5...), and a high sampling frequency compared to the bandwidth of the signal to be encoded. This device shapes the spectrum of the quantization noise by pushing its power into a frequency band not occupied by the spectrum of the useful signal. This spectral separation makes it possible, thanks to filtering of the encoded signal, to maintain a signal-to-noise ratio conforming to given specifications.

[0072] The sigma-delta 12 encoder generally comprises:

[0073] an input for receiving a digital input signal, coded on M bits;

[0074] an output for delivering a digital output value coded on T bits, here delivered to DAC 13; and,

[0075] one or more sigma-delta cells arranged between the input and the output of the encoder.

[0076] Each cell may be of first order or higher order. In particular, some may be of first order, and others of higher order.

[0077] Each sigma-delta cell comprises at least the following elements:

[0078] a subtractor comprising a first input, a second input, and an output, the first input being coupled to the encoder input to receive the input signal or a value derived from it;

[0079] an integrator comprising an input coupled to the output of the subtractor, and an output; and,

[0080] a quantizer having an input coupled to the output of the integrator and an output coupled to the second input of the subtractor; the output of the quantizer is further coupled to the output of the encoder to provide the digital output value or a contribution thereto.

[0081] The encoder 12 is characterized by its transfer function, connecting its output signal S_OUT to its input signal S_IN and to the quantization noise Q and therefore the expression of the transform in z is as follows:

[0082] S_OUT(z) = S_IN.STF(z) + Q.NTF(z)

[0083] where STF is the signal transfer function of the input signal of the encoder 12 (in English Signal Transfer Function) and NTF is the noise transfer function of the encoder 12 (in English Noise Transfer Function), expressed here as a function of the frequency; Q is the quantization noise, equal to the difference between the quantizer output and the quantizer input.

[0084] The signal transfer function operates with a gain of 1 in the bandwidth of interest. The noise transfer function is a high-pass filter function, providing noise shaping: strong suppression of quantization noise in low frequencies while quantization noise at high frequencies outside the bandwidth is increased.

[0085] The steps of a processing method 100 implemented by a transmitter device 10 in one embodiment are now described with reference to [Fig. 10]. They are carried out for example in a calibration step, which can for example take place prior to the operational phase and which in one embodiment, uses predetermined input signals.

[0086] Thus, in a step 101, the following sub-steps are implemented: - 101_l: a digital input signal input_bb is provided at time r in input of the transmitter device 10; the DPD 11 performs a predistortion of this signal, according to predistortion coefficients recently provided by the estimation block 19 (or else initialized to a predetermined value, for example to zero); - 101_2: the input signal thus predistorted is supplied to the encoder input sigma-delta 12, which reduces the number of coding bits to T bits; the signal resulting from processing by the sigma-delta 12 encoder at the output of the latter is provided at the input of the DAC 13; - 101_3: the DAC 13 performs an analog conversion of this signal which is supplied as input and delivers this analog signal to the filter input analog 14; - 101_4: filter 14 performs low-pass filtering of the signal supplied at the input and delivers it to the frequency changing stage 15_1; - 101_5: the frequency changing stage 15_1 transposes the signal which has was input at the RF carrier frequency and delivers the transposed signal to PA 16; - 101_6: the PA 16 amplifies this signal which is supplied to it, with non-linearities, and supplies it to the RF transmitting antenna.

[0087] In a step 102, which for example is carried out in parallel with the sub-steps 101_3 to 101_6, the quantization noise (Q(z)) of the sigma-delta coder 12 is extracted (for example using a subtractor, not shown, calculating the difference between the input of the sigma-delta coder 12 and the output of the sigma-delta coder 12 connected to the DAC 13 considered in step 101_2). Optionally, this quantization noise is then multiplied, in the noise shaping block 20, by the noise shaping transfer function NTZ(z) of the sigma-delta coder 12, to obtain Q(z)*NTF(z) which is the shaped quantization noise (alternatively, in certain types of coder 12, Q(z)*NTF(z) is extracted directly). The block 20, optionally, delivers a filtered portion of the quantization noise, i.e. Q(z)*H(z) or Q(z)*H(z)*NTF(z), where H(z) is a low-pass or band-pass filter, which makes it possible to limit the amount of data passing between blocks 12 and 19.The quantization noise thus optionally filtered and / or shaped is provided to the estimation block 19. .

[0088] In a step 103, the following sub-steps are implemented: - 103_l: the amplified signal at the end of sub-step 101_6 above is received at the input of the return path (for example via a coupler coupling the output of PA 16 to the antenna and the input of the return path); the signal as received is called Feedback_bb; - 103_2: the frequency changing stage 15_2 transposes the signal Feedback_bb of the baseband RF carrier frequency and delivers the transposed signal to filter 17; - 103_3: filter 17 filters the transposed signal (it removes high frequencies) and delivers the filtered signal to the ADC 18; - 103_4: ADC 18 converts the signal supplied to it as input into digital format and delivers the resulting analog signal Feedback_bb_d to estimation block 19.

[0089] In a step 104, the estimation block 19 estimates the predistortion coefficients as a function of the signal input_bb considered in step 101, of the signal obtained in step 103_4 delivered by the ADC 18 and of the signal Q(z)*NTF(z) (or Q(z)*H(z)*NTF(z) or Q(z)*H(z) or Q(z)) supplied to the estimation block in step 102. The predistortion coefficients estimated distortions are provided to DPD block 11.

[0090] In one embodiment, the steps of the method 100 are iterated cyclically after an initialization phase, the predistortion coefficients used by the DPD block 11 thus being cyclically updated; in another embodiment, they are implemented only during a preliminary calibration phase.

[0091] [Fig.7] represents a diagram of the estimation block 19 in one embodiment.

[0092] Optionally, a filter 21 filters the Feedback_bb_d signal and then a decimator 22 downsamples the filtered Feedback_bb_d signal by a factor K: this makes it possible to return to a sampling frequency similar to the baseband signal. Optionally also, a suppression block 23 eliminates the DC component (DC offset of the signal which can be significant).

[0093] Then a normalization block 24 normalizes the power of the signal supplied to it by the suppression block 23: in fact, it is necessary that when the reference signal (baseband input signal input_bb) and the transmitted signal (feedback signal Feddback_bb_d) are then compared, they have the same power. A delay estimation and compensation block 25 compensates for the delay between the input signal and the feedback signal: in fact, the feedback signal having passed through the transmission path and the return path is delayed relative to the input signal input_bb considered at time r and from which it results: it must therefore be resynchronized on the input signal to be able to make a more relevant comparison. At the end of these treatments, the delay estimation and compensation block 25 delivers the signals “normalized input_bb” and “normalized feedback_bb”.

[0094] An additional delay determination block 26 aims to determine the time shift to be applied between these signals and the signal representative of the quantization noise delivered by the noise shaping block 20 to resynchronize them with each other (i.e. they all result from the input signal input_bb considered at time r). This (optional) consideration makes the estimation of the predistortion coefficients then carried out by block 19 more relevant.

[0095] Operations are for example applied to this quantization noise Q(z)*NTF(z) (or Q(z)*H(z)*NTF(z) or Q(z)*H(z) or Q(z)) before it is subtracted from the signal coming from the return signal: it is filtered by a low-pass filter 27, then sub-sampled by a factor L by a decimator 28: this is intended to adapt the sampling frequency and the bandwidth of the quantization noise as required, in particular to the baseband signal. It is also possible to filter the quantization noise to reproduce the influence of the analog filter 14 which is in the transmission processing channel as well as the analog filter 17 which is in the return channel. It is also possible to use a band-pass filter 27 rather than a low-pass filter.

[0096] A subtracter 40 subtracts from the normalized (and synchronized) Feedback_bb return signal the signal representing the quantization noise after these filterings and decimations if necessary. And a coefficient estimation sub-block 29 then calculates the predistortion coefficients, in a known manner, to compensate for the gain variations of the amplifier according to the power of the input signal, as a function of the normalized and synchronized input_bb signals and the signal resulting from the subtraction, for example by comparing them.

[0097] In one embodiment, section Sect of estimation block 19 in [Fig.7] is replaced, in another embodiment, by the content of [Fig.8]. Similar elements between the two figures have the same reference numbers. This embodiment makes it possible to take into account more precisely the influence (and its variations over time) of the transfer functions implemented in analog, for example of the low-pass filter 14 following the DAC 13 and, in the return path, (to a lesser extent) of the filter 17 preceding the ADC 18, which can have an influence on the spectrum of the quantization noise as it is found in the return signal.

[0098] In one embodiment, the modeling of these transfer functions is carried out by a digital filter, with finite impulse response 32 (FIR for “Impulse Response Filter” in English) and the transfer function of this filter is estimated by an FIR estimation block 31 which evaluates the coefficients of this real transfer function by comparing the signal received by the return channel and the quantization noise, isolated by extraction from the sigma-delta coder.

[0099] The quantization noise cancellation operation is performed in estimation block 19, in the example detailed above, because it includes signal power normalization and delay compensation; but it can instead be performed outside this block 19.

[0100] In [Fig.5] are represented:

[0101] - the spectrum (hatched) of the input signal of the transmitting device in the mode of considered embodiment of the invention occupying +B / -B,

[0102] - the Q3 spectrum of the quantization noise as it will appear in the block estimation of coefficients 19 after cancellation of quantization noise in the + / -5B band,

[0103] - analog filtering FA3 by filter 14 arranged after DAC 13,

[0104] - the non-linearity noise due to amplifier 16, in dotted line.

[0105] The sigma-delta coder has the same behavior as in the case of [Fig.3], but in the passband, the quantization noise is canceled to access the intermodulation signal present in the band from B to 5B.

[0106] Thus, in a transmitting device according to the invention, the same sampling frequency as considered in the case of [Fig.3] can be retained even if a predistortion is performed.

[0107] DECORRELATION

[0108] One of the effects of integrating the sigma delta encoder into the transmit processing path, as described above, is to reduce the number of bits at the input of the DAC 13, which is particularly critical in multi-path transmit devices.

[0109] In addition, the proposed invention makes it possible to reduce oversampling frequency constraints, which is particularly critical in millimeter systems. In addition, these systems generally have less significant adjacent channel noise constraints. This is therefore more particularly suited to the proposed invention since a part of the quantization noise is emitted in the bands from B to 5B and is emitted in the spectrum, which pollutes the adjacent channels. In the case of millimeter systems, the adjacent channel noise constraints are generally less severe.

[0110] It is often necessary, in transmitter device architectures used for example to perform digital beamforming, to have a large number of these channels in parallel in order to obtain a significant gain in the intended direction. This large number of channels results in a large silicon surface area and can result in a large number of inputs / outputs. To limit the number of inputs / outputs, it is possible to use the same digital signal as the input of all the channels (I, respectively Q) and to perform the beamforming by operating a different delay (and / or by operating a phase shift of the signals) on each of the channels, to compensate for the difference in time of flight between the signals.

[0111] In the case of multi-channel architectures, the use of a sigma delta encoder as proposed in [Fig.6] can be problematic since if the input signal is the same on each of the transmission processing channels, then the quantization noise will be the same on each of the channels and will therefore be added coherently during beam formation, which increases the in-band and out-of-band noise. For this purpose, WO2020105040Al proposes to implement the sigma-delta modulators in a differentiated manner from each other in order to reduce the correlation of the noise between the different channels.

[0112] The implementation of predistortion in each of the channels of a transmitter device comprising N channels Vi, i = 1 to N (N integer strictly greater than 1) each comprising a transmission processing channel and a return channel as considered above contributes to solving the problem of correlation of quantization noises. Indeed, it is very likely that the coefficients of the different power amplifiers are different from each other which in itself will make the input signals different from each other and will therefore decorrelate the quantization noises.

[0113] In one embodiment, such a multi-channel transmitter device further comprises a control block (not shown) implementing processing for:

[0114] (i) ensuring that noises are decorrelated before predistortion is activated; and / Or

[0115] (ii) ensure that the sets of predistortion coefficients used by the DPDs at the same time in two different channels Vi, Vj, i ^j, remain very different.

[0116] One of the solutions is to oversize the size of the data paths in the predistortion circuits (i.e. the number of coding bits of the coefficients of the nonlinear predistortion function and the associated calculations, a number B >0 of determined bits thus taking, due to the oversize, always the digital value 0 and then giving rise to measurement errors in analog). Thus, it will be possible in the first phase before initial estimation of the nonlinear parameters of the PA, in particular before calibration, to generate linearity errors below the levels which degrade the performance of the transmitting device. These values ​​will be different for each of the channels and will thus make it possible to decorrelate the quantization noises.This will also increase the probability of having different coefficients for each of the channels or allow them to be modified slightly in the event of equality without this having an impact on the linearity performance. This will avoid modifying the sigma delta encoders on each of the channels, these encoders will therefore remain very simple and identical on all channels.

[0117] Another alternative is to send deterministic signals to the input of the sigma-delta modulators at low levels and at frequencies such that these signals are filtered by the analog pass filter used in the transmitter, so that these signals do not appear at the output of the PA 16. The amplitude or frequency of these signals can be simply varied between the different channels. These signals can be simple square signals which are simple to generate. Finally, it can also be ensured that after beam-forming, these deterministic signals cancel each other out in the intended emission direction. Such deterministic signals are easy to generate, out of band (and therefore suppressed) and they are only used during the initial phase, before estimation of the signal distortion coefficients in the operational phase.

[0118] [Fig.9] represents a transmitter device 50 of the type the transmitter device 10 represented in [Fig.6], comprising N channels VI, ... VN, each channel Vi comprising a transmission processing channel and a return channel as described with reference to [Fig.6]. A delay block 41 is further inserted between the sigma-delta encoder 12 and the DAC 13. A part of this delay can also be implemented at the level of the DAC 13. This delay makes it possible to carry out the beamforming operation more efficiently than with a phase shift. Thanks to the digital nature of the signal, the implementation of this delay is greatly simplified.

[0119] The invention has been described above in a context of radiofrequency emission, however the invention can of course be implemented in relation to other emitting devices, for example of audiofrequency signals, ultrasounds, etc.

Claims

Claims

1. Transmitter device (10; 50) comprising N channels Vi, i = 1 to N > 1, each of said channels Vi comprising: - a transmission processing channel adapted to process an input signal and comprising a predistortion block (11), a digital-to-analog converter (13), a first analog filter (14) and a power amplifier (16), said transmitter device comprising a transducer device coupled to said processing channel and receiving the input signal once processed by said processing channel, the predistortion block (11) being adapted to apply a predistortion to said input signal, as a function of predistortion coefficients, to compensate for the non-linearities of the power amplifier;- a return channel associated with the transmission processing channel, adapted to receive the signal processed by the processing channel and comprising a second analog filter (17), an analog-to-digital converter (18), and a pre-distortion coefficient estimation block (19), said distortion coefficient estimation block (19) being adapted to estimate, as a function of at least said input signal and a signal, called the return signal, obtained following at least the successive operations, on the received processed signal, of filtering by the second filter and of conversion by the analog-to-digital converter, pre-distortion coefficients and to deliver said estimated pre-distortion coefficients to the pre-distortion block; in which the processing channel further comprises a sigma-delta encoder (12) between the pre-distortion block and the digital-to-analog converter, adapted to carry out in particular a quantization; said transmitter device (10;50) being characterized in that the predistortion coefficient estimation block (19) is adapted to estimate said predistortion coefficients as a function of at least said input signal and a signal resulting from the subtraction, from said return signal, of a signal representing the quantization noise resulting from the quantization carried out by the sigma-delta coder (12).;

2. A wireless transmitter device (10; 50) according to claim 1, wherein said signal representing quantization noise is a function of said quantization noise shaped by the noise transfer function NTF(z) of the sigma-delta encoder.

3. Wireless transmitter device (10; 50) according to claim 1 or 2, adapted to obtain said signal representing the quantization noise by performing at least one of said operations among: - filtering and decimation to adapt the sampling frequency and the bandwidth of the signal representing the quantization noise to the input signal; - filtering to reproduce at least the effect of the first and second filters; - FIR type filtering to reproduce the effect of the first and second filters, the transfer function of said FIR filtering being estimated as a function of the comparison between said return signal and a signal obtained from the quantization noise extracted from the sigma-delta coder.

4. Transmitter device (50) according to one of the preceding claims, in which N >1.

5. A transmitter device (50) according to claim 4, wherein the size of data paths in the predistortion coefficient estimation block (19) in each channel is oversized, to introduce additional non-linearities; and / or the transmitter device comprises a control block adapted to, in a prior transient phase, trigger the provision of deterministic signals to the input of the digital-to-analog converters (13), such that given any two distinct channels, the amplitude and / or frequency of these deterministic signals is different between said channels.

6. A method of predistortion in a transmitter device (10; 50) comprising N channels Vi, i = 1 to N > 1, each of said channels Vi comprising: - a transmission processing channel adapted to process an input signal and comprising a predistortion block (11), a digital-to-analog converter (13), a first analog filter (14) and a power amplifier (16), said transmitter device comprising a transducer device coupled to said processing channel and receiving the input signal once processed by said processing channel, the predistortion block (11) being adapted to apply a predistortion to said input signal, as a function of predistortion coefficients, to compensate for the non-linearities of the power amplifier;- a return channel associated with the transmission processing channel, adapted to receive the signal processed by the processing channel and comprising a second analog filter (17), an analog-digital converter;

7.

8.

9. (18), and a pre-distortion coefficient estimation block (19); in which the processing channel further comprises a sigma-delta coder (12) between the pre-distortion block and the digital-to-analog converter, adapted to carry out in particular a quantization; said method comprising a step of estimating the predistortion coefficients by said distortion coefficient estimation block (19), as a function of at least said input signal and a signal, called return signal, obtained following at least the successive operations, on the received processed signal, of filtering by the second filter and of conversion by the analog-digital converter, said estimated predistortion coefficients being delivered to the predistortion block; said method being characterized in that the predistortion coefficient estimation block (19) estimates said predistortion coefficients as a function of at least said input signal and a signal resulting from the subtraction, from said return signal, of a signal representing the quantization noise resulting from the quantization carried out by the sigma-delta coder (12). A method of predistortion in a transmitter device (10; 50) according to claim 6, wherein said signal representing the quantization noise is a function of said quantization noise shaped by the noise transfer function NTF(z) of the sigma-delta encoder. A method of predistortion in a transmitter device (10; 50) according to one of claims 6 to 7, wherein said signal representing the quantization noise is obtained by performing at least one of said operations among: - filtering and decimation to adapt the sampling frequency and bandwidth of the signal representing the quantization noise to the input signal; - filtering to reproduce at least the effect of the first and second filters; - FIR type filtering to reproduce the effect of the first and second filters, the transfer function of said FIR filtering being estimated as a function of the comparison between said return signal and a signal obtained from the quantization noise extracted from the sigma-delta coder. Method for decorrelating quantization noise in a transmitting device (50), according to which a predistortion calculation method according to claims 6 to 8 is implemented on each of the channels Vi, i = 1 to N, with N integer >1.

10. A method for decorrelating quantization noise in a transmitter device according to claim 9, comprising the following steps: the size of data paths in the predistortion coefficient estimation block (19) in each channel is oversized, to introduce additional non-linearities; and / or in a prior transient phase, deterministic signals are provided to the input of the digital-to-analog converters (13), such that given any two distinct channels, the amplitude and / or frequency of these deterministic signals is different between said channels.

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