Dual-band filter for IFF / ADS-B receiver chain

Abstract

The invention relates to a dual-band analog filter (300) for passing IFF and ADS-B signals in a first and a second frequency band, and for rejecting signals received outside these bands, implemented using multi-planar technology, and comprising successively: a layer (301) comprising an input port and an output port, a layer (302) comprising a ground plane having openings opposite the input and output ports, a layer (303) comprising N bandpass resonators (921 to 926), a layer (304) comprising N bandstop resonators (931 to 936), arranged such that: the bandpass resonators are coupled in pairs, the first and last bandpass resonators are electromagnetically coupled to the input and output ports, each bandpass resonator is electromagnetically coupled to a resonator Strip cutter. Figure for the abbreviation: Fig. 3

Description

Title of the invention: Dual-band filter for IFF / ADS-B reception chain Technical field

[0001] The invention lies in the field of air traffic control and anti-collision devices for air traffic. It relates more specifically to a dual-band analog filter usable in a reception chain configured to allow simultaneous reception and on separate frequencies of IFF (Identification Friend of Foe) interrogations and ADS-B (Automatic Dependent Surveillance-Broadcast) interrogations. The dual-band filter according to the invention must meet strong constraints of rejection of third-party signals and volume. Prior art

[0002] IFF is an encrypted aircraft identification system, allowing civil or military approach radars to recognize cooperating aircraft, known as "friendly", and to determine their heading and distance. IFF is also used in flight in military aircraft to identify friendly and enemy aircraft. IFF interrogations are made by ground stations or aircraft on a frequency band centered around the 1030 MHz frequency, typically 14 MHz.

[0003] ADS-B is a cooperative surveillance system for air traffic control. An aircraft equipped with ADS-B determines its position using a Global Navigation Satellite System (GNSS) and periodically sends it to ground stations and other aircraft flying nearby. The position transmission rate depends on the phase of flight. ADS-B transmissions are made on a frequency band centered around 1090 MHz, typically 16 MHz.

[0004] Most aircraft, particularly airplanes, carry the equipment necessary to implement these two standards, in particular the equipment necessary to listen to TIFF on the 1030 MHz frequency and ADS-B on the 1090 MHz frequency.

[0005] These reception equipments are subject to strong performance constraints by the standards DO-260 B or C (for ADS-B), AIMS 03-1000 (for TIFF), and ED73 E or F (ADS-B). The constraints relate in particular to minimum rejection levels for the reception radio chains. In particular, the rejection levels to be achieved are very high (greater than 60 dB) on the 1053 MHz - 1065 MHz band which is between the TIFF reception frequency and that of ADS-B.

[0006] In order to meet these strong rejection constraints, state-of-the-art aircraft generally carry separate equipment for TIFF and ADS-B. This equipment has head analog filters designed to achieve the rejection levels defined by the standards, designed by cascading a large number of bandpass filters of various technologies: ceramic cavity filters, SAW filters (Surface Acoustic Waves), BAW filters (Bulk Acoustic Waves), etc.

[0007] Some equipment, such as the TSC4000 from Thailand, performs the reception of IFF interrogations and ADS-B messages in the same equipment. However, these receptions are done on two separate radio channels.

[0008] Embarking two separate receiving equipment or equipment with two separate receiving chains in an aircraft to perform the IFF reception and ADS-B reception functions has obvious consequences in terms of volume and weight, which may not be compatible with small aircraft. In addition, the consumption of the equipment is not optimal, which is a major drawback for drone-type applications. Finally, the components of each receiving chain are relatively expensive, and the higher their number, the higher the series cost.

[0009] For this reason, the applicant filed patent application FR 2214616, relating to a reception chain allowing continuous reception of IFF interrogations around the frequency 1030 MHz and ADS-B transmissions around the frequency 1090 MHz, while respecting the normative constraints associated with these receptions.

[0010] Designing a radio chain allowing the simultaneous reception of IFF and ADS-B signals, with stable filtering in the frequency bands of interest and respecting the normative constraints on the rejection levels of signals outside these bands, proves to be particularly difficult. While it is possible to design analog filters making it possible to achieve the desired rejection levels around one or other of the IFF or ADS-B frequency bands, designing analog filters that are passable and stable simultaneously around the two frequencies of interest while guaranteeing the required rejection levels outside these bands (greater than 60 dB), in particular in the frequency band located between the IFF band and the ADS-B band, requires the use of particularly high-order filters.Such filters are particularly complex and expensive to produce, have band ripple problems, and have significant weight and volume.

[0011] The realization of a high-order digital filter which is passable and stable on the two frequency bands of interest and has the desired rejection properties is theoretically possible. However, such a filter would be positioned in the reception chain after the analog-digital converter, which consequently would not be not protected against high-level emissions occurring in frequency bands close to those of TIFF and ADS-B, in particular those occurring in the frequency band located between the two bands of interest. These unrejected emissions can have the effect of making the ADC work at saturation. In practice, saturation of the ADC leads to non-linear distortion which at the very least creates broadband spectral noise, which destroys the dynamics of the receiver, and at worst will cause it to deteriorate.

[0012] For these reasons, the invention which is the subject of application FR 2214616 defines a hybrid analog / digital radio reception chain allowing simultaneous reception of IFF signals and ADS-B signals. [Fig.l] schematically represents the main elements of this reception chain. It comprises an analog part 101 with: - an antenna 103 configured to receive on a frequency band comprising the IFF frequency band (band centered around the frequency 1030 MHz) and the ADS-B frequency band (band centered around the frequency 1090 MHz), - a dual-band filtering device 104 configured to filter the radiofrequency signal acquired on the receiving antenna 101, advantageously a dual-band filter passing around the frequency bands of the IFF (centered around 1030 MHz) and ADS-B (centered around 1090 MHz) signals while rejecting the signals outside these bands, in particular those located in the intermediate frequency band, - a low noise amplifier 105 configured to amplify the signal received on the antenna, - a mixer 106, configured to mix the amplified signals with a sinusoidal signal so as to transpose the IFF and ADS-B signals to a lower frequency compatible with the operating band of the analog-to-digital converter (ADC) 107, - an analog-to-digital converter 107.

[0013] It also comprises a digital part 102 configured to duplicate (108) the signals digitized by the CAN, and to process the duplicated signals respectively by a processing path 110 for the reception of the IFF signals and by a processing path 120 for the reception of the ADS-B signals.

[0014] The IFF path 110 comprises a digital filter 111 and an I / Q demodulator 112, configured to reject the signals around the IFF signal and transpose the IFF signal to the baseband. The positions of the digital filter 111 and the demodulator 112 can be inverted. It then comprises means 113 for processing the demodulated samples, configured to decode the IFF signals.

[0015] The ADS-B path 120 comprises a digital filter 121, and an I / Q demodulator 122, configured to reject the signals around the ADS-B signal and transpose the ADS-B signal to the baseband. The positions of the digital filter 121 and the demodulator 122 can be inverted. It then comprises means 123 for processing the demodulated samples, configured to decode the ADS-B signals.

[0016] The dual-band filtering device 104, the digital filter 111 of the IFF path and the digital filter 121 of the ADS-B path together contribute to the filtering of the received signals, and in particular to compliance with the rejection constraints defined by the DO-260, AIMS 03-1000 and ED73 standards. In particular: - the first dual-band analog filtering level 104 is intended to provide part of the desired rejection of signals located outside the bands of interest, in particular those received between the IFF signal and the ADS-B signal, in order to avoid saturation of the CAN linked to non-essential radiation, - the second level of digital filtering 111 and 121, adapted to one or other of the signals of interest, is configured so that the two levels of filtering together make it possible to achieve the normative requirements.

[0017] Thus, the dual-band filtering device 104 contributes to the rejection of signals outside the IFF and ADS-B bands, but is not bound by the constraints defined by the standards, which is why its template can be more relaxed, thus making the production of such an analog filter possible in reduced dimensions. The filter must however sufficiently reject the adjacent bands, in particular the frequency band located between the IFF band and the ADS-B band, to avoid the phenomenon of saturation of the ADC due to non-essential radiation. An aim of the invention is to define such a filter. Summary of the invention

[0018] To this end, the present invention describes a dual-band analog filter configured to be passable for IFF signals received in a first frequency band centered on the frequency 1030 MHz and ADS-B signals received in a second frequency band centered on the frequency 1090 MHz, and to reject signals received outside the first frequency band and the second frequency band, in particular at least one frequency band between the first and second frequency bands.

[0019] The dual-band analog filter according to the invention is produced using multiplanar technology, and comprises a plurality of superimposed layers, among which successively: - a first layer comprising an input port and an output port, - a second layer comprising a ground plane having openings opposite the input port and the output port, - a third layer comprising N bandpass resonators, with N greater than or equal to 1, - a fourth layer comprising N band-stop resonators.

[0020] The elements of the layers are arranged so that: - the N bandpass resonators are connected two by two by electromagnetic coupling, - the first resonator of the N bandpass resonators is connected by electromagnetic coupling with the input port, - the last resonator of the N bandpass resonators is connected by electromagnetic coupling with the output port, - each bandpass resonator is connected by electromagnetic coupling to one of the bandstop resonators.

[0021] Advantageously, the dimensions of the band-pass resonators and the band-stop resonators are chosen so that each pair of band-pass / band-stop resonators forms a dual-band resonator configured to be passable in said first and second frequency bands, and to reject signals received outside these two frequency bands, in particular in the frequency band between the first and second frequency bands.

[0022] According to an embodiment of a dual-band analog filter according to the invention, where N is greater than or equal to 3, two non-consecutive bandpass resonators among the N bandpass resonators are connected by electromagnetic coupling.

[0023] Advantageously, the spacing between the two non-consecutive bandpass resonators connected by electromagnetic coupling is chosen so as to form zeros in the impulse response of the filter positioned so as to increase the slope on the edges of the filter. In one embodiment, the two non-consecutive bandpass resonators connected by electromagnetic coupling are the first and the last bandpass resonator.

[0024] According to one embodiment, the dual-band analog filter according to the invention is configured to reject by at least 30 dB the signals received in the frequency band between the first and second frequency bands. Brief description of the drawings

[0025] The invention will be better understood and other characteristics, details and advantages will appear more clearly on reading the following description, given without limitation, and thanks to the appended figures, given by way of example.

[0026] [Fig. 1] [Fig. 1] schematically represents a hybrid radio reception chain allowing the simultaneous reception of IFF and ADS-B signals as defined in the patent application FR 2214616.

[0027] [Fig.2] [Fig.2] gives an example of a dual-band analog filter template that can be used for implementing a dual-band IFF / ADS-B receiver.

[0028] [Fig.3] [Fig.3] represents the structure of a multi-layer dual-band analog filter according to one embodiment of the invention.

[0029] [Fig.4] [Fig.4] represents the principle of coupling between elements located in different layers of a multilayer filter.

[0030] [Fig.5] [Fig.5] shows the theoretical form of a ladder low-pass filter.

[0031] [Fig.6a] [Fig.6a] illustrates the equivalent structure of a dual-band resonator.

[0032] [Fig.6b] [Fig.6b] represents an assembly equivalent to that of [Fig.6a].

[0033] [Fig.7] [Fig.7] represents the structure of a dual-band filter of order N.

[0034] [Fig.8] [Fig.8] is a coupling diagram of the 6th order multilayer dual-band filter according to the embodiment of the invention.

[0035] [Fig.9] Figure 9 represents a top view of a multi-layer dual-band filter of order N - 6 according to an embodiment of the invention, in which the different layers are represented in superimposition.

[0036] [Fig. 10] [Fig. 10] represents the theoretical frequency response of the multi-layer dual-band filter according to the embodiment of the invention represented in [Fig.9].

[0037] Identical references may be used in different figures when they designate identical or comparable elements. Description of the embodiments

[0038] [Fig.2] gives an example of a dual-band analog filter template 103 that can be used for implementing a dual-band IFF / ADS-B receiver where the filtering is carried out jointly in an analog and digital manner.

[0039] This template is defined as passing between the frequencies fQi = 1005 MHz and fQ2 = 1055 MHz, which includes the frequency band associated with the reception of IFF signals. The oscillation of the signals in the sub-band fQOi = 1016 MHz to fQo2 = 1044 MHz is limited to 6 dB, so as to minimize distortions of the IFF signal.

[0040] This filter is also defined as passing between the frequencies fAi = 1065 MHz and f A2 = 1115 MHz, which includes the frequency band associated with the reception of ADS-B signals. The oscillation of the signals in the sub-band fAOi = 1076 MHz to fA02 = 1104 MHz is limited to 6 dB, so as to minimize distortions of the ADS-B signal.

[0041] Outside these two frequency bands, the rejection level of the filter is greater than 30 dB, in particular in the frequency band 1055 MHz - 1063 MHz, in order to ensure sufficient rejection of the signals transmitted in this band to avoid the phenomenon of saturation of the CAN.

[0042] The template in [Fig.2] is given for illustration purposes only. It can be modified depending on the desired performance, the efficiency of the digital filters implemented after the analog filter, the performance of the analog-to-digital converter, etc. However, the rejection values ​​taken in the example in [Fig.2] constitute a good compromise between rejection of unwanted signals, in particular in the band between the IFF signal and the ADS-B signal, and compactness of implementation. The stability of the signal sought in the passbands of the filter in [Fig.2] also results from a good compromise between distortions made to the spectrum of the useful signals and complexity of implementation.

[0043] Several filter technologies can be used to perform analog filtering functions. For example, we know volume technologies (cavity filters), planar technologies (microstrip filters, coplanar filters, multilayer filters, membrane filters) and SAW filters (acronym for Surface Acoustic Wave) or BAW filters (acronym for Bulk Acoustic Wave).

[0044] The invention relates to a filter in planar technology, and more precisely to a multilayer filter. Such filters consist of a plurality of superimposed dielectric layers. They have the advantage of having an extremely small footprint, very good electrical performance, and of being simple and inexpensive to produce. The thesis by Y.Wu, E.Fourn, P.Besnier and C.Quendo, “Direct synthesis of multiband bandpass filters with generalized frequency transformation methods”, IEEE Transactions on microwave theory and techniques, vol. 69, no. 8, pp. 3820-3831, Aug. 2021 describes the principle of multi-band planar filters.

[0045] [Fig. 3] represents the structure of a multi-layer dual-band analog filter according to an embodiment of the invention. This multi-layer filter 300 has several superimposed layers 301 to 304, among which successively: • a first layer 301, comprising an input port and an output port allowing the filter to interface with the other elements of the reception chain, • a second layer 302, located directly under the first layer, comprising a ground plane having openings facing the input port and the output port, • a third layer 303 comprising N band-pass resonators (in English band-pass resonators with N greater than 1. The band-pass resonators are connected two by two by electromagnetic coupling, and • a fourth layer 304 comprising N band-stop resonators. Each of the N band-stop resonators is connected by electromagnetic coupling to a single resonator among the N resonators bandpass.

[0046] The purpose of the second layer 302 is to isolate the first layer 301, comprising the filter ports, from the layers 303 and 304, comprising the N bandpass resonators, so as to avoid the appearance of interference generated by an unwanted coupling effect, for example between the input / output ports and the filter components, or between the filter components and external components, disturbances which would disturb the frequency response of the filter. The openings made in the ground plane of the layer 302 have the effect of allowing the connection between the resonators of the layer 303 and the input / output ports by electromagnetic coupling.

[0047] In this embodiment of the invention, the first resonator of the N bandpass resonators of the layer 303 is connected by electromagnetic coupling with the input port through an opening made in the ground plane of the layer 302. Similarly, the last resonator of the N bandpass resonators of the layer 303 is connected by electromagnetic coupling with the output port, through an opening made in the ground plane of the layer 302.

[0048] Advantageously, the filter may comprise a ground plane 305 located above the layer 301 comprising the access ports, and / or a ground plane 306 located under the layer 304 comprising the band-stop resonators. These ground planes, which cover the entire surface of the filter, make it possible to protect it against interference linked to elements external to the filter. Advantageously, these ground planes may be connected to the ground plane 302 by a metallized hole passing through the layers (via) 307 so that the ground planes are exactly at the same potential.

[0049] The various components of the multi-layer dual-band filter according to the invention are connected to each other by electromagnetic coupling. The term "connected by electromagnetic coupling" means that two elements not connected by conductive tracks are arranged close enough to induce an exchange of energy between them. The value of the coupling coefficient depends directly on the distance separating the elements and the length of the lines located opposite each other. Depending on their respective arrangement, they can be electrically or magnetically coupled.

[0050] Connecting the resonators and the input / output ports by electromagnetic coupling makes it possible, among other things, to simplify the production of the multi-layer dual-band filter according to the invention, since it is not necessary to introduce metal tracks between the elements of the same layer, or metallized holes between the elements of different layers. Its complexity and manufacturing cost are therefore reduced.

[0051] [Fig. 4] represents, by way of illustration, the principle of coupling between elements located in different layers of a multilayer filter, in the case of two elements separated by a ground plane having openings. Reference 401 designates a port of layer 301, for example the input port of the filter. Reference 402 designates for example the first resonator of the N bandpass resonators of layer 303. Reference 403 designates the ground plane of layer 302. Port 401 and resonator 402 are located plumb above each other, and ground plane 403 is interrupted between these two elements, so as to allow coupling effects. Coupling is also possible between components located in adjacent layers of a multilayer filter, in the absence of an intervening ground plane.

[0052] It is well known that an N-order low-pass filter can be represented in the form of series inductances and parallel capacitances, as illustrated in [Fig.5].

[0053] It is also known that this structure can be modified, to be represented in the form of parallel capacitors only, separated by admittance inverters, which make it possible to transform series coils into parallel capacitors.

[0054] From the filter template given in [Fig.2], and as described in the thesis of Y.Wu cited previously, it is possible to determine the transfer function of a dual-band filter in the form: O - h ( - -2 'l - 1 «J ^49

[0055] with: - Q the transfer function of the filter, - and b0 the resonance frequency and the susceptance, or slope parameter, of a bandpass resonator equivalent to the first part of the equation, - and b} the resonance frequency and the susceptance, or slope parameter, of a band-stop resonator equivalent to the second part of the equation.

[0056] This dual-band resonator is composed of a band-pass resonator, equivalent to an inductance and a capacitance connected in parallel, positioned in parallel with a band-stop resonator, equivalent to an inductance and a capacitance connected in series. [Fig.6a] illustrates the equivalent structure of a dual-band resonator, composed of a band-pass resonator formed by the capacitance Co and the inductance Lo, connected in parallel with a band-stop resonator formed by the capacitance Ci and the inductance Lp

[0057] The values ​​of the capacitances and inductances Co, Ci, Lo and Li can be obtained by the following formulas:

[0058] Lq = —h- [°059] Q = 7-777

[0060] L1=—L-

[0061]

[0062] where CLP is the value of the original low-pass prototype capacitance obtained using the Chebyshev or Butterworth shape coefficients associated with the response expected.

[0063] Furthermore, the dual-band resonator of [Fig.6a] can be transformed into that of [Fig.6b], to comprise only LC components in parallel, by introducing an admittance inverter of impedance J, with: with A “ MiCi-

[0064] By taking the ladder low-pass filter structure of [Fig.5], modified to present only parallel capacitors separated by admittance inverters, and by using the dual-band resonator of [Fig.6b] in place of the parallel capacitors, we obtain the structure of an N-order dual-band filter as shown in [Fig.7], with G s the source impedance and GL the load impedance.

[0065] The impedances of the admittance inverters are respectively:

[0066] PW [°067] _ ± Jü+1 b^g^

[0068] _ /

[0069] where the coefficients 8, are the Chebyshev or Butterworth shape coefficients associated with the expected response.

[0070] These steps make it possible to express a dual-band filter of order N according to the invention by a succession of parallel dual-band filters separated by admittance inverters, each dual-band filter itself being able to be seen as a band-pass filter positioned in parallel with an admittance inverter and a band-stop filter, the band-pass filter and the band-stop resonator being represented in the form of a parallel LC circuit.

[0071] In microstrip technology, a parallel LC filter can be implemented in the form of a resonator, i.e. an open line of length 4 / 4 and impedance Z. The parameter 4 is the wavelength in the substrate, which can be derived directly from the resonant frequency a'o of the bandpass filter, or wi of the bandstop filter. The impedance Z of a lossless transmission line is approximately ^L / C ■ This impedance makes it possible to calculate the width of the transmission line, also depending on the type of substrate. These parameters make it possible to determine the dimensions of open lines used to implement the resonators of the dual-band filter according to the invention.

[0072] One way to implement an admittance inverter is to replace it with a quarter-wave line. It is also possible to replace it with a coupling circuit. The coupling coefficients are related to the impedances of the admittance inverters.

[0073] The invention proposes to implement a dual-band filter passing the frequency bands associated with the IFF and ADS-B signals (i.e., passbands centered around the frequencies 1030 MHz and 1090 MHz) while rejecting the other frequencies, in particular frequencies intermediate to these two bands. If we refer to the template of [Fig. 2], this means that the dual-band filter is passing at least in the frequency bands from fQOi to fQ02 and from fAOi to fA02, and that it rejects by at least 30 dB the signals whose frequency is lower than fQ, higher than fA, and between fQ2 and fAb

[0074] The implementation is done by means of a multilayer circuit composed of resonators equivalent to parallel LC circuits connected by electromagnetic coupling, which makes it possible both to do away with transmission lines and thus to simplify the design of the filter, and to implement the admittance inverters necessary for the implementation of the dual-band filter according to the invention.

[0075] Figure 8 is a coupling diagram of a multilayer dual-band filter of order N = 6 according to an embodiment of the invention. The order of the filter is the result of a compromise between, on the one hand, the performance of the filter (steepness of the slopes of the spectral template, rejection level, oscillation in the band) and, on the other hand, the surface occupied by the filter. The invention applies for any value of N greater than or equal to 1.

[0076] There is an input port (In) and an output port (Out), connected respectively to an input (Source) and an output (Load) of the filter. The filter is implemented here in the form of N = 6 dual-band resonators each comprising a band-pass resonator Pi and a band-stop resonator. The two resonators are associated by coupling, with coefficient Ksj., so as to replace the admittance inverter J of Figure 6b. In practice, we have Ks,i — J. Each band-pass resonator is connected by coupling to another band-pass resonator, with a coupling coefficient Ku+c so as to replace the admittance inverters of impedance Jij+\, with i 6 (1, N), of Figure 7. In practice, we have K ij+i — J

[0077] There is also an input port (in English Source) and an output port (in English Load), respectively connected by electromagnetic coupling to the first and last of the N bandpass resonators. The connection is made by coupling, respectively of coefficients Qes and Qel, so as to implement the admittance inverters Jo.i and Jnjv+i of figure 7. The coupling between the input / output ports and the bandpass resonators is made through the openings of the ground plane 801, and is adjusted according to the desired coupling coefficients. In practice, we have _ / j \2 and O ~ (............V'

[0078] The multilayer structure of the dual-band filter according to the invention makes it possible to interface all the resonators by coupling effects while avoiding parasitic couplings, which would be difficult to achieve on a single-layer filter. It also makes it possible to freely interface the position of the input / output ports with respect to the resonators, thanks to the ground plane isolating the layer comprising the bandpass resonators from the input / output ports. Another advantage is to be able to adjust the type of substrate of each layer, so as to adapt the dimensions of the lines of the different layers so that they have widths close to each other, which makes it possible to simplify the design of the filter and simplifies the couplings between resonators of different layers.

[0079] Figure 9 shows a top view of a multilayer dual-band filter of order N - 6 according to an embodiment of the invention, given for illustration purposes only, in which the different layers are shown in overprint. There is an input port 901 and an output port 902, and a ground plane 913 covering the entire surface of the filter. The ground plane 913 has two openings 911 and 912, positioned between metal tracks 904 and 905 extending the input output ports on the layer 301, and bandpass resonators 921 and 926 positioned on the layer 303. The input ports and the bandpass resonators are thus connected two by two by coupling effects. The metal tracks 904 and 905 and the ground plane 913 allow the input / output ports 901 and 902 to be positioned freely over the entire surface of the filter.

[0080] The filter also comprises N = 6 bandpass resonators 921 to 926, connected two by two by electrical or magnetic coupling effect. In the example, resonators 921 and 922 are connected by magnetic coupling, while resonators 923 and 924 are connected by electrical coupling. The N = 6 resonators thus form a loop between the input port and the output port. The dimensions of the resonators are obtained theoretically from the template of the filter, as described previously. However, they can be adjusted to the margin to compensate for possible imperfections during the production of the circuit or possible undesirable interference. Their shapes and positions are chosen so as to limit the size of the filter while respecting the coupling constraints between elements.

[0081] The filter also comprises N = 6 band-stop resonators 931 to 936. Each band-stop resonator is positioned so as to be connected to a single band-pass resonator 921 to 926 by electrical or magnetic coupling. In the example of [Fig.9], the band-stop resonator 931 is connected by magnetic coupling to the band-pass resonator 921, etc. The dimensions of the band-stop resonators are obtained theoretically from the filter template, as described previously, and can be adjusted to the margin. Their shapes and positions are chosen so as to limit the size of the filter while respecting the coupling constraints between elements, in particular the absence of significant coupling between band-stop resonators.

[0082] In the example of [Fig.9], the resonators are open-end resonators. However, the invention applies to any type of resonator, such as open-stub resonators, ring resonators, etc.

[0083] Advantageously, in one embodiment of a dual-band IFF / ADS-B filter according to the invention, two of the non-consecutive bandpass resonators are connected by coupling effect. In the example of Figure 9, these are the bandpass resonator 921 and the bandpass resonator 926, the ends of which are brought together so that they are connected by electrical coupling with a coupling coefficient corresponding to $ in [Fig.8]. Indeed, the coupling of non-consecutive bandpass resonators of the series of bandpass resonators makes it possible to create zeros in the frequency response of the dual-band filter.

[0084] The zeros of a filter constitute the frequencies for which the theoretical attenuation of the filter is infinite. The multi-layer dual-band filter according to the invention uses the coupling between two non-consecutive bandpass resonators of layer 303 to increase the slope of the frequency response at the edges of the filter. The other non-consecutive resonators of the filter are arranged sufficiently far apart so as not to generate significant coupling effects.

[0085] In the example of [Fig.9], resonator 921 and resonator 926, which constitute the first and last of the series of bandpass resonators, are connected by electromagnetic coupling. However, other pairs of non-consecutive bandpass resonators make it possible to produce equivalent results.

[0086] [Fig. 10] represents the theoretical frequency response of the multi-layer dual-band filter according to the embodiment of the invention shown in [Fig.9]. Curve 1001 represents the reflection coefficient of the filter, or SI 1, as a function of frequency. Curve 1002 represents the attenuation of the filter, or S21, as a function of frequency.

[0087] It can be seen that this filter conforms to the spectral template shown in [Fig.2], that is to say that it has a rejection greater than 30 dB for frequencies below 1005 MHz (in practice, those below 1100 MHz), for frequencies above 1115 MHz, and in the intermediate frequency band 1055 MHz - 1065 MHz.

[0088] In addition to the zeros linked to the response of the dual-band filter, such as the zero 1010 close to the frequency 1060 Hz, the filter has two zeros 711 and 712 located on the edges of the filter passband, linked to the coupling between two non-consecutive band-pass resonators PI and P2, and which have the effect of making the slope of the filter steeper, thus allowing high rejection levels to be quickly achieved outside the two IFF and ADS-B operating bands. These zeros are symmetrical with respect to the central band of the filter (i.e. 1060 MHz). In practice, the distance between the non-consecutive bandpass resonators coupled to generate these zeros is adjusted empirically, in simulations, so as to obtain the optimal position of the zeros.

[0089] The filter according to the invention can therefore be positioned at the reception head of a device configured to simultaneously process IFF signals and ADS-B signals. It makes it possible to significantly reject signals transmitted in adjacent frequencies, in particular those transmitted in the frequency band located between the IFF signals (1030 MHz) and the ADS-B signals (1090 MHz). It can therefore be used to implement a common IFF / ADS-B reception chain satisfying strict rejection requirements, in particular the DO-260, AIMS 03-1000 and ED73 standards.

[0090] The filter according to the invention has very good rejection of out-of-band signals, a very good response in the bands of interest and good selectivity, with a reduced volume, thanks to the combination of the following techniques: - a multi-layer structure, making it possible to design filters of reduced size and to easily arrange elements connected by coupling effects, - the connection of the components (ports and resonators) by coupling effects, making it possible to limit the components of the filter to the strict minimum and to eliminate all the metallic elements (tracks, vias) necessary to connect the components together, and therefore to reduce the volume and consumption of the filter, - the addition of specific couplings between resonators, allowing zeros to be positioned in the frequency response of the filter in order to increase its frequency selectivity.

Claims

Claims

1. Dual-band analog filter (300) configured to be passable for IFF signals received in a first frequency band (fQOi - fQ02) centered on the frequency 1030 MHz and ADS-B signals received in a second frequency band (fAoi - fA02) centered on the frequency 1090 MHz, and to reject signals received outside the first frequency band and the second frequency band, in particular at least one frequency band (fQ2 - fAi) between the first and second frequency bands, said dual-band analog filter being produced in multi-planar technology, and being characterized in that it comprises a plurality of superimposed layers among which successively: - a first layer (301) comprising an input port (901) and an output port (902), - a second layer (302) comprising a ground plane having openings (911, 912) facing the port input and output port,- a third layer (303) comprising N bandpass resonators (921 to 926), with N greater than or equal to 1, - a fourth layer (304) comprising N bandstop resonators (931 to 936), and in that the elements of said layers are arranged so that: - the N bandpass resonators are connected two-by-two by electromagnetic coupling (Ki>2, K2>3, K3>4, K4 5, K5>6), - the first resonator of the N bandpass resonators is connected by electromagnetic coupling (Qe,s) with the input port, - the last resonator of the N bandpass resonators is connected by electromagnetic coupling (Qe,i) with the output port, - each bandpass resonator is connected by electromagnetic coupling (Ks.i, KSj2, KSj3, KSj4, KSj5, KSîô) to a of said band-stop resonators.,

2. Dual-band analog filter according to claim 1, wherein the dimensions of the band-pass resonators and the band-stop resonators are chosen so that each pair of band-pass resonators band / bandstop forms a dual-band resonator configured to be conductive in said first and second frequency bands, and to reject signals received outside these two frequency bands, in particular in the frequency band between the first and second frequency bands.

3. A dual-band analog filter according to one of the preceding claims, wherein N is greater than or equal to 3, and wherein two non-consecutive bandpass resonators (921 / 926, P1 / P6) among the N bandpass resonators are connected by electromagnetic coupling (Ci,6).

4. A dual-band analog filter according to claim 3, wherein the spacing between the two non-consecutive bandpass resonators connected by electromagnetic coupling is chosen so as to form zeros (1011, 1012) in the impulse response of the filter positioned so as to increase the slope at the edges of the dual-band filter.

5. Dual-band analog filter according to one of claims 3 and 4, in which the two non-consecutive bandpass resonators connected by electromagnetic coupling are the first (921, PI) and the last (926, P6) bandpass resonator.

6. Dual-band analog filter according to one of the preceding claims, configured to reject by at least 30 dB the signals received in said frequency band (fQ2 - fAi) between the first and second frequency bands.

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