Magnetic resonance type wireless power supply device

Abstract

To provide a magnetic resonance type wireless power supply device capable of performing a selection fixing of at least one of two resonance loops with a method for not applying a limitation to a layout of a power supply coil and a power reception coil in regardless of an environment of a circumference.SOLUTION: A magnetic resonance type wireless power supply device comprises: a power supply side resonance circuit that contains a power supply coil; and a power reception resonance circuit that contains a power reception coil, and performs both of the power supply coil and the power reception coil to be in a magnetic resonance by using a PT (Parity-Time) symmetry. In a power supply side resonance circuit 1, an inverter 3 and a current sensor 4 are connected, and a timing that is a time when switching elements 33 and 34 in an internal part of the inverter 3 are turned on / turned off is determined on the basis of a pulse generated by a signal after a phase relation is adjusted by a phase adjustment circuit 30, and thus any one of two resonance loops is selected and fixed.SELECTED DRAWING: Figure 9

Description

[Technical field]

[0001] The present invention relates to a magnetic resonance type wireless power feeder that utilizes parity-time symmetry (hereinafter referred to as "PT symmetry") to feed power in a non-contact manner by magnetically resonating a feeding coil and a receiving coil. [Background technology]

[0002] Conventionally, there are several known wireless power transfer technologies, such as those based on electromagnetic induction and magnetic resonance. Of these, the wireless power transfer technology based on electromagnetic induction is used, for example, for charging mobile phones, and has coils arranged above and below each other. In other words, based on the same principle as a transformer, power can be transferred only when the distance (transmission distance) between the power transfer coil and the power receiving coil is very short.

[0003] However, because the transmission distance of electromagnetic induction wireless power transfer technology is short, at around a few millimeters, the distance between the power transfer coil and the power receiving coil cannot be large. In addition, even the slightest misalignment or separation between the power transfer coil and the power receiving coil makes charging and power transfer impossible. In other words, because this technology is sensitive to misalignment, it is difficult to apply it to artificial devices attached inside the human body, such as artificial hearts, or devices that rotate in multiple directions or have an axial misalignment, such as robotic arms.

[0004] In addition, magnetic resonance wireless power transfer technology has a long transmission distance of several centimeters to several meters, so the distance between the power transfer coil and the power receiving coil can be made larger than with the electromagnetic induction method, and it is approaching a level where it can be put to practical use. However, power cannot be transferred unless the distance between the power transfer coil and the power receiving coil is kept constant, and if they get closer or farther than that distance, or if there is an angle, the transmission efficiency decreases and the required power cannot be sent. In other words, this method is also vulnerable to positional deviations, so there is a problem that it is difficult to apply it to devices that rotate in multiple directions or have an axis deviation (rotating power supply target) such as a robot arm.

[0005] One of the wireless power supply methods is a technology called wireless power supply that uses Parity-Time symmetry (hereinafter referred to as "PT symmetry"). This wireless power supply that uses PT symmetry is a physical system with a non-Hermitian Hamiltonian, and is a new concept of wireless power supply that was first announced in 2017 (see Non-Patent Document 1).

[0006] When PT symmetry is preserved, the eigenenergy of the Hamiltonian becomes real, so that the system behaves as if its energy is preserved, even though it is a non-Hermitian system. In this case, the energy transferred between the power supply resonant circuit and the power receiving resonant circuit per unit time does not depend on the coupling coefficient between the two resonant circuits. As a result, even if the transmission distance changes or the position of the power supply coil and the power receiving coil is misaligned, the transmitted power and power transfer efficiency will always be kept constant as long as PT symmetry is preserved.

[0007] Furthermore, many methods have been proposed for automatically adjusting the frequency of inverters used as AC power sources in order to suppress fluctuations in transmission power and power transmission efficiency with changes in transmission distance (see Non-Patent Document 2). However, these control methods cannot completely separate the transmission power and power transmission efficiency from the transmission distance. This is because all of these systems are not designed as non-Hermitian physical systems and do not preserve PT symmetry.

[0008] Wireless power supply using PT symmetry is a technology in which the AC power supply in conventional magnetic resonance wireless power supply technology is replaced with an inverter that behaves electrically in the same way as a negative resistor, that is, an inverter that behaves as a negative resistor.

[0009] This is a known technology as disclosed in, for example, Patent Document 1, but to explain it in more detail, an inverter that behaves as a negative resistance is an inverter whose switching frequency and voltage amplitude are not fixed in advance, has a circuit configuration in which the switching frequency is determined by the apparent resonance frequency of the wireless power supply circuit as seen from the output end of the inverter, and has a switching frequency that responds quickly to changes in the apparent resonance frequency of the wireless power supply circuit that may change due to changes in the transmission distance of the coil or positional deviation. Here, the wireless power supply circuit is a circuit that includes a power supply side resonant circuit and a power receiving side resonant circuit, and all subsequent circuits connected thereto. In addition, the apparent resonance frequency means a substantial resonance frequency that takes into account the interaction between the power supply side resonant circuit and the power receiving side resonant circuit.

[0010] In addition, in wireless power supply using PT symmetry, the inverter described above is used in a self-oscillating manner. There are two modes of oscillation, and PT symmetry can be preserved in either mode. Here, these two modes will be explained in more detail. Figure 1 shows an equivalent circuit of the SP topology, and is the same as Figure 14 in Patent Document 1.

[0011] As shown in Fig. 1, the power supply side resonant circuit 1 and the power receiving side resonant circuit 2 of the SP topology have a mutual inductance k m The L in FIG. 1 represents the self-inductance of the power supply coil 11 and the power receiving coil 21. L(1-k m ) represent the leakage inductance of the power supply coil 11 and the power receiving coil 21, respectively. Also, r1' and r2' represent the winding resistance of the power supply coil 11 and the power receiving coil 21, respectively. r crepresents the iron loss equivalent resistance. C represents the capacitance (electrostatic capacity) of the capacitor. In the circuit shown in FIG. 1, there are two resonant loops (the "resonant loop of loop I" and the "resonant loop of loop II") in which the resonant current flowing through the multi-resonant circuit can circulate. As for frequency, the resonant frequency of loop I is ω l , the resonant frequency of loop II is ω h Let us assume that.

[0012] However, as shown in the experimental results in Non-Patent Document 1, the two oscillation modes (resonance loops) may switch unintentionally during operation. When the oscillation mode (resonance loop) switches, the operation becomes unstable and power transmission itself becomes difficult, so a method for reliably selecting one of the oscillation modes (resonance loops) was needed.

[0013] As one method for solving such problems, Patent Document 1 proposes a method for selecting one of the resonant loops by adjusting the magnitude of the Q value (selectivity) of the two resonant loops through coil design in a magnetic resonance type wireless power supply device that utilizes PT symmetry to supply power contactlessly by magnetically resonating a power supply coil and a power receiving coil. [Prior art documents] [Patent documents]

[0014] [Patent Document 1] JP 2022-121324 A [Non-patent literature]

[0015] [Non-Patent Document 1] Sid Assawaworrarit, Xiaofang Yu & Shanhui Fan, “Robust wireless power transfer using a nonlinear parity-time-symmetric circuit”, Nature, 15 JUNE 2017, volume 546, p.387-390 [Non-Patent Document 2] APSample, DTMeyer & JRSmith, “Analysis, experimental results, and range adaptation of magnetically coupled resonators for wireless power transfer”, IEEE Trans. Ind. Electron., 2011, vol.58, no.2, pp.544-554 [Non-Patent Document 3] J. Zhou, B. Zhang, W. Xiao, D. Qiu, and Y. Chen, “Nonlinear parity-time-symmetric model for constant efficiency wireless power transfer: application to a drone-in-flight wireless charging platform”, IEEE Trans. Ind. Electron., Aug. 2019, vol.66, no.5, pp.4097-4107 [Non-Patent Document 4] H. Ishida, T. Kyoden, and H. Furukawa, “Application of parity-time symmetry to low-frequency wireless power transfer system”, IEEJ J. Ind. Appl., 2022, vol.11, no.1, pp.59-68 Summary of the Invention [Problem to be solved by the invention]

[0016] However, in a magnetic resonance type wireless power supply device as shown in Patent Document 1, for example, a method is adopted in which one of the two resonant loops is selected by adjusting the magnitude of the Q value (selectivity) of the two resonant loops through the coil design. This imposes restrictions on the design of the power supply coil and the power receiving coil, posing an application problem in that the coils cannot be freely shaped or sized.

[0017] The present invention has been made to solve the above-mentioned problems, and has an object to provide a magnetic resonance type wireless power feeder that utilizes PT symmetry to feed power contactlessly by magnetically resonating a feed coil and a receiving coil, and that is capable of selecting and fixing one of two resonant loops in a manner that is not influenced by the surrounding environment and does not restrict the design of the feed coil and the receiving coil. [Means for solving the problem]

[0018] In order to achieve the above object, the present invention provides a magnetic resonance type wireless power feeder that includes a power feed coil and a power receiving coil, and that utilizes Parity-Time symmetry to feed power in a non-contact manner by magnetically resonating the power feed coil and the power receiving coil. When the power feed coil and the power receiving coil are regarded as a multi-resonant circuit coupled to each other by mutual inductance, there are two resonant loops (a "resonant loop of loop I" and a "resonant loop of loop II") through which a resonant current flowing in the multi-resonant circuit can circulate. The power feed coil includes an inverter, and a current or magnetic field of the power feed coil is connected to the multi-resonant circuit. a sensor that detects the phase relationship between the AC voltage applied to the power supply side resonant circuit and the AC current flowing through the power supply coil is connected to the inverter, and the inverter is equipped with a phase adjustment circuit that can adjust the phase relationship between the AC voltage applied to the power supply side resonant circuit and the AC current flowing through the power supply coil based on a current phase based on the current of the power supply coil detected by the sensor or a magnetic field phase based on the magnetic field of the power supply coil, and a switch timing, which is the time to turn on or off a switching element inside the inverter, is determined based on a pulse generated by a signal after the phase relationship is adjusted by the phase adjustment circuit, thereby selecting and fixing one of the two resonant loops. Effect of the Invention

[0019] According to this invention, in a magnetic resonance type wireless power supply device that utilizes PT symmetry to supply power contactlessly by magnetically resonating a power supply coil and a power receiving coil, a phase adjustment circuit is provided to adjust the phase relationship between the AC voltage applied to the power supply side resonant circuit and the AC current flowing through the power supply coil. This makes it possible to select and fix one of the two resonant loops in a manner that is not influenced by the surrounding environment and does not restrict the design of the power supply coil and the power receiving coil. [Brief description of the drawings]

[0020] [Figure 1] FIG. 1 shows an equivalent circuit of an SP topology. [Diagram 2]FIG. 1 is a conceptual diagram showing two typical circuit configurations of wireless power transfer using magnetic resonance technology that utilizes PT symmetry. [Diagram 3] 1 is a graph showing a result of a numerical calculation showing a typical example of the relationship between the magnetic coupling coefficient km and the transmission distance d between two coils (a power supply coil and a power receiving coil). [Figure 4] 13 is a graph showing the relationship between the resonant frequency ωh of loop II and the resonant frequency ωl of loop I and the transmission distance d between two coils (a power supply coil and a power receiving coil). [Diagram 5] This is a circuit diagram of a class D inverter connected to a power supply side resonant circuit as an inverter used for PT symmetric wireless power supply. [Figure 6] 6 is a schematic diagram showing the phase relationship of each waveform of the power supply side resonant circuit shown in FIG. 5. [Figure 7] 4 shows the experimental results showing the phase relationship between AC current i1 and input voltage v1. [Figure 8] FIG. 2 is a circuit diagram showing an equivalent circuit of a wireless power supply connected to an inverter. [Figure 9] 1 is a circuit diagram showing an example of an inverter having a phase adjustment circuit and a power supply side resonant circuit according to a first embodiment of the present invention. [Figure 10] 10 is a schematic diagram showing voltage waveforms at various points in the phase adjustment circuit (preamplifier 36 and phase delay circuit 35) shown in FIG. [Figure 11] 1 is a circuit diagram showing an example of an inverter having a time delay circuit and a power supply side resonant circuit according to a first embodiment of the present invention. [Figure 12] 1 is a circuit diagram showing an example of an inverter having an all-pass filter and a power supply side resonant circuit in accordance with a first embodiment of the present invention. [Figure 13] 1 is a circuit diagram showing an example of an inverter having a PLL (phase-locked loop) and a power supply side resonant circuit according to a first embodiment of the present invention. [Figure 14] 1 is a block diagram showing an example of a specific internal function of a PLL (phase-locked loop) in accordance with a first embodiment of the present invention. [Figure 15]1 is a circuit diagram showing an example of an inverter having an all-pass filter and a PLL (phase-locked loop), and a power supply side resonant circuit in accordance with a first embodiment of the present invention. [Figure 16] 11 is a graph showing measured waveforms of AC current i1 and input voltage v1 when the input voltage v1 is adjusted to have a leading phase with respect to the AC current i1 in an actual device according to the first embodiment of the present invention. [Figure 17] 17 is a graph showing the results of measuring the change in resonant frequency when the distance between the power supply coil 11 provided in the power supply side resonant circuit 1 and the power receiving coil 21 provided in the power receiving side resonant circuit 2, i.e., the transmission distance d between these two coils, is changed in the phase relationship of FIG. [Figure 18] 13 is a graph showing the results of measuring the transmitted power when the transmission distance d between the two coils is changed. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0021] The present invention relates to a magnetic resonance type wireless power feeder that utilizes parity-time symmetry (hereinafter referred to as "PT symmetry") to feed power in a non-contact manner by magnetically resonating a feeding coil and a receiving coil. Hereinafter, an embodiment of the present invention will be described in detail with reference to the drawings.

[0022] Embodiment 1 FIG. 2 is a conceptual diagram showing two typical circuit configurations of wireless power transfer using magnetic resonance method utilizing PT symmetry. In both circuit examples shown in FIG. 2(a) and (b), a power supply side resonant circuit 1 is connected to an inverter 3 as an AC power source. On the other hand, a power receiving side resonant circuit 2 is connected to a load resistor R L In these circuits, L1 represents the self-inductance of the power supply coil 11, C1 represents the capacitance of the power supply side capacitor, L2 represents the self-inductance of the power receiving coil 21, and C2 represents the capacitance of the power receiving side capacitor. Additionally, r1 and r2 represent the resistance components included in the power supply side resonant circuit 1 and the power receiving side resonant circuit 2, respectively.

[0023] The circuit example shown in Fig. 2(a) is called an SS topology because the coil and capacitor of the power supply side resonant circuit 1 are connected in series, and the coil and capacitor of the power receiving side resonant circuit 2 are also connected in series. The circuit example shown in Fig. 2(b) is called an SP topology because the coil and capacitor of the power supply side resonant circuit 1 are connected in series, and the coil and capacitor of the power receiving side resonant circuit 2 are connected in parallel. Non-Patent Document 3 discloses an example of wireless power supply in which PT symmetry is preserved in an SS topology, and Non-Patent Document 4 discloses an example of wireless power supply in which PT symmetry is preserved in an SP topology.

[0024] Here, to repeat, taking the SP topology equivalent circuit shown in FIG. 1 as an example, the power supply side resonant circuit 1 and the power receiving side resonant circuit 2 of the SP topology have a mutual inductance k m The L in FIG. 1 represents the self-inductance of the power supply coil 11 and the power receiving coil 21. L(1-k m ) represent the leakage inductance of the power supply coil 11 and the power receiving coil 21, respectively. Also, r1' and r2' represent the winding resistance of the power supply coil 11 and the power receiving coil 21, respectively. r c represents the iron loss equivalent resistance. C represents the capacitance (electrostatic capacity) of the capacitor. In the circuit shown in FIG. 1, there are two resonant loops (the "resonant loop of loop I" and the "resonant loop of loop II") in which the resonant current flowing through the multi-resonant circuit can circulate. As for frequency, the resonant frequency of loop I is defined as ω l , the resonant frequency of loop II is ω h Let us assume that.

[0025] Also, as described above, the wireless power feeder utilizing the PT symmetry in the first embodiment of the present invention is the same as the conventional one in that the inverter is self-oscillated and used, there are two oscillation modes, and the PT symmetry can be preserved in either mode. It goes without saying that the power feed side resonant circuit 1 includes the power feed coil 11, and the power receiver side resonant circuit 2 includes the power receiver coil 21, as shown in Figs. 1 and 2.

[0026] In either of the topologies shown in Fig. 2(a) and (b), the wireless power supply circuit viewed from the output end of the inverter 3 forms a multi-resonant circuit, and two resonant loops exist, but the exact mathematical expressions for the two resonant loops cannot be derived from the circuit diagram shown in Fig. 2. Strictly speaking, this can be found from the coupled mode theory, and the derivation process is as shown in Non-Patent Document 4 (as shown in Non-Patent Document 4 by one of the applicants of the present application), but here we will omit a detailed explanation and the derivation process and describe only the results, which are shown in Equations (1) and (2).

[0027]

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[0028]

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[0029] That is, ω shown in Eq. (1) h A resonant loop with a resonant frequency of ω l There is a resonant loop with a resonant frequency of ω h >ω l ω0 is the natural angular resonance frequency when the power supply resonant circuit and the power receiving resonant circuit are far apart and not completely coupled, and can be expressed by equation (3). Note that Γ in equations (1) and (2) 20 and Γ Lare all parameters in the coupled mode theory (CMT parameters), and as explained in detail in Patent Document 1 (as one of the applicants of the present application has shown in Patent Document 1 and Non-Patent Document 4), detailed explanations of these are omitted here because they are not very relevant to the explanation of the present invention and the explanation of the definitions would be too long.

[0030]

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[0031] In addition, k is a parameter called a coupling rate, which is a parameter related to the number of times energy is exchanged between the power supply side resonant circuit and the power receiving side resonant circuit per unit time. Furthermore, the coupling rate k is expressed as the magnetic coupling coefficient k between the power supply coil and the power receiving coil. m Using this, it can be expressed as equation (4).

[0032]

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[0033] Here, for example, in the case of the SP topology of FIG. 2(b), if equations (1) and (2) are applied, they can be expressed as the following equations (5) and (6).

[0034]

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[0035]

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[0036] The above formulas (5) and (6) can also be approximately expressed as formulas (7) and (8). In Patent Document 1, these approximate formulas (7) and (8) are used in the explanation.

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[0039] From equations (5) and (6), the resonant frequency ω of loop II is h and the resonant frequency ω of loop I l is the magnetic coupling coefficient k m In the case of a complex coil shape, the magnetic coupling coefficient k m It is difficult to formulate the relationship between the distance d (transmission distance) between the power supply coil and the power receiving coil, and it is generally obtained by numerical calculation using a computer. m The relationship that is inversely proportional to the transmission distance d between the coils is universal. m 13 is a graph showing a result of a numerical calculation showing a typical example of the relationship between the power supply coil and the transmission distance d between two coils (a power supply coil and a power receiving coil).

[0040] As shown in Figure 3, the magnetic coupling coefficient k m The relationship between the transmission distance d between the two coils (the power supply coil and the power receiving coil) can be calculated numerically. From equations (5) and (6), the resonant frequency ω h and the resonant frequency ω of loop I l The relationship between the resonant frequency ω of loop II and the transmission distance d between the two coils (the power supply coil and the power receiving coil) can also be calculated. h and the resonant frequency ω of loop I l 4 is a graph showing the relationship between the transmission distance d between two coils (a power supply coil and a power receiving coil).

[0041] As shown in Figure 4, the resonant frequency ω h As the transmission distance d between the two coils becomes longer, the resonant frequency ω lbecomes higher as the transmission distance d becomes longer. In the example shown in Fig. 4, the resonant frequencies of the two resonant loops match when the transmission distance d between the two coils is about 70 mm. Since the PT symmetry is not preserved at transmission distances beyond the match, these are excluded from the discussion of the present invention.

[0042] As mentioned above, in the case of PT symmetric wireless power transfer, the inverter is used in self-oscillation, so the resonant frequency ω h and the resonant frequency ω of loop I l One of the resonant loops is selected and self-oscillation occurs. However, when the transmission distance d changes during operation, the resonant frequency ω h and the resonant frequency ω of loop I l If the mode is suddenly switched unintentionally, the frequency of the self-oscillation will suddenly and greatly fluctuate (discontinuously), and the power transmission will be interrupted. In addition, since the magnitude of the transmitted power and the power transmission efficiency differ depending on the mode, the resonant frequency ω h and the resonant frequency ω of loop I l There is also a problem in that a step occurs in the magnitude of the transmission power when the switching occurs.

[0043] Therefore, in this invention, even if the transmission distance d varies, if the resonant loop is fixed to one side, the self-oscillation frequency changes smoothly (continuously), which is the idea that stable wireless power supply is possible. Based on this idea, the resonant frequency ω h or the resonant frequency ω of loop I l In order to select and fix one of the resonant loops having the resonant loops, a phase adjustment circuit is provided to advance / delay the phase of the voltage relative to the current of the power supply coil 11.

[0044] First, the circuit configuration of an inverter without a phase adjustment circuit as in the present invention will be described. A class D inverter or a class E inverter is used as an inverter for PT symmetric wireless power supply. FIG. 5 is a circuit diagram of a case where a class D inverter is connected to a power supply side resonant circuit as an inverter for PT symmetric wireless power supply. In reality, there is a power receiving side resonant circuit that is wirelessly supplied with power from the power supply side resonant circuit 1 shown in FIG. 5, and a load resistor connected to the power receiving side resonant circuit, but these are omitted in FIG. 5. FIG. 6 is a schematic diagram showing the phase relationship of each waveform of the power supply side resonant circuit 1 shown in FIG. 5.

[0045] As explained in the section on the prior art, wireless power transfer using PT symmetry replaces the AC power supply in conventional magnetic resonance wireless power transfer technology with an inverter that behaves electrically in the same way as a negative resistor, that is, an inverter that behaves as a negative resistor.

[0046] An inverter that behaves as a negative resistance is an inverter whose switching frequency and voltage amplitude are not fixed in advance, has a circuit configuration in which the switching frequency is determined by the apparent resonant frequency of the wireless power supply circuit as seen from the output end of the inverter, and has a switching frequency that quickly responds to changes in the apparent resonant frequency of the wireless power supply circuit, which may change due to changes in the transmission distance between two coils (a power supply coil and a power receiving coil) or positional deviation. The wireless power supply circuit is a circuit that includes a power supply side resonant circuit and a power receiving side resonant circuit, and all circuits connected to them. The apparent resonant frequency means the actual resonant frequency that takes into account the interaction between the power supply side resonant circuit and the power receiving side resonant circuit.

[0047] 5, an inverter 3 and a current sensor 4 are connected to the power supply side resonant circuit 1, and the inverter 3 and the current sensor 4 realize a negative resistance circuit. Elements constituting the inverter 3 include a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34. The high-side FET 33 and the low-side FET 34 are switching elements within the inverter 3.

[0048] 5 receives as input a detection signal of the AC current i1 flowing through the power feeding coil 11 detected by the current sensor 4, and when an input pulse that is the output of the comparator 31 is input to the gate driver 32, the gate driver 32 outputs two-phase output pulses (a first-phase output pulse and a second-phase output pulse) with a phase difference of 180° for alternately energizing / de-energizing (turning on / off) the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3. The input pulse input to the gate driver 32 is generated based on the AC current i1 flowing through the power feeding coil 11 or the AC magnetic field generated in the power feeding coil 11.

[0049] At this time, the switch timing of the output pulses alternately output from the gate driver 32 to the high-side FET 33 and low-side FET 34, which are switching elements inside the inverter 3, is determined based on the input pulse to the gate driver 32. Here, the "switch timing" refers to the time at which the high-side FET 33 and low-side FET 34, which are switching elements inside the inverter 3, are turned on or off. Then, a detection signal of the AC current i1 detected by the current sensor 4 installed between the inverter 3 and the power supply side resonant circuit 1 is input to the comparator 31, which generates the above-mentioned input pulse.

[0050] That is, the drive current of the power supply coil 11 in the power supply side resonant circuit 1, which is the output of the inverter 3, is fed back to the control side of the inverter 3 to control the output current of the inverter 3, which is called "feedback control". Here, the detection signal of the AC current i1 is an AC voltage signal in phase with the AC current i1. In principle, the AC current i1 and the input voltage v1 of the power supply resonant circuit are in phase (no phase difference), so self-oscillation occurs due to positive feedback. This self-oscillation excites the power supply coil, realizing wireless power supply.

[0051] However, in an actual circuit, the AC current i1 and the input voltage v1 are not in phase. In reality, a time delay occurs in the process in which the voltage signal propagates through the circuit, so the input voltage v1 lags behind the AC current i1. This phase difference is called the initial phase difference. The phase compensation circuit described in Patent Document 1 is a circuit for compensating for this initial phase difference and making the AC current i1 and the input voltage v1 in phase, and has a significantly different purpose from that of the present invention.

[0052] As mentioned above, making the AC current i1 and the input voltage v1 in phase is theoretically correct, but actually making the AC current i1 and the input voltage v1 in phase causes the problem of frequent switching between the two resonant loops as mentioned above. Conversely, experiments have shown that if a phase difference is intentionally created between the AC current i1 and the input voltage v1, one of the resonant loops becomes stable.

[0053] FIG. 7 shows the experimental results showing the phase relationship between AC current i1 and input voltage v1. FIG. 7(a) shows the case where the input voltage v1 lags behind AC current i1, FIG. 7(b) shows the case where AC current i1 and input voltage v1 are in phase, and FIG. 7(c) shows the case where the input voltage v1 leads AC current i1.

[0054] As shown in FIG. 7(a), when the input voltage v1 lags the AC current i1, the resonant frequency ω lAs shown in FIG. 7(b), when the AC current i1 and the input voltage v1 are almost in phase, the resonance loop becomes unstable. Also, as shown in FIG. 7(c), when the input voltage v1 leads the AC current i1, the resonance frequency ω h The resonant loop is selected.

[0055] Next, the principle of selecting a resonant loop by adjusting the phase relationship between the AC current i1 and the input voltage v1 will be described. As described above, the inverter of the wireless power supply device in the first embodiment of the present invention self-oscillates by positively feeding back the current waveform of the AC current i1. Therefore, this inverter can be regarded as an AC power source with an unfixed oscillation frequency. As shown in Fig. 5, the inverter 3 is connected to the power supply side resonant circuit 1. The power receiving side resonant circuit 2 (see Figs. 1 and 2) and the load resistance R L (see FIGS. 1 and 2) are also indirectly connected to the inverter 3.

[0056] The circuit seen from the output end of the inverter 3 forms a resonant circuit, and this resonant circuit is represented as a simple equivalent circuit in Fig. 8. Fig. 8 is a circuit diagram showing an equivalent circuit of a wireless power supply connected to the inverter. Here, the resistance component of the circuit is R, the inductance component is L, and the capacitance component is C. The inverter has a resonant frequency ω c When self-oscillation occurs, the magnitudes of the inductive reactance and capacitive reactance must satisfy the following equation (9).

[0057]

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[0058] Therefore, the reactance component of the resonant circuit seen from the output terminal of the inverter is zero, and it is equivalent to a circuit in which only resistance R is connected to the inverter. This is the original equivalent circuit when PT symmetry is preserved. And, as mentioned above, in the original equivalent circuit, since only resistance component R is connected to the inverter, AC current i1 and input voltage v1 are in phase. However, in the present invention, a phase difference between the AC current i1 and the input voltage v1, which are essentially in phase, is forcibly created.

[0059] For example, as shown in Figure 7(a), if the input voltage v1 is delayed with respect to the AC current i1, the equivalent circuit connected to the inverter becomes capacitive. In other words, if the input voltage v1 is delayed with respect to the AC current i1, the resonant frequency ω becomes smaller than the original resonant frequency ω c It can also be said that if the equivalent circuit connected to the inverter deviates slightly from the normal, the equivalent circuit becomes capacitive.

[0060] On the other hand, as shown in FIG. 7(c), when the input voltage v1 is advanced relative to the AC current i1, the resonant frequency ω becomes equal to the original resonant frequency ω c If the input voltage is slightly off, the equivalent circuit connected to the inverter becomes inductive. The impedance Z of the equivalent circuit connected to the inverter can be expressed by the following equation (10): The inductive condition is given by equation (11).

[0061]

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[0063] Furthermore, from equation (9), the inductance component L can be expressed as equation (12). By substituting equation (12) into equation (11), the inductive condition becomes equation (13).

[0064]

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[0065]

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[0066] In this way, when the input voltage v1 is advanced relative to the AC current i1, the resonant frequency ω at that time is the original resonant frequency ω c will be slightly higher than

[0067] As mentioned above, the resonant frequency ω c is the resonant frequency ω of loop II h or the resonant frequency ω of loop I l Also, ω h >ω l Therefore, as shown in equation (13), c A sufficient condition for satisfying the relation is ω>ω h However, ω is ω h Since ω is a slightly higher frequency than ω h In other words, if the input voltage v1 advances with respect to the AC current i1, the resonant frequency ω h Therefore, the resonant loop is selected as follows.

[0068] On the other hand, the capacitive condition is given by equation (14). From equations (9) and (14), the capacitive condition is given by equation (15).

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[0071] In this way, when the input voltage v1 is delayed with respect to the AC current i1, the resonant frequency ω at that time is the original resonant frequency ω c will be slightly lower than

[0072] As mentioned above, the resonant frequency ω c is the resonant frequency ω of loop II h or the resonant frequency ω of loop I l Also, ω h >ω l Therefore, as shown in equation (15), c A sufficient condition for satisfying the relation is ω<ω l However, ω is ω l Since ω is a slightly lower frequency than ω l In other words, if the input voltage v1 is delayed with respect to the AC current i1, the resonant frequency ω l Therefore, the resonant loop is selected as follows.

[0073] An actual circuit example will now be described. As mentioned above, it has been confirmed that if a phase difference is intentionally provided between the AC current i1 and the input voltage v1, it can be stabilized in one of the resonant loops (see Figs. 16 to 18 for the experimental results). Therefore, the resonant frequency ω h To advance the input voltage v1 relative to the AC current i1 in order to select and fix the resonant loop, a phase lead circuit can be inserted before the AC current detected by the current sensor is input to the comparator. However, a phase lead circuit also functions as a high-pass filter, preferentially passing high-frequency noise components, which worsens the S / N ratio and makes it unsuitable for stable operation.

[0074] On the other hand, the phase delay circuit also functions as a low-pass filter and blocks high-frequency noise components, making it suitable for stable operation. Of course, since it is a phase delay circuit, it is not possible to create a phase lead circuit. Therefore, in an actual circuit, as shown in Figure 9, a phase adjustment circuit 30 is formed by combining a preamplifier 36 and a phase delay circuit 35 to realize a phase lead. This makes it possible to selectively fix one of the two resonant loops and realize stable operation.

[0075] Fig. 9 is a circuit diagram showing an example of an inverter having a phase adjustment circuit and a power supply side resonant circuit in the first embodiment of the present invention. As shown in Fig. 9, an inverter 3 is connected to a power supply side resonant circuit 1, and the elements constituting the inverter 3 are the same as those in the circuit diagram shown in Fig. 5 in that they include a comparator 31, a gate driver 32, a high side FET (field effect transistor) 33, and a low side FET (field effect transistor) 34.

[0076] Also, in Figure 9, only the power supply side resonant circuit 1 and the inverter 3 and current sensor 4 connected thereto are shown, but the premise is the same as in Figures 1 and 2, that is, a magnetic resonance type wireless power supply device that has a power supply side resonant circuit 1 including a power supply coil 11 and a power receiving side resonant circuit 2 including a power receiving coil 21, and uses PT symmetry (Parity-Time symmetry) to supply power contactlessly by magnetically resonating the power supply coil 11 and the power receiving coil 21, and when the power supply side resonant circuit 1 and the power receiving side resonant circuit 2 are considered to be a multi-resonant circuit coupled to each other by mutual inductance, there are two resonant loops (a "resonant loop of loop I" and a "resonant loop of loop II") through which the resonant current flowing in the multi-resonant circuit can circulate.

[0077] 9, that is, in one example of the circuit in the first embodiment of the present invention, a phase adjustment circuit 30 including a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35 is provided before the input to the comparator 31 provided inside the inverter 3, and the AC current i1 based on the detection of the current or magnetic field of the power feeding coil 11 is delayed by more than 180° by the phase adjustment circuit 30 (the inverting amplifier circuit 36 ​​and the phase delay circuit 35) before being input to the comparator 31, thereby performing phase control to control the AC current i1 as an advanced phase. This makes it possible to select and fix the "resonant loop of loop II" out of the two resonant loops.

[0078] 9 (preamplifier (inverting amplifier circuit) 36 and phase delay circuit 35) are not provided, a detection signal of the AC current i1 flowing through the power feeding coil 11 detected by the current sensor 4 is input to the comparator 31 of the inverter 3, and when an input pulse output from the comparator 31 is input to the gate driver 32, the gate driver 32 outputs two-phase output pulses (a first-phase output pulse and a second-phase output pulse) with a phase difference of 180° for alternately energizing / deenergizing (turning on / off) the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3. The input pulse input to the gate driver 32 is generated based on the AC current i1 flowing through the power feeding coil 11 or the AC magnetic field generated in the power feeding coil 11.

[0079] 9, however, a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35 are provided before the input to the comparator 31, and a signal in which the AC current i1 has been delayed by more than 180° by the inverting amplifier circuit 36 ​​and the phase delay circuit 35 is input to the comparator 31 before the AC current i1 based on the detection of the current or magnetic field of the power feeding coil 11 is input to the comparator 31. When an input pulse that is the output of the comparator 31 is input to the gate driver 32, the gate driver 32 outputs two-phase output pulses (a first-phase output pulse and a second-phase output pulse) with a phase difference of 180° for alternately energizing / de-energizing (turning on / off) the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, and the input pulses input to the gate driver 32 are generated based on the AC current i1 flowing in the power feeding coil 11 or the AC magnetic field generated in the power feeding coil 11, just like the inverter 3 shown in FIG.

[0080] At this time, the switch timing of the output pulse alternately output from the gate driver 32 to the high-side FET 33 and low-side FET 34, which are switching elements inside the inverter 3, is determined based on the input pulse to the gate driver 32. Here, as described above, the "switch timing" refers to the time at which the high-side FET 33 and low-side FET 34, which are switching elements inside the inverter 3, are turned on or off. Then, the detection signal of the AC current i1 detected by the current sensor 4 installed between the inverter 3 and the power supply side resonant circuit 1 is phase-adjusted by the preamplifier (inverting amplifier circuit) 36 and the phase delay circuit 35, and the signal is input to the comparator 31 to generate the above-mentioned input pulse.

[0081] That is, in the case of FIG. 9, the switch timing of the output pulses alternately output from the gate driver 32 to the high-side FET 33 and low-side FET 34, which are switching elements inside the inverter 3, is determined based on an input pulse generated by inputting to the comparator 31 a signal after phase control has been performed by feedback control of a leading phase or lagging phase based on the current phase or magnetic field phase of the power feeding coil 11 detected by the current sensor 4, that is, a signal after the phase relationship has been adjusted by the phase adjustment circuit 30.

[0082] Here, since the preamplifier 36 is an inverting amplifier circuit, when the detection signal of the AC current i1 is input to the preamplifier 36, the phase of the output signal is inverted by 180° with respect to the input signal, as shown in FIG. 10. FIG. 10 is a schematic diagram showing the voltage waveforms of each part of the phase adjustment circuit 30 (preamplifier 36 and phase delay circuit 35) shown in FIG. 9. Thereafter, the phase is further delayed at the output of the phase delay circuit (low-pass filter) 35 using an operational amplifier. For example, when a phase delay of 130° is generated in the phase delay circuit 35, the phase delay including the preamplifier 36 is 180+130=310°. In other words, since the phase is advanced by 360-310=50° with a delay of one period (360°), an equivalent phase advance can be realized. In other words, the AC current i1 is delayed by more than 180° and input to the comparator 31.

[0083] 9, the inverter 3 and the current sensor 4 are described as being used as the negative resistance circuit connected to the power supply side resonant circuit 1, but a magnetic sensor may be used instead of the current sensor 4. Since the current phase and the magnetic field phase coincide with each other, the current may be detected by the current sensor 4, the current phase may be detected, and the AC current may be detected based on the current phase, or the magnetic field may be detected by the magnetic sensor, the magnetic field phase may be detected, and the AC current may be detected based on the magnetic field phase. In other words, the sensor constituting the negative resistance circuit together with the inverter 3 may be any sensor capable of detecting the current or magnetic field of the power supply coil 11 in the power supply side resonant circuit 1. This also applies to FIG. 5, FIG. 8, and FIG. 11 and subsequent figures described later.

[0084] 9 is a device that detects the current or magnetic field of the power supply coil 11 to oscillate the inverter 3 connected to the power supply side resonant circuit 1, detects zero crossings of the voltage or current of the power supply coil 11 based on the current or magnetic field, and causes self-oscillation. h In order to select and fix a resonant loop having a resonant loop ("resonant loop of loop II"), the AC current i1 detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is delayed by more than 180° before being input to the comparator 31 in the inverter 3, and thus the voltage phase is controlled as a leading phase, by inserting a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35. In other words, the leading phase is realized by rotating the lag phase by more than 180°, and as a result, the problem that the leading phase control is vulnerable to noise and thus causes unstable operation can be solved.

[0085] In this way, the power supply side resonant circuit 1 is connected to the inverter 3 and a sensor (current sensor 4 in FIG. 9) that detects the current or magnetic field of the power supply coil 11, and the inverter 3 is equipped with a phase adjustment circuit 30 that performs phase control by feedback control of leading or lagging phase based on the current phase based on the current of the power supply coil 11 detected by the sensor (or the magnetic field phase based on the magnetic field of the power supply coil 11), i.e., the phase adjustment circuit 30 that can adjust the phase relationship between the AC voltage applied to the power supply side resonant circuit 1 and the AC current flowing through the power supply coil 11, and the switch timing, which is the time to turn on or off the switching element inside the inverter 3, is determined based on a pulse generated by a signal after the phase relationship has been adjusted by the phase adjustment circuit 30, thereby selecting and fixing one of the two resonant loops.

[0086] 9, a comparator 31 is provided inside the inverter 3, and a phase delay circuit 35 is provided as a phase adjustment circuit 30 before the input to the comparator 31. In the inverter 3, the phase delay circuit 35 delays the AC current based on the current or magnetic field of the power feeding coil 11 detected by a sensor (current sensor 4 in FIG. 9) by more than 180° and inputs it to the comparator 31 before the AC current is input to the comparator 31, in order to select the "resonance loop of loop II" out of the two resonant loops. This adjusts the phase relationship by leading the phase of the AC voltage applied to the power feeding side resonant circuit 1 with respect to the AC current flowing through the power feeding coil 11.

[0087] 9, a low-pass filter using an operational amplifier is used as the phase delay circuit 35 as a specific example of the phase adjustment circuit 30, but if a large phase delay of about 130° is generated by the low-pass filter, the gain (output amplitude / input amplitude) of the low-pass filter becomes too small, which may cause a problem that the amplitude of the output signal becomes too small. In that case, a method of using a time delay circuit instead of the low-pass filter is available.

[0088] Fig. 11 is a circuit diagram showing an example of an inverter having a time delay circuit and a power supply side resonant circuit in the first embodiment of the present invention. As shown in Fig. 11, an inverter 3 is connected to a power supply side resonant circuit 1, and the elements constituting the inverter 3 are the same as those in the circuit diagrams shown in Figs. 5 and 9 in that they include a comparator 31, a gate driver 32, a high side FET (field effect transistor) 33, and a low side FET (field effect transistor) 34.

[0089] 11, that is, another example of the circuit in the first embodiment of the present invention, a preamplifier (inverting amplifier circuit) 36 is provided before the input to the comparator 31 provided inside the inverter 3, and a time delay circuit 37 is provided after the output of the comparator 31, so that phase control is performed by controlling the delay time. In the case of FIG. 11, the phase adjustment circuit 30 includes at least the time delay circuit 37.

[0090] That is, the circuit shown in FIG. 11 has a resonant frequency ω h In order to select and fix a resonant loop having this resonant loop ("resonant loop of loop II"), the AC current i1 detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is input to a comparator 31 in the inverter 3 in order to advance the phase of the voltage relative to the current of the power supply coil 11. A time delay circuit 37 is inserted after the output of the comparator 31, and phase control is performed by controlling the delay time.

[0091] A large phase delay of about 130° is generated by connecting multiple time delay circuits 37, which combine a NOT type logic element with a simut trigger and an RC integrator circuit, in the rear of the comparator 31. In other words, real-time control is achieved by using a NOT circuit and an RC integrator circuit to control the switch signal input to the comparator 31, and as a result, the problem that the phase delay of an analog circuit is frequency dependent can be solved. Note that the logic element does not necessarily have to have a simut trigger. Also, the logic element can be realized by using a buffer circuit instead of a NOT type.

[0092] 11, a comparator 31 is provided inside the inverter 3, and an AC current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 11) is input to the comparator 31, and a time delay circuit 37 serving as a phase adjustment circuit 30 is provided after the output of the comparator 31. The inverter 3 controls the phase relationship by controlling the delay time by the time delay circuit 37, and thus one of the two resonance loops is selected and fixed.

[0093] In the actual circuit examples of Fig. 9 and Fig. 11, a current sensor 4 is attached to the connection line of the power supply coil 11 to detect the current of the power supply coil 11 (the current of the power supply side resonant circuit 1), thereby detecting the phase of the AC current i1. A similar effect can be achieved by attaching a magnetic sensor (not shown) near the power supply coil 11 to detect the magnetic field of the power supply coil 11 (the magnetic field of the power supply side resonant circuit 1). The magnetic sensor may be a Hall element or a magnetic sensor called a loop coil in which an electric wire is formed into a loop. In this case, the negative resistance circuit described above is constituted by the inverter 3 and the magnetic sensor.

[0094] 9, the phase is controlled by inserting a phase delay circuit (low-pass filter) 35 before the comparator 31, while in Fig. 11, the phase is controlled by inserting a time delay circuit 37 after the comparator 31. As another possible embodiment, it is possible to use a circuit capable of controlling only the phase, called an all-pass filter, instead of the time delay circuit 37 shown in Fig. 11.

[0095] Fig. 12 is a circuit diagram showing an example of an inverter having an all-pass filter and a power supply side resonant circuit in the first embodiment of the present invention. As shown in Fig. 12, an inverter 3 is connected to a power supply side resonant circuit 1, and the elements constituting the inverter 3 are the same as those in the circuit diagrams shown in Figs. 5, 9 and 11 in that they include a comparator 31, a gate driver 32, a high side FET (field effect transistor) 33 and a low side FET (field effect transistor) 34.

[0096] However, in the case of the circuit of FIG. 12, that is, in yet another example of the circuit in embodiment 1 of the present invention, a preamplifier (inverting amplifier circuit) 36 is provided before the input to the comparator 31 provided inside the inverter 3, and an all-pass filter 38 is provided after the output of the comparator 31, and phase control is performed by this all-pass filter 38.

[0097] That is, the circuit shown in FIG. 12 has a resonant frequency ω h In order to select and fix a resonant loop having the resonant loop ("resonant loop of loop II"), the AC current i1 detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is input to a comparator 31 in the inverter 3 in order to advance the phase of the voltage relative to the current of the power supply coil 11. An all-pass filter 38 is inserted after the output of the comparator 31, and phase control is performed by this all-pass filter 38.

[0098] 12, a comparator 31 is provided inside the inverter 3, and an AC current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 12) is input to the comparator 31, and an all-pass filter 38 serving as a phase adjustment circuit 30 is provided after the output of the comparator 31. The inverter 3 controls the adjustment of the phase relationship by the all-pass filter 38, whereby one of the two resonant loops is selected and fixed.

[0099] Furthermore, as shown in Fig. 13, instead of the time delay circuit 37 shown in Fig. 11 and the all-pass filter 38 shown in Fig. 12, a method of controlling the phase using a PLL (phase-locked loop) 39 can be considered. Fig. 13 is a circuit diagram showing an example of an inverter having a PLL (phase-locked loop) and a power supply side resonant circuit in the first embodiment of the present invention. As shown in Fig. 13, an inverter 3 is connected to the power supply side resonant circuit 1, and the elements constituting the inverter 3 are the same as those in the circuit diagrams shown in Figs. 5, 9, 11, and 12 in that a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34.

[0100] However, in the case of the circuit of FIG. 13, that is, yet another example of the circuit in embodiment 1 of the present invention, a preamplifier (inverting amplifier circuit) 36 is provided before the input to the comparator 31 provided inside the inverter 3, and a PLL (phase-locked loop) 39 is provided after the output of the comparator 31, and phase control is performed by this PLL (phase-locked loop) 39.

[0101] That is, the circuit shown in FIG. 13 has a resonant frequency ω hIn order to select and fix a resonant loop having this resonant loop ("resonant loop of loop II"), the AC current i1 detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is input to a comparator 31 in the inverter 3 in order to advance the phase of the voltage relative to the current in the power supply coil 11. A PLL (phase-locked loop) 39 is inserted after the output of the comparator 31, and phase control is performed by controlling the phase lead / lag using this PLL (phase-locked loop) 39.

[0102] 13, a comparator 31 is provided inside the inverter 3, and an AC current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 13) is input to the comparator 31, and a PLL (phase-locked loop) 39 serving as a phase adjustment circuit 30 is provided after the output of the comparator 31. The inverter 3 controls the phase relationship by using the PLL (phase-locked loop) 39 to advance or delay the phase of the AC voltage applied to the power supply side resonant circuit 1 with respect to the AC current flowing through the power supply coil 11, thereby selecting and fixing one of the two resonant loops.

[0103] 14 is a block diagram showing an example of a specific internal function of the PLL (phase-locked loop) 39. As shown in FIG. 14, the PLL (phase-locked loop) 39 includes at least a PFD (phase detector) 91 and a VCO (voltage-controlled oscillator) 92. The PFD (phase detector) 91 calculates the phase difference between an input signal serving as a reference frequency and a feedback signal output from the VCO (voltage-controlled oscillator) 92, whose frequency changes according to the voltage, and inputs the phase difference to the VCO (voltage-controlled oscillator) 92 to synchronize the phases of the input signal and the output signal. Therefore, by applying some offset voltage to the input voltage of the VCO (voltage-controlled oscillator) 92, it is possible to limit the frequency range of the output signal of the PLL (phase-locked loop) 39.

[0104] For example, the resonant frequency ω of loop IIh The frequency range of is 70kHz to 90kHz, and the resonant frequency of loop I is ω l If the frequency range of is 50kHz to 70kHz, the resonant frequency ω of loop I can be set to 70kHz to 90kHz by setting the frequency range of the output signal of PLL (phase-locked loop) 39 to 90kHz. l Therefore, the resonant frequency of loop II, ω h In other words, by controlling the input voltage of the VCO (voltage controlled oscillator) 92, the oscillation frequency of the VCO (voltage controlled oscillator) 92 is limited, and the frequency range of the output signal of the PLL (phase-locked loop) 39 is limited, thereby improving the operational stability when one of the two resonant loops is selected.

[0105] Here, a signal obtained by inverting an AC signal based on the current detected by the current sensor 4 shown in FIG. 13 using a preamplifier (inverting amplifier circuit) 36 is input to a comparator 31, and an input pulse generated in the comparator 31 is an input signal to a PLL (phase-locked loop) 39. Therefore, when the output signal of the PLL (phase-locked loop) 39 is input to the gate driver 32, the frequency or phase can be controlled by controlling the input voltage of a VCO (voltage-controlled oscillator) 92 (controlling the control value of the VCO 92), as compared to the case of positive feedback without phase control. Utilizing this, the resonant frequency ω h or the resonant frequency ω of loop I l Alternatively, a limit may be imposed on the oscillation frequency of the VCO (Voltage Controlled Oscillator) 92 of the PLL (Phase Locked Loop) 39, and the resonant frequency ω h Or, the resonant frequency ω of loop I l It is also possible to control the oscillation so that only the first oscillation occurs.

[0106] In this way, a limit is placed on the range of variation of the oscillation frequency of the VCO (Voltage Controlled Oscillator) 92 in the PLL (Phase Locked Loop) 39, and the resonant frequency ω lSo, if we make it impossible to oscillate, the resonant frequency ω of loop II will inevitably be h Since only the resonant frequency ω h can be selected and fixed to. In other words, by using a PLL (phase-locked loop) 39 that imposes a frequency limit on a VCO (voltage-controlled oscillator) 92, the output frequency of the PLL (phase-locked loop) 39 can be limited, so that the resonant loop can be fixed to one of the frequencies, thereby preventing unstable switching.

[0107] This makes it possible to prevent the problem of the current loop randomly switching due to changes in the positions of the power supply coil and the power receiving coil during wireless power supply. As described above, by adding PLL (phase-locked loop) 39, not only is it possible to perform phase control using a PLL, but it also has the advantage of improving the stability of operation by reducing phase fluctuations and other issues, thereby enabling stable operation.

[0108] Furthermore, as shown in Fig. 15, instead of the time delay circuit 37 shown in Fig. 11, the all-pass filter 38 shown in Fig. 12, or the PLL (phase-locked loop) 39 shown in Fig. 13, a method of controlling the phase using both the all-pass filter 38 and the PLL (phase-locked loop) 39 may be considered. Fig. 15 is a circuit diagram showing an example of an inverter having an all-pass filter and a PLL (phase-locked loop) and a power supply side resonant circuit according to the first embodiment of the present invention. As shown in Fig. 15, the power supply side resonant circuit 1 is connected to an inverter 3, and the elements constituting the inverter 3 are the same as those in the circuit diagrams shown in Figs. 5, 9, 11, 12, and 13 in that a comparator 31, a gate driver 32, a high-side FET (field-effect transistor) 33, and a low-side FET (field-effect transistor) 34.

[0109] 16 is a graph showing the measured waveforms of AC current i1 and input voltage v1 when the input voltage v1 is adjusted to have a leading phase with respect to the AC current i1 in an actual device according to the first embodiment of the present invention, where the dashed line shows the AC current i1 and the solid line shows the input voltage v1. That is, FIG. 16 shows the phase difference between the AC voltage applied to the power supply side resonant circuit 1 and the AC current flowing through the power supply coil 11. FIG. 17 is a graph showing the results of measuring the change in resonant frequency when the distance between the power supply coil 11 provided in the power supply side resonant circuit 1 and the power receiving coil 21 provided in the power receiving side resonant circuit 2, i.e., the transmission distance d between these two coils, is changed in the phase relationship of FIG. 16, where the ● marks indicate the experimental results and the dashed lines indicate the calculation results.

[0110] In the experiment, when the phase difference shown in FIG. 16 is 30 to 35 degrees, that is, when the AC voltage v1 is made to lead the AC current i1 by 30 to 35 degrees, the resonant frequency ω h It was confirmed that the resonant frequency ω of loop II was selected and fixed at any transmission distance, i.e., even if the transmission distance d was changed, the resonant frequency ω h It can be seen that the resonant loop of is selected and fixed.

[0111] Figure 18 is a graph showing the results of measuring the transmitted power when the transmission distance d between the two coils is changed, with the circles indicating the experimental results and the dashed lines indicating the calculation results. Looking at Figure 18, it can be seen that power supply is always possible regardless of the transmission distance, that is, even when the transmission distance d is changed.

[0112] As described above, according to the present invention, in a magnetic resonance type wireless power feeder that utilizes PT symmetry and feeds power contactlessly by magnetically resonating a power feed coil and a power receiving coil, a phase adjustment circuit for adjusting the phase is provided, making it possible to select and fix a resonance loop without being influenced by the surrounding environment and without imposing constraints on the design of the power feed coil and the power receiving coil. That is, it is possible to select and fix one of the two resonance loops without being influenced by the surrounding environment and the selection of a frequency suitable for wireless power feed, the core shape and material of the coil, etc. (no adjustments in coil design are necessary), and without imposing constraints on the design of the power feed coil and the power receiving coil, and the coil can be freely shaped and sized.

[0113] It should be noted that, within the scope of the present invention, any of the components of the embodiments may be modified or omitted. [Industrial Applicability]

[0114] The magnetic resonance type wireless power supply device of the present invention can be widely applied to wireless power supply in a wide variety of environments, including not only short-range wireless power supply such as charging a mobile phone, but also environments surrounding water such as underwater drones, and environments surrounding a lot of metal such as power supply to equipment in a factory. [Explanation of symbols]

[0115] 1 Power supply side resonant circuit 2 Receiving side resonant circuit 3 Inverter 4 Current Sensor 11 Power coil 21 Receiving coil 30 Phase adjustment circuit 31 Comparator 32 Gate Driver 33 High-side FET 34 Low-side FET 35 Phase delay circuit 36 Preamplifier (inverting amplifier circuit) 37 Time Delay Circuit 38 Allpass Filter 39 PLL (phase locked loop) 91 PFD (Phase Detector) 92 VCO (Voltage Controlled Oscillator)

Claims

1. A magnetic resonance type wireless power supply device that includes a power supply side resonant circuit including a power supply coil and a power receiving side resonant circuit including a power receiving coil, and that uses parity-time symmetry to supply power in a non-contact manner by magnetically resonating the power supply coil and the power receiving coil, When the power supplying side resonant circuit and the power receiving side resonant circuit are regarded as a multi-resonant circuit mutually coupled by mutual inductance, there are two resonant loops ("resonant loop I" and "resonant loop II") through which a resonant current flowing in the multi-resonant circuit can circulate, an inverter and a sensor for detecting a current or a magnetic field of the power supply coil are connected to the power supply side resonant circuit; the inverter includes a phase adjustment circuit that adjusts a phase relationship between an AC voltage applied to the power feeding side resonant circuit and an AC current flowing through the power feeding coil, based on a current phase based on a current in the power feeding coil detected by the sensor or a magnetic field phase based on a magnetic field of the power feeding coil; A switch timing, which is a time to turn on or off a switching element inside the inverter, is determined based on a pulse generated by a signal after the phase relationship is adjusted by the phase adjustment circuit, and one of the two resonance loops is selected and fixed. A magnetic resonance type wireless power supply device.

2. a comparator is provided inside the inverter, and a phase delay circuit is provided as the phase adjustment circuit before an input to the comparator; In the inverter, for the purpose of selecting "resonance loop II" of the two resonant loops, the phase delay circuit delays the AC current based on the current or magnetic field of the power supply coil detected by the sensor by inputting the AC current to the comparator by more than 180°, thereby controlling the phase relationship to be adjusted as a leading phase that leads the phase of the AC voltage applied to the power supply side resonant circuit with respect to the AC current flowing through the power supply coil.

2. The magnetic resonance type wireless power feeder according to claim 1.

3. a comparator is provided inside the inverter, and an AC current based on the current or magnetic field of the power supply coil detected by the sensor is input to the comparator, and a time delay circuit serving as the phase adjustment circuit is provided after the output of the comparator; The inverter performs control to adjust the phase relationship by controlling the delay time of the time delay circuit.

2. The magnetic resonance type wireless power feeder according to claim 1.

4. a comparator is provided inside the inverter, and an AC current based on the current or magnetic field of the power supply coil detected by the sensor is input to the comparator, and an all-pass filter serving as the phase adjustment circuit is provided after the output of the comparator; The inverter performs control to adjust the phase relationship by the all-pass filter.

2. The magnetic resonance type wireless power feeder according to claim 1.

5. a comparator is provided inside the inverter, and an AC current based on the current or magnetic field of the power supply coil detected by the sensor is input to the comparator, and a PLL (phase-locked loop) serving as the phase adjustment circuit is provided after an output of the comparator, The inverter performs control to adjust the phase relationship by leading or lagging the phase of the AC voltage applied to the power supply side resonant circuit with respect to the AC current flowing through the power supply coil using the PLL (phase locked loop).

2. The magnetic resonance type wireless power feeder according to claim 1.

6. The PLL (phase-locked loop) includes at least a VCO (voltage-controlled oscillator), and limits the oscillation frequency of the VCO (voltage-controlled oscillator) by controlling the input voltage of the VCO (voltage-controlled oscillator), thereby limiting the frequency range of the output signal of the PLL (phase-locked loop).

6. The magnetic resonance type wireless power feeder according to claim 5.

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