Rotary electric machine control method, and rotary electric machine control device

Abstract

To provide a rotary electric machine control method capable of reducing current ripple when driving a wound field type rotary electric machine with one inverter, and a rotary electric machine control device.SOLUTION: A field winding current If which is a current flowing in a field winding 15 and stator winding currents (IU, IV and IW) which are currents flowing in a stator winding 16 of a rotary electric machine 10 are detected and based on the detected field winding current If and stator winding currents (IU, IV and IW), field magnetic flux Ψ is changed such that zero voltage timing that is timing in which stator winding voltages (VU, VV and VW) which are voltage applied to the stator winding 16 become zero is made close to zero current timing that is timing in which the stator winding currents (IU, IV and IW) become zero.SELECTED DRAWING: Figure 2

Description

[Technical field]

[0001] The present invention relates to a rotating electric machine control method and a rotating electric machine control device for controlling a wound-field type rotating electric machine. [Background technology]

[0002] Patent Document 1 discloses that when outputting a rectangular wave voltage to a load, two inverters are used and each inverter is controlled so that harmonic components of specific orders generated by the operation of each inverter cancel each other out. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Patent No. 7002924 Summary of the Invention [Problem to be solved by the invention]

[0004] Rotating electric machines such as electric motors are controlled by pulse width modulation control (hereinafter referred to as PWM control). At this time, ripples (hereinafter referred to as current ripples) occur in the current input from a battery, which is a DC power source, to an inverter that drives the rotating electric machine.

[0005] When an accessory is connected to a battery that supplies power to a rotating electric machine, the current ripple may flow into a smoothing capacitor (input capacitor) for the accessory, causing unnecessary heat generation. For this reason, it is particularly desirable to reduce the current ripple when an accessory is connected to a battery that supplies power to a rotating electric machine.

[0006] In this case, it is possible to use two inverters to drive the rotating electric machine and offset the current ripple generated by the operation of each inverter. However, providing a second inverter just to reduce the current ripple is usually not possible because it has significant disadvantages in terms of system size and cost.

[0007] An object of the present invention is to provide a rotating electric machine control method and a rotating electric machine control device that can reduce current ripples when a wound-field type rotating electric machine is driven by a single inverter. [Means for solving the problem]

[0008] One aspect of the present invention is a method for controlling a rotating electric machine in a system including a wound-field type rotating electric machine that generates a field flux by a field winding, an auxiliary machine, and a battery that supplies power to the rotating electric machine and the auxiliary machine. In this rotating electric machine control method, a field winding current that is a current flowing through the field winding and a stator winding current that is a current flowing through the stator winding of the rotating electric machine are detected, and the field flux is changed based on the detected field winding current and stator winding current, thereby bringing closer the zero voltage timing at which the stator winding voltage that is the voltage applied to the stator winding becomes zero and the zero current timing at which the stator winding current becomes zero. Effect of the Invention

[0009] According to the present invention, it is possible to provide a rotating electric machine control method and a rotating electric machine control device that can reduce current ripples when a wound-field type rotating electric machine is driven by a single inverter. [Brief description of the drawings]

[0010] [Figure 1] FIG. 1 is a block diagram showing a schematic configuration of a vehicle system. [Diagram 2] FIG. 2 is a block diagram showing the configuration of the controller. [Diagram 3]FIG. 3 is an explanatory diagram showing the control mode switching timing. [Figure 4] FIG. 4 is a vector diagram showing the case where the power factor of the power supplied from the battery to the rotating electrical machine (inverter) is 1. [Diagram 5] FIG. 5 is a vector diagram showing the change due to the change in field flux. [Figure 6] FIG. 6 is a vector diagram of a scene in which the stator winding current is limited. [Figure 7] FIG. 7 is a circuit diagram showing a configuration of an inverter. [Figure 8] FIG. 8 is a graph showing the U-phase current, the U-phase voltage, and the like when square wave control is performed in the comparative example. [Figure 9] FIG. 9 is a graph showing the U-phase current, the U-phase voltage, and the like when square wave control is performed in this embodiment. [Figure 10] FIG. 10 is a graph showing the U-phase current, the U-phase voltage, etc. in a scene in which the stator winding current is limited when square wave control is performed in this embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0011] Hereinafter, an embodiment of the present invention will be described with reference to the drawings.

[0012] [First embodiment] Fig. 1 is a block diagram showing a schematic configuration of a vehicle system 100. As shown in Fig. 1, the vehicle system 100 includes a rotating electric machine 10, a battery 11, an inverter 12, an auxiliary machine 13, and a controller .

[0013] The rotating electric machine 10 is an electric motor or a generator. When the rotating electric machine 10 is an electric motor, the vehicle system 100 is a drive source for an electric vehicle such as an electric vehicle or a hybrid vehicle. When the rotating electric machine 10 is a generator, the vehicle system 100 is a power generation system that generates power to charge the battery 11. In this embodiment, the rotating electric machine 10 is an electric motor.

[0014] In particular, the rotating electric machine 10 is a so-called wound field type AC synchronous machine. That is, the rotating electric machine 10 has a field magnetic flux Ψ generated by a field winding 15 provided on a rotor and a stator magnetic flux Φ generated by a stator winding 16. s It rotates due to the interaction of and .

[0015] In this embodiment, the rotating electric machine 10 is a three-phase synchronous machine, and the stator winding 16 is composed of a U-phase winding, a V-phase winding, and a W-phase winding. s is the magnetic flux Φ generated by the stator winding 16 of each phase. U ,Φ V ,Φ W This results in a composite magnetic flux that forms a so-called rotating magnetic field.

[0016] In the following, the current flowing through the field winding 15 is referred to as the field winding current I f The current flowing through the stator winding 16 is called the stator winding current. The stator winding current is the current in the UVW coordinate system, that is, the U-phase current I U , V phase current I V , and W-phase current I W The stator winding current is expressed as the current in the dq coordinate system that rotates together with the rotor, that is, the d-axis current I d and q-axis current I q In addition, the voltage applied to the stator winding 16 is called the stator winding voltage.

[0017] The battery 11 is a DC power source that supplies power to the rotating electric machine 10. In particular, in this embodiment, the battery 11 supplies power not only to the rotating electric machine 10 but also to the auxiliary machine 13 as necessary. In other words, the battery 11 is a power source common to the rotating electric machine 10 and the auxiliary machine 13. The DC voltage V dc can be acquired at any timing as required by a voltage sensor (not shown). The battery 11 is, for example, a lithium ion battery.

[0018] The inverter 12 converts the DC power output by the battery 11 into AC power and supplies it to the rotating electric machine 10. The inverter 12 includes a smoothing capacitor 17 (input capacitor) and a power module 18. The smoothing capacitor 17 smoothes the power supplied from the battery 11 to the inverter 12. The power module 18 includes a bridge circuit configured with a plurality of switching elements. The timing of turning on or off each switching element of the power module 18 is controlled by PWM control. In this embodiment, the power module 18 not only supplies power to the stator windings 16 of each of the UVW phases, but also includes a module that supplies a predetermined power to the field winding 15 through the line X.

[0019] The auxiliary machine 13 is one or more devices that consume electric power other than the rotating electric machine 10. In an electric vehicle, the auxiliary machine 13 is, for example, an electric compressor of an air conditioner, an on-board charger, and a DC / DC converter. As described above, the auxiliary machine 13 uses the battery 11 that supplies electric power to the rotating electric machine 10 as a power source. For this reason, the auxiliary machine 13 is connected to the battery 11 in parallel with the rotating electric machine 10 via a smoothing capacitor 19 (input capacitor). If a ripple occurs in the current input from the battery 11 to the inverter 12, this current ripple may flow into the smoothing capacitor 19 for the auxiliary machine 13, causing the smoothing capacitor 19 to generate heat unnecessarily.

[0020] The controller 14 is a control device that comprehensively controls each part of the vehicle system 100, and is configured by, for example, one or more computers or other circuits. In this embodiment, the controller 14 functions as a rotating electric machine control device that controls the rotating electric machine 10, that is, a device that controls the operation of the inverter 12 to drive the rotating electric machine 10. The controller 14 also functions as an auxiliary machine control device that controls the operation of the auxiliary machine 13. A program (rotating electric machine control program) for causing the controller 14 to function as a rotating electric machine control device and a program (auxiliary machine control program) for causing the controller 14 to function as an auxiliary machine control device are stored in advance in a storage (not shown), but may be provided in a state of being stored in a predetermined storage medium or by wireless communication or the like.

[0021] Fig. 2 is a block diagram showing the configuration of the controller 14. Fig. 2 shows the main parts of a rotating electrical machine control device, and omits other elements.

[0022] As shown in FIG. 2, the controller 14 includes a current command value calculation unit 21, a voltage command value calculation unit 22, a coordinate conversion unit 23, a PWM signal generation unit 24, a rotation detection unit 25, a coordinate conversion unit 26, and a power factor correction calculation unit 27.

[0023] The current command value calculation unit 21 calculates a torque command value τ e * , the angular velocity ω [rad / s] of the rotating electric machine 10 (rotor), and the DC voltage V dc Based on this, the d-axis current command value I d * , q-axis current command value I q * , and the field winding current command value I f * Calculate the d-axis current command value I d * is the d-axis component of the current to be passed through the stator winding 16, d The q-axis current command value I q * is the q-axis current I qIn the following, the d-axis current command value I d * and q-axis current command value I q * The dq-axis current command value I d * ,I q * The field winding current command value I f * is the field winding current I f The field magnetic flux Ψ generated by the field winding 15 is determined by the field winding current I f The torque command value τ is adjusted by setting and changing the e * is determined based on the accelerator operation amount, etc. In this embodiment, for simplicity, the torque command value τ e * is already given. The angular velocity ω is obtained from the rotation detection unit 25. The DC voltage V dc As described above, is appropriately acquired by a voltage sensor (not shown).

[0024] The voltage command value calculation unit 22 calculates the d-axis voltage command value V d * and the q-axis voltage command value V q * (Hereinafter, the dq-axis voltage command value V d * ,V q * The dq-axis voltage command value V d * ,V q * is the dq-axis current command value I d * ,I q * d-axis current I according to d and q-axis current I q (Hereinafter, dq axis current I d ,I q The voltage command value calculation unit 22 calculates the field winding voltage command value V f * The field winding voltage command value V f *The field winding current command value I f * The field winding current I according to f This is the voltage command for flowing.

[0025] In principle, the voltage command value calculation unit 22 calculates the dq-axis current I d ,I q and the field winding current I f is the dq axis current command value I d * ,I q * and the field winding current command value I f * The dq-axis voltage command value V d * ,V q * and the field winding voltage command value V f * That is, in principle, the voltage command value calculation unit 22 calculates the fed back dq-axis current I d ,I q and the field winding current I f and the dq-axis current command value I d * ,I q * and the field winding current command value I f * Based on this, the dq-axis voltage command value V d * ,V q * and the field winding voltage command value V f * Calculate the following.

[0026] In this embodiment, as will be described later, the voltage command value calculation unit 22 receives the dq-axis current I d ,I q and the field winding current I f , i.e., the corrected dq-axis current I d ^,I q ^ and corrected field winding current I fTherefore, the voltage command value calculation unit 22 of this embodiment uses the corrected dq-axis current I d ^,I q ^ and corrected field winding current I f ^ and the dq-axis current command value I d * ,I q * and the field winding current command value I f * Based on this, the dq-axis voltage command value V d * ,V q * and the field winding voltage command value V f * Calculate the following.

[0027] In addition, the voltage command value calculation unit 22 receives the dq-axis voltage command value V d * ,V q * and the field winding voltage command value V f * There are two modes for calculating the dq-axis voltage command value V d * ,V q * and the field winding voltage command value V f * In the second mode, the dq-axis voltage command value V d * ,V q * and the field winding voltage command value V f * This is a mode in which the voltage command value calculation unit 22 calculates the voltage command value. The voltage command value calculation unit 22 switches between the current vector control and the voltage phase control according to the modulation factor M (not shown) of the PWM control. The switching between the current vector control and the voltage phase control will be described in detail later.

[0028] The voltage command value calculation unit 22 calculates the dq-axis voltage command value V d * ,V q * and the field winding voltage command value V f* In this embodiment, the illustration and description of specific blocks for the decoupling control are omitted, but the dq-axis voltage command value V d * ,V q * and the field winding voltage command value V f * is decoupled.

[0029] The coordinate conversion unit 23 converts the dq-axis voltage command value V d * ,V q * Based on this, the U-phase voltage command value V U * , V-phase voltage command value V V * , and W-phase voltage command value V W * (Hereinafter, the three-phase voltage command value V U * ,V V * ,V W * The U-phase voltage command value V U * is the U-phase voltage V applied to the U-phase stator winding 16. U The V-phase voltage command value V V * is the V-phase voltage V applied to the V-phase stator winding 16. V Similarly, the W-phase voltage command value V W * is the W-phase voltage V applied to the W-phase stator winding 16. W In the following, the U-phase voltage V U , V phase voltage V V , and W-phase voltage V W Putting these together, we get the three-phase voltage V U ,V V ,V W The three-phase voltage V U ,V V ,V W is the stator winding voltage. Also, U-phase current I U, V phase current I V , and W-phase current I W Putting these together, the three-phase current I U ,I V ,I W The coordinate conversion from the dq axis coordinate system to the UVW coordinate system is performed based on the electrical angle θ [deg] of the rotating electric machine 10.

[0030] The PWM signal generator 24 generates a three-phase voltage command value V U * ,V V * ,V W * Based on this, a PWM signal that determines the timing at which the switching elements of the UVW phases are turned on / off is generated and input to the inverter 12. As a result, the stator winding 16 is supplied with a three-phase voltage command value V U * ,V V * ,V W * Three-phase voltage V according to U ,V V ,V W is applied, resulting in a three-phase current I U ,I V ,I W The actual three-phase current I U ,I V ,I W is detected by the current detector 31. U ,I V ,I W is the stator winding current.

[0031] In this embodiment, the field winding voltage command value V f * Based on this, the field winding voltage V f The inverter 12 generates a PWM signal (or other control signal) for controlling the field winding voltage command value V f * The field winding voltage V according to f is applied, resulting in a field winding current I f The actual field winding current If is detected by the current detector 32.

[0032] The PWM signal generating unit 24 has two operating modes. The first mode is a mode in which so-called asynchronous PWM control is performed. The second mode is a mode in which so-called synchronous PWM control is performed. The asynchronous PWM control is a control that uses a triangular wave or the like of a predetermined frequency as a carrier signal regardless of the rotation state of the rotating electric machine 10 (rotor). The synchronous PWM control is a control that uses a carrier signal synchronized with the rotation of the rotating electric machine 10 (rotor). The PWM signal generating unit 24 switches between asynchronous PWM control and synchronous PWM control depending on the modulation factor M. The switching between asynchronous PWM control and synchronous PWM control will be described in detail later.

[0033] The rotation detection unit 25 calculates parameters representing the rotation state of the rotating electric machine 10 (rotor) based on a signal output by a rotation detector 33 such as a resolver or an encoder. Specifically, the rotation detection unit 25 calculates an electrical angle θ, an angular velocity ω, a rotation speed N [rpm], and the like.

[0034] The coordinate conversion unit 26 converts the detected three-phase current I U ,I V ,I W Based on this, the dq axis current I d ,I q Calculate the dq axis current I d ,I q is the stator winding current. Note that the coordinate conversion from the UVW coordinate system to the dq axis coordinate system is performed based on the electrical angle θ of the rotating electric machine 10.

[0035] The power factor correction calculation unit 27 corrects the detected dq-axis current I so that the power factor of the electric power supplied from the battery 11 to the rotating electric machine 10 (inverter 12) (hereinafter simply referred to as the power factor) approaches 1 (maximum value) as much as possible. d ,I q (stator winding current) and field winding current I f Then, the power factor correction calculation unit 27 corrects the corrected dq-axis current I d ,I qThe corrected dq-axis current I d ^,I q ^(corrected stator winding current) and corrected field winding current I f The corrective field winding current I f ^ is fed back to the voltage command value calculation unit 22.

[0036] In this way, the corrected dq axis current I d ^,I q ^ and corrected field winding current I f By feeding back the voltage command value calculation unit 22, the power factor correction calculation unit 27 changes the field flux Ψ. As a result, the power factor correction calculation unit 27 changes the stator winding current (I U ,I V ,I W ) becomes zero, and the stator winding voltage (V U ,V V ,V W ) becomes zero, and

[0037] The power factor correction calculation unit 27 calculates, for example, the detected dq-axis current I d ,I q and the field winding current I f Based on this, the power factor and the torque τ e Estimate the estimated power factor and torque τ e Based on this, the corrected dq axis current I d ^,I q ^ and corrected field winding current I f In this embodiment, for simplicity, the power factor correction calculation unit 27 calculates (sets) the dq-axis current I d ,I q and the field winding current I f and the corrected dq axis current I d ^,I q ^ and corrected field winding current I f Therefore, the power factor correction calculation unit 27 refers to the correction map to calculate the detected dq-axis current I d ,Iq and the field winding current I f According to the correction dq axis current I d ^,I q ^ and corrected field winding current I f Set the power factor correction, i.e., the correction dq axis current I d ^,I q ^ and corrected field winding current I f The calculation of ^ will be described in detail later.

[0038] <Switching control modes> 3 is an explanatory diagram showing the control mode switching timing. The horizontal axis of FIG. 3 is the rotation speed N, and the vertical axis is the torque τ e The solid line shows the torque τ that can be output according to the rotation speed N. e The maximum torque that the rotating electrical machine 10 can output over the entire rotation speed N is τ e-MAX And the maximum torque τ e-MAX The limit speed at which the rotation can be maintained is N lim It is.

[0039] Line L1 is the boundary where the modulation factor M of the PWM control is 1.00. In the control region where the rotation speed N is lower than line L1, the modulation factor M is 1.00 or less and the three-phase current I U ,I V ,I W In the normal control region (hereinafter referred to as normal region 36) where the rotation speed N is higher than that of the line L1, the modulation factor M exceeds 1.00 and the three-phase current I U ,I V ,I W This is a region (hereinafter, referred to as an overmodulation region 37) where the waveform of the signal L1 is controlled in a partially limited overmodulation state. That is, the line L1 is an overmodulation arrival line.

[0040] Line L2 is a boundary line where the modulation factor M of PWM control becomes 1.05. Line L2 is set to provide hysteresis to the switching of the control mode and to suppress frequent switching of the control mode.

[0041] Line L3 is the boundary line where the modulation factor M of the PWM control is 1.10. In the control region where the rotation speed N is higher than line L3, the modulation factor M exceeds 1.10 and the three-phase current I U ,I V ,I W This is a control region (hereinafter referred to as a square wave region 38) in which the waveform of the square wave is substantially a square wave. That is, the line L3 is a square wave arrival line. Control in the square wave region 38 (square wave control) is sometimes called one-pulse control.

[0042] In principle, in the normal region 36, the rotating electric machine 10 is controlled by current vector control and asynchronous PWM control, and in the overmodulation region 37, the rotating electric machine 10 is controlled by voltage phase control and synchronous PWM control. That is, in principle, when the rotating electric machine 10 is controlled in the normal region 36, the voltage command value calculation unit 22 selects current vector control, and the PWM signal generation unit 24 selects asynchronous PWM control. When the rotating electric machine 10 is controlled in the overmodulation region 37, the voltage command value calculation unit 22 selects voltage phase control, and the PWM signal generation unit 24 selects synchronous PWM control.

[0043] In this embodiment, when the control of the rotating electric machine 10 transitions from control in the normal region 36 to control in the overmodulation region 37, the control mode is switched from voltage vector control and asynchronous PWM control to voltage phase control and synchronous PWM control when the line L2 is crossed. On the other hand, when the control of the rotating electric machine 10 transitions from control in the overmodulation region 37 to control in the normal region 36, the control mode is switched from voltage phase control and synchronous PWM control to current vector control and asynchronous PWM control when the line L1 is crossed.

[0044] <Power factor correction> FIG. 4 is a vector diagram when the power factor of the electric power supplied from the battery 11 to the rotating electrical machine 10 (inverter 12) is 1.

[0045] As shown in Figure 4, in the dq-axis coordinate system, the stator winding current is expressed by the current vector I a The current vector Ia is the d-axis current vector (I d ) and the q-axis current vector (I q ) is the composite vector of the stator winding current. The phase β of the stator winding current is the current vector I a is expressed by the angle it makes with the q axis.

[0046] The field flux Ψ is represented by a positive vector on the d-axis. The magnetic flux vector Φ0 of the rotating electric machine 10 is the sum of the field flux Ψ and the magnetic flux generated by the stator winding 16 (hereinafter, the stator magnetic flux Φ s The stator flux Φ s is the magnetic flux generated in the q-axis direction (L q I q ) and the magnetic flux generated in the d-axis direction (L d I d ) is the combination of the magnetic flux (L q I q ) is the q-axis inductance L of the stator winding 16 q and the q-axis current I q Similarly, the magnetic flux generated in the d-axis direction (L d I d ) is the d-axis inductance L of the stator winding 16 d and d-axis current I d And the rotating magnetic flux is the stator magnetic flux Φ s The phase δ of the stator flux Φ s It is represented by the angle between the vector representing and the q axis.

[0047] Current vector I a and the stator flux Φ s The vectors representing the stator winding current phase β and the stator flux Φ are parallel. s When the phases δ of the current vector I a and the magnetic flux vector Φ0 becomes perpendicular.

[0048] In the control in the normal region 36, the torque command value τ e * The torque τ corresponding to eIn order to output the required torque, the dq-axis current I d ,I q In other words, the phase β of the stator winding current can be selected substantially arbitrarily. Therefore, in the control in the normal region 36, the combination of the phase β of the stator winding current and the stator magnetic flux Φ s The dq-axis current I d ,I q is set (calculated).

[0049] On the other hand, in the control in the overmodulation region 37 (particularly, the rectangular wave region 38), the stator winding current and the stator winding voltage are limited due to the performance limitations of the inverter 12, etc., and the dq-axis current I d ,I q In other words, the degree of freedom in selecting the phase β is reduced. For this reason, for example, when the rotating electric machine to be controlled is a permanent magnet embedded type and the field flux Ψ cannot be controlled, in the overmodulation region 37 and the square wave region 38, the dq-axis current I is usually adjusted so that the phase β and the phase δ are equal. d ,I q As a result, a drop in the power factor is unavoidable.

[0050] However, in this embodiment, since the rotating electric machine 10 is a wound field type, the field magnetic flux Ψ is equal to the field winding current I f Therefore, in this embodiment, the dq-axis current I d ,I q Together with this, the field winding current I f By adjusting the field flux Ψ, the dq-axis current I d ,I q Therefore, in this embodiment, even if the phase β is limited, the power factor is maintained at 1 by changing the field flux Ψ. Specifically, as described above, the power factor correction calculation unit 27 sets the dq-axis current I d ,I q and the field winding current I f This is achieved by correcting the dq-axis current Id ^,I q ^ and corrected field winding current I f ^ is set to modify the field flux Ψ to maintain unity power factor.

[0051] The change in the field flux Ψ will be described in detail below.

[0052] Torque τ of the rotating electric machine 10 e is generally expressed by the following formula (1). "P" is the number of pole pairs of the rotating electrical machine 10. "L d " is the d-axis inductance [H] of the stator winding 16, as described above. Similarly, "L q " is the d-axis inductance [H] of the stator winding 16.

[0053]

number

[0054] However, since the rotating electric machine 10 of the present embodiment is a wound field type, the reluctance torque is very small compared to the torque (so-called magnet torque) generated by the field magnetic flux Ψ. For this reason, in the formula (1), the second term (P(L d -L q )I d I q ) can be omitted. Therefore, the torque τ e is equal to the torque generated by the field flux Ψ, as shown in the following equation (2). Also, the field flux Ψ is equal to the torque generated by the field winding current I f The proportionality coefficient k is predetermined by the specific configuration of the field winding 15.

[0055]

number

[0056] Therefore, as is clear from the formulas (2) and (3), the torque τ of the rotating electric machine 10, which is a wound field type, e is expressed by the following equation (4).

[0057]

number

[0058] Moreover, the voltage equation of the rotating electric machine 10 is generally expressed by the following equation (5). d " is the d-axis voltage, and "V q " is the q-axis voltage. These dq-axis voltages V d ,V q is the stator winding voltage. "R a " is the resistance [Ω] of the stator winding 16. "s" is the differential operator (d / dt). Also, "ω" is the angular velocity [rad / s] of the rotating electric machine 10, as described above.

[0059]

number

[0060] When phase β and phase δ are equal and the power factor is 1, the dq axis current I d ,I q Ratio of and dq-axis voltage V d ,V q The ratio of is equal. That is, I d / I q =V d / V q Therefore, using equation (3) for the field flux Ψ, I d / I q =V d / V q If we solve the voltage equation (5) under these conditions, the d-axis current I d is expressed by the following equation (6). Also, the q-axis current I q is expressed by the following equation (7).

[0061]

number

[0062] Torque command value τ e *The torque τ that the rotating electrical machine 10 should output is e Once this is determined, the q-axis current I q and the field winding current I f When the required torque is determined, the q-axis current I q and the field winding current I f are roughly inversely proportional to the field winding current I f If you increase the q-axis current I q decreases and the field winding current I f If you reduce the q-axis current I q Also, the d-axis current I d As shown in equation (6), the q-axis current I q and the field winding current I f For example, when the field winding current I f If you increase the d-axis current I d The magnitude of the field winding current I f If we reduce the d-axis current I d The magnitude of decreases.

[0063] FIG. 5 is a vector diagram showing the change caused by changing the field flux Ψ. For the sake of explanation, as shown in FIG. 5(A), a specific dq-axis current I d ,I q and the field winding current I f The influence of the change in the field flux Ψ will be described with reference to a scene in which the rotating electric machine 10 is controlled by the torque τ e shall be maintained.

[0064] As shown in Fig. 5(B), the field winding current I f I f - When the field flux Ψ is reduced to Ψ - At this time, the q-axis current I q I q + and the d-axis current I d I d - Therefore, the stator flux Φ s is Φ s' and its phase δ changes to δ - Note that the magnetic flux vector Φ0 changes to Φ0'.

[0065] As shown in Fig. 5(C), the field winding current I f I f + When the field flux Ψ is increased to Ψ + At this time, the q-axis current I q I q - and the d-axis current I d I d + Therefore, the stator flux Φ s is Φ s " and its phase δ is δ + Note that the magnetic flux vector Φ0 changes to Φ0″.

[0066] In this way, in the wound field type rotating electric machine 10, the field winding current I f By adjusting the field flux Ψ, the torque τ e (required torque) while maintaining the stator flux Φ s The phase δ of the dq-axis current I d ,I q changes, but according to the change in field flux Ψ, the dq axis current I d ,I q The change in the current vector I can be kept within the range of the stator winding current and the stator winding voltage due to the performance limit of the inverter 12, etc. a Phase β and stator flux Φ s This makes it easier to bring the phase δ of the inverter closer together and maintain the power factor at or near 1.

[0067] FIG. 6 is a vector diagram of a scene in which the stator winding current is limited. In FIG. 6, the maximum value of the stator winding current (maximum current I am The current limit ellipse representing the maximum induced voltage V according to the rotation speed N is shown by a dashed line. amThe voltage limit ellipse representing the maximum torque τ that can be output at each rotation speed N of 3000 rpm, 4000 rpm, 5000 rpm, and 6000 rpm under the limitation of the stator winding current is shown by the dotted line. e When outputting the current vector I a I a1 ,I a2 ,I a3 ,I a4 In addition, these current vectors I a1 ,I a2 ,I a3 ,I a4 For the current vector I, the field magnetic flux Ψ for maintaining the power factor at 1 (β=δ) is Ψ1, Ψ2, Ψ3, and Ψ4. However, Ψ4 is not shown. a1 ,I a2 ,I a3 ,I a4 and the field stator flux Φ corresponding to the field fluxes Ψ1, Ψ2, Ψ3, and Ψ4 s Φ s1 , Φ s2 , Φ s3 , Φ s4 Similarly, the current vector I a1 ,I a2 ,I a3 ,I a4 And the magnetic flux vector Φ0 corresponding to the field magnetic fluxes Ψ1, Ψ2, Ψ3, and Ψ4 is Φ 01 , Φ 02 , Φ 03 , Φ 04 In addition, the current vector I a1 ,I a2 ,I a3 ,I a4 and each phase δ of the field fluxes Ψ1, Ψ2, Ψ3, and Ψ4 are omitted in the drawing.

[0068] As shown in Figure 6, when there is a limit to the stator winding current, the current vector I a1 ,I a2 ,I a3 ,I a4 As shown in the figure, the phase β increases according to the rotation speed N. Therefore, when the rotation speed N increases while the stator winding current is limited, the power factor can be maintained at 1 by increasing the field flux Ψ.

[0069] In addition, the field flux Ψ (field current I f ) is limited. If the stator winding current is limited, for example, the current vector I a3 Even if it is possible to realize the field flux Ψ3 corresponding to the current vector I a4 In such a case, the power factor correction calculation unit 27 changes the field flux Ψ to adjust the corrected dq-axis current I d ^,I q ^ and corrected field winding current I f For example, set the current vector I a4 When a field flux Ψ4 corresponding to the above is necessary, but the field flux Ψ4 is not realizable and only the field flux Ψ3 can be realized, the power factor correction calculation unit 27 changes the field flux Ψ to the field flux Ψ3 and adjusts the corrected dq axis current I d ^,I q ^ and corrected field winding current I f Set ^.

[0070] Fig. 7 is a circuit diagram showing the configuration of inverter 12. Fig. 7 shows only the configuration for passing stator winding current, and omits illustration of other elements.

[0071] As shown in FIG. 7, the power module 18 of the inverter 12 includes legs L corresponding to the UVW phases. U ,L V ,L W Each phase leg L U ,L V ,L W The inverter 12 (rotating electric machine 10) includes an upper arm and a lower arm, and the stator windings of each phase are connected between them. The upper arm and the lower arm are each configured with a switching element and a free wheel diode. In this embodiment, the DC current supplied from the battery 11 to the inverter 12 (rotating electric machine 10) is dc The currents flowing through the upper arms of the UVW phases are I1, I2, and I3, respectively. Also, the currents flowing through the smoothing capacitor 17 and the legs L of each phase are U ,L V,L W The DC current flowing between these terminals is I4.

[0072] Hereinafter, the operation of the controller 14 (rotating electric machine control device) configured as above will be described.

[0073] FIG. 8 shows the U-phase current I U and U phase voltage V U 8 is a graph showing the relationship between the power factor and the field flux Ψ in the comparative example, which is a control example of a rotating electric machine with embedded permanent magnets, in which the power factor is not corrected by changing the field flux Ψ. The other control configurations are the same as those of the present embodiment.

[0074] FIG. 8A shows the U-phase current I U and U phase voltage V U In FIG. 8(A), the U-phase voltage V U The zero voltage timing is indicated by a ◇, and the U-phase current I U The zero current timing is indicated by a ▽ mark. In FIG. 8(A), the U-phase current I U and U phase voltage V U FIG. 8(B) shows currents I1, I2, I3, and I4 flowing through various parts of the inverter 12 in the comparative example. In FIG. 8(B), the currents I1, I2, I3, and I4 are indicated by a thin solid line, a dashed line, a dashed dotted line, and a thick solid line, respectively. FIG. 8(C) shows DC current I supplied from the battery 11 to the inverter 12 (rotating electric machine 10) in the comparative example. dc The horizontal axis in each graph in FIG. 8 represents time.

[0075] As shown in FIG. 8A, the U-phase stator winding 16 outputs a rectangular-wave U-phase voltage V U is applied, and a sinusoidal U-phase current I U flows. At this time, in the comparative example, the zero voltage timing (▽) and the zero current timing (◇) are shifted. The difference Δ between the zero voltage timing (▽) and the zero current timing (◇) becomes noticeable in the control in the overmodulation region 37, and particularly in the control in the square wave region 38. In addition, the power factor drops to about 0.9.

[0076] At this time, U phase leg L U The switching element of U phase voltage V U As a result, as shown in FIG. 8B, the U-phase leg L U The current I1 flowing through the upper arm of the U-phase voltage V U When the V-phase leg L changes from negative to positive, the current I1 temporarily becomes negative. V and W-phase leg L W The currents I2 and I3 flowing through the upper arms of the MOSFETs 11 and 12 also become negative temporarily, just like the current I1. U ,L V ,L W The current I4 flowing between these two terminals is the sum of the currents I1, I2, and I3. Therefore, as shown in FIG. 8B, when the currents I1, I2, and I3 temporarily become negative, the current I4 drops at that timing.

[0077] In other words, if there is a difference Δ between the zero voltage timing (▽) and the zero current timing (◇), each phase leg L U ,L V ,L W The currents I1, I2, and I3 flowing through the upper arm of the inverter temporarily become negative, causing a ripple in the current I4. The effective value of the current I4 shown in FIG. 8B is about 550A, and the amplitude of the ripple (the difference in the current value between the peaks) is about 300A.

[0078] Although the ripple of the current I4 is smoothed by the smoothing capacitor 17, as shown in FIG. 8C, the DC current I dc A large ripple also occurs in the DC current I dc The effective value of is about 540 A, and the amplitude of the ripple is about 180 A. As described above, the battery 11 is also connected to the auxiliary device 13, so that the DC current I dc The ripple generated in the smoothing capacitor 19 flows into the smoothing capacitor 19 that connects the auxiliary device 13, causing the smoothing capacitor 19 to generate heat.

[0079] FIG. 9 shows the U-phase current I U and U phase voltage V U FIG. 9A is a graph showing the U-phase current I U and U phase voltage V U Shows.

[0080] In Figure 9(A), the U-phase voltage V U and U-phase current I U The timing when the U-phase current I U and U phase voltage V U FIG. 9(B) shows currents I1, I2, I3, and I4 flowing through various parts of the inverter 12 in this embodiment. In FIG. 9(B), the currents I1, I2, I3, and I4 are indicated by a thin solid line, a dashed line, a dashed dotted line, and a thick solid line, respectively. FIG. 9(C) shows DC current I supplied from the battery 11 to the inverter 12 (rotating electric machine 10) in this embodiment. dc The horizontal axis in each graph in FIG. 9 represents time.

[0081] As shown in FIG. 9A, in this embodiment, the power factor is corrected by changing the field flux Ψ, so that the U-phase voltage V U and U-phase current I U That is, as indicated by the circles, the zero voltage timing and the zero current timing coincide with each other, and the power factor becomes approximately 1.

[0082] For this reason, as shown in FIG. 9(B), the U-phase leg L U The current I1 flowing through the upper arm of the V-phase leg L is always greater than or equal to zero (effectively only positive current components). V and W-phase leg L W The currents I2 and I3 flowing through the upper arms of the inverter 11 are always equal to or greater than zero, just like the current I1. U ,L V ,L WThe ripple of the current I4 flowing between is reduced compared to the comparative example. In particular, in FIG. 9(B), the zero voltage timing and the zero current timing are perfectly matched, so the ripple of the current I4 is minimized. The effective value of the current I4 shown in FIG. 9(B) is about 600 A, and the amplitude of the ripple is about 100 A.

[0083] Since the ripple of the current I4 is suppressed, the DC current I supplied from the battery 11 to the inverter 12 is reduced as shown in FIG. dc The ripple of the DC current I shown in FIG. 9C is also reduced compared to the comparative example. As a result, the ripple flowing into the smoothing capacitor 19 connecting the auxiliary device 13 is also reduced, and the heat generation of the smoothing capacitor 19 is suppressed. dc The effective value of is about 600A and the amplitude of the ripple is about 60A.

[0084] FIG. 10 shows a U-phase current I in a scene where the stator winding current is limited when performing square wave control in this embodiment. U and U phase voltage V U Graph showing the above.

[0085] Figure 10(A) shows the U-phase current I U and U phase voltage V U In FIG. 10(A), the U-phase voltage V U The timing when becomes zero is indicated by ◇, and the U-phase current I U The timing when the U-phase current I U and U phase voltage V U FIG. 10(B) shows currents I1, I2, I3, and I4 flowing through various parts of the inverter 12. In FIG. 10(B), the currents I1, I2, I3, and I4 are indicated by a thin solid line, a dashed line, a dashed dotted line, and a thick solid line, respectively. FIG. 10(C) shows the DC current I supplied from the battery 11 to the inverter 12 (rotating electric machine 10) in a comparative example. dc The horizontal axis in each graph in FIG. 10 represents time.

[0086] As shown in FIG. 10A, in a scene where the stator winding current is limited, it may not be possible to completely match the zero voltage timing (▽) and the zero current timing (◇) by changing the field flux Ψ and maintain the power factor at 1. However, the power factor correction calculation unit 27 changes the field flux Ψ within a feasible range and adjusts the corrected dq axis current I d ^,I q ^ and corrected field winding current I f ^ is set. That is, the power factor correction calculation unit 27 changes the field flux Ψ so as to minimize the difference Δ between the zero voltage timing (▽) and the zero current timing (◇). Therefore, the difference Δ between the zero voltage timing (▽) and the zero current timing (◇) is reduced more than in the comparative example, and the power factor becomes about 0.95.

[0087] For this reason, as shown in FIG. 10(B), the U-phase leg L U Although the current I1 flowing through the upper arm of the V-phase leg L temporarily becomes negative, the magnitude of the negative current is reduced compared to the comparative example. V and W-phase leg L W Although the currents I2 and I3 flowing through the upper arms of the inverter 11 and 12 also temporarily become negative like the current I1, the magnitude of the negative current is reduced compared to the comparative example. U ,L V ,L W Although the current I4 flowing between the currents I1, I2, and I3 drops when the currents I1, I2, and I3 temporarily become negative, the degree of this drop is suppressed compared to the comparative example. In other words, the amplitude of the ripple generated in the current I4 is suppressed compared to the comparative example. The effective value of the current I4 shown in FIG. 10(B) is about 580A, and the amplitude of the ripple is about 150A.

[0088] In this way, the ripple of the current I4 is suppressed, so that the DC current I supplied from the battery 11 to the inverter 12 is reduced as shown in FIG. dcThe ripple of the DC current I shown in FIG. 10C is also reduced compared to the comparative example. As a result, the ripple flowing into the smoothing capacitor 19 connecting the auxiliary device 13 is also reduced, and the heat generation of the smoothing capacitor 19 is suppressed. dc The effective value of is about 580A and the amplitude of the ripple is about 120A.

[0089] As described above, according to the present embodiment, the power factor is corrected to be as close to 1 as possible by changing the field magnetic flux Ψ, so that the DC current I dc As a result, the ripple flowing into smoothing capacitor 19 connecting auxiliary device 13 is reduced, and heat generation in smoothing capacitor 19 is suppressed.

[0090] In particular, in this embodiment, when a winding-field type rotating electric machine 10 is driven by one inverter 12, a DC current I dc In other words, if the power factor is corrected by changing the field magnetic flux Ψ as in this embodiment, the DC current I dc Therefore, the heat generation of the smoothing capacitor 19 can be suppressed.

[0091] In the above embodiment, the power factor is corrected by changing the field flux Ψ in the control in the square wave region 38, but this is not limiting. In the control in the overmodulation region 37, the difference Δ between the zero voltage timing (▽) and the zero current timing (◇) becomes particularly noticeable, so it is preferable to correct the power factor by changing the field flux Ψ at least in the control in the overmodulation region 37. Furthermore, in the control in the normal region 36, the difference Δ between the zero voltage timing (▽) and the zero current timing (◇) is relatively inconspicuous, but in the control in the normal region 36, if the power factor is corrected by changing the field flux Ψ as in the control in the overmodulation region 37 and the square wave region 38, the DC current I dc Therefore, the heat generation of the smoothing capacitor 19 can be suppressed.

[0092] As described above, the rotating electric machine control method according to this embodiment is a rotating electric machine control method for controlling the rotating electric machine 10 in a system (100) having the wound field type rotating electric machine 10 that generates the field magnetic flux Ψ by the field winding 15, the auxiliary machine 13, and the battery 11 that supplies power to the rotating electric machine 10 and the auxiliary machine 13. In this rotating electric machine control method, the field winding current I f and the stator winding current (I U ,I V ,I W ) and the detected field winding current I f and the stator winding current (I U ,I V ,I W ) based on the field flux Ψ, the stator winding voltage (V U ,V V ,V W ) becomes zero, and the stator winding current (I U ,I V ,I W ) becomes zero.

[0093] In this way, by changing the field flux Ψ to bring the zero voltage timing and the zero current timing closer to each other, even when the rotating electric machine 10 is driven by one inverter 12, the DC current I dc As a result, the ripple flowing into the smoothing capacitor 19 connecting the auxiliary device 13 is reduced, and the heat generation of the smoothing capacitor 19 is suppressed. When two inverters are used to control the rotating electric machine 10, the stator winding current (I U ,I V ,I W ) may not be an ideal sine wave, but this embodiment does not have such a problem.

[0094] In the rotating electrical machine control method according to the above embodiment, the zero voltage timing and the zero current timing are made to coincide with each other in principle.

[0095] In this way, when the zero voltage timing and the zero current timing are matched, the power factor is maintained at 1 and the DC current I supplied from the battery 11 to the inverter 12 is dc Therefore, the ripple flowing into the smoothing capacitor 19 to which the auxiliary device 13 is connected is also minimized, and the heat generation of the smoothing capacitor 19 is minimized.

[0096] In the rotating electric machine control method according to the above embodiment, the stator winding current (I a ) is limited and the zero voltage timing and the zero current timing cannot be made to coincide, the field flux Ψ is changed so that the difference Δ between the zero voltage timing and the zero current timing is minimized.

[0097] Thus, the stator winding current (I a ) is limited and it is not possible to match the zero voltage timing and the zero current timing, if the field flux Ψ is changed so that the difference Δ between them is minimized within a feasible range, the DC current I supplied from the battery 11 to the inverter 12 will be longer than if the field flux Ψ is not changed at all. dc Therefore, the ripple flowing into the smoothing capacitor 19 to which the auxiliary device 13 is connected can be suppressed, and the heat generation of the smoothing capacitor 19 can be reduced.

[0098] In the rotating electric machine control method according to the above embodiment, the rotating electric machine 10 is controlled by pulse width modulation control (PWM control), and the field flux Ψ is changed at least when the modulation factor M in the pulse width modulation control is greater than 1.

[0099] In the overmodulation region 37 (particularly the square wave region 38) where the modulation factor M is greater than 1, the difference Δ between the zero voltage timing and the zero current timing becomes significant, and the DC current I supplied from the battery 11 to the inverter 12 dcTherefore, as described above, it is preferable to change (adjust) the field flux Ψ and reduce heat generation in the smoothing capacitor 19 at least in the control in the overmodulation region 37 (particularly the square wave region 38).

[0100] In the rotating electric machine control method according to the above embodiment, the detected field winding current I f and the stator winding current (I U ,I V ,I W ) so that the power factor of the electric power supplied from the battery 11 to the rotating electric machine 10 approaches 1.

[0101] In this way, if the field flux Ψ is changed so that the power factor of the electric power supplied from the battery 11 to the rotating electric machine 10 approaches 1, the zero voltage timing and the zero current timing can be easily brought closer to each other. dc As a result, the ripple flowing into smoothing capacitor 19 connecting auxiliary device 13 is reduced, and heat generation in smoothing capacitor 19 is suppressed.

[0102] In the rotating electric machine control method according to the above embodiment, specifically, the required torque (τ e * ) based on the field winding current I f The field winding current command value I f * and the stator winding current (I U ,I V ,I W ) is determined by the stator winding current command value (I d * ,I q * ) and calculate the detected field winding current I based on the field flux Ψ when the power factor is 1. f and the stator winding current (I d ,I q ) to obtain the corrected field winding current I f ^ and corrected stator winding current (I d ^,I q^) and corrected field winding current I f ^ and corrected stator winding current (I d ^,I q ^) is the field winding current command value I f * and the stator winding current command value (I d * ,I q * ) to provide feedback.

[0103] In this way, the detected field winding current I f and the stator winding current (I d ,I q ) to obtain the corrected field winding current I f ^ and corrected stator winding current (I d ^,I q By calculating the zero voltage timing and the zero current timing and feeding it back, the zero voltage timing and the zero current timing can be easily brought closer to each other. dc As a result, the ripple flowing into the smoothing capacitor 19 connecting the auxiliary device 13 is reduced, and the heat generation of the smoothing capacitor 19 is suppressed.

[0104] The rotating electric machine control device according to the above embodiment is a rotating electric machine control device (controller 14) that controls the rotating electric machine 10 in a system (100) that includes a wound field type rotating electric machine 10 that generates a field magnetic flux Ψ by a field winding 15, an auxiliary machine 13, and a battery 11 that supplies power to the rotating electric machine 10 and the auxiliary machine 13. This rotating electric machine control device controls a field winding current I f and the stator winding current (I U ,I V ,I W ) and the detected field winding current I f and the stator winding current (I U ,I V ,I W) based on the field flux Ψ, the stator winding voltage (V U ,V V ,V W ) becomes zero, and the stator winding current (I U ,I V ,I W ) becomes zero.

[0105] In this way, by changing the field flux Ψ to bring the zero voltage timing and the zero current timing closer to each other, even when the rotating electric machine 10 is driven by one inverter 12, the DC current I dc As a result, the ripple flowing into smoothing capacitor 19 connecting auxiliary device 13 is reduced, and heat generation in smoothing capacitor 19 is suppressed.

[0106] Although the embodiments of the present invention have been described above, the configurations described in the above embodiments merely show some of the application examples of the present invention, and are not intended to limit the technical scope of the present invention. For example, in the above embodiments and modified examples, the vehicle system 100 is exemplified as a system equipped with a winding-field type rotating electric machine 10, but the present invention is also suitable for systems other than vehicles as long as the system is equipped with a winding-field type rotating electric machine 10. [Explanation of symbols]

[0107] 10: rotating electric machine, 11: battery, 12: inverter, 13: auxiliary machine, 14: controller, 15: field winding, 16: stator winding, 17: smoothing capacitor, 18: power module, 19: smoothing capacitor, 21: current command value calculation unit, 22: voltage command value calculation unit, 23: coordinate conversion unit, 24: PWM signal generation unit, 25: rotation detection unit, 26: coordinate conversion unit, 27: power factor correction calculation unit, 31: current detector, 32: current detector, 33: rotation detector, 36: normal region, 37: overmodulation region, 38: square wave region, 100: vehicle system

Claims

1. A rotating electric machine control method for a system including a rotating electric machine of a wound field type that generates a field magnetic flux by a field winding, an auxiliary machine, and a battery that supplies power to the rotating electric machine and the auxiliary machine, comprising: Detecting a field winding current that is a current flowing through the field winding and a stator winding current that is a current flowing through a stator winding of the rotating electric machine; changing the field flux based on the detected field winding current and stator winding current, thereby bringing closer a zero voltage timing, which is a timing at which a stator winding voltage, which is a voltage applied to the stator winding, becomes zero, and a zero current timing, which is a timing at which the stator winding current becomes zero, A rotating electrical machine control method.

2. 2. A rotating electric machine control method according to claim 1, comprising: The zero voltage timing and the zero current timing are made to coincide with each other. A rotating electrical machine control method.

3. 3. A rotating electrical machine control method according to claim 2, comprising: if the stator winding current is limited and the zero current timing and the zero voltage timing cannot be made to coincide, modifying the field flux so that the difference between the zero voltage timing and the zero current timing is minimized. A rotating electrical machine control method.

4. A rotating electric machine control method according to any one of claims 1 to 3, The rotating electric machine is controlled by pulse width modulation control; changing the field flux at least when a modulation rate in the pulse width modulation control is greater than 1; A rotating electrical machine control method.

5. A rotating electric machine control method according to any one of claims 1 to 3, changing the field flux based on the detected field winding current and the detected stator winding current so that a power factor of the electric power supplied from the battery to the rotating electric machine approaches 1; A rotating electrical machine control method.

6. A rotating electrical machine control method according to claim 5, Calculating a field winding current command value that determines the field winding current and a stator winding current command value that determines the stator winding current based on a required torque; calculating a corrected field winding current and a corrected stator winding current by correcting the detected field winding current and the detected stator winding current based on the field flux when the power factor becomes 1; The corrective field winding current and the corrective stator winding current are fed back to the field winding current command value and the stator winding current command value. A rotating electrical machine control method.

7. A rotating electric machine control device for controlling a rotating electric machine in a system including a wound field type rotating electric machine that generates a field magnetic flux by a field winding, an auxiliary machine, and a battery that supplies power to the rotating electric machine and the auxiliary machine, comprising: Detecting a field winding current that is a current flowing through the field winding and a stator winding current that is a current flowing through a stator winding of the rotating electric machine; changing the field flux based on the detected field winding current and stator winding current, thereby bringing closer a zero voltage timing, which is a timing at which a stator winding voltage, which is a voltage applied to the stator winding, becomes zero, and a zero current timing, which is a timing at which the stator winding current becomes zero, Rotating electric motor control device.

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