DC-DC converter

Abstract

To provide a converter capable of reducing switching loss by performing ZVS operation in simple control, even in initial charging.SOLUTION: A DC-DC converter 1 is configured so that a first full-bridge circuit 10 and a second full-bridge circuit 20 are connected through a transformer T and an inductor L1. A control circuit 30 performs soft switching control of each switching element in the first full-bridge circuit 10 and the second full-bridge circuit 20. The control circuit 30 is switchable between an initial charging operation which gradually increases an output voltage value to a target voltage value and a normal operation which stably outputs a desired target voltage value. The control circuit 30 executes an output angle modulation mode which adjusts a voltage output period for each of the first full-bridge circuit 10 and the second full-bridge circuit 20 regardless of the value of the output voltage value in the initial charging operation, and selects one control mode from a plurality of control modes including the output angle modulation mode according to the value of output power and executes it in the normal operation.SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present invention relates to a DC-DC converter.

Background Art

[0002] In a power conversion device such as a DC-DC converter, zero voltage switching (hereinafter referred to as ZVS) is used to reduce switching loss and perform power transmission with high efficiency, and also to reduce noise, suppress switching surges, and use inexpensive elements with low withstand voltage.

[0003] Patent Document 1 discloses a DC-DC converter that enables ZVS operation and high-efficiency power transmission when the voltage difference between the primary-side DC voltage and the secondary-side DC voltage is large. In the DC-DC converter described in Patent Document 1, power is detected on each of the primary side and the secondary side, and the duty of the primary-side switch and the duty of the secondary-side switch are increased or decreased so that the difference between these two powers becomes minimum. Thereby, ZVS operation is achieved.

Prior Art Documents

Patent Documents

[0004]

Patent Document 1

Patent Document 2

Summary of the Invention

Problems to be Solved by the Invention

[0005] In Patent Document 1, in order to perform ZVS operation, it is necessary to detect power and perform switching control on each of the primary side and the secondary side. Therefore, the circuit configuration and its control become complicated, and it is difficult to improve productivity and reduce costs.

[0006] In contrast, Patent Document 2 discloses a method of performing ZVS operation with simple control and reducing switching loss by executing a plurality of modes, such as an output angle modulation mode for adjusting the voltage output period of the full-bridge circuits 10 and 20 on the primary side and the secondary side respectively, a frequency modulation mode for adjusting the switching frequency, and a phase control mode for changing the phase difference of the AC voltages of the full-bridge circuits 10 and 20 on the primary side and the secondary side, according to the target power to be output.

[0007] In this method, the mode is selected according to the target power. Therefore, in the initial charging, since the charging destination has a low voltage, the output power value may change significantly from a small value to a large value, and thus the mode may be changed midway. In that case, magnetic bias of the transformer due to the mode change occurs, the transformer current does not reach the desired value, ZVS cannot be realized, and there is a possibility of element breakdown or an error in the detection system due to noise.

[0008] Therefore, an object of the present invention is to provide a DC-DC converter that performs ZVS operation even in initial charging and reduces switching loss.

Means for Solving the Problem

[0009] To solve the above problems, the first invention of the present application is a DC-DC converter, comprising: a first full-bridge circuit having four switching elements, including a capacitor which is a parasitic capacitance or an externally connected capacitor connected in parallel; a second full-bridge circuit having four switching elements, including a capacitor which is a parasitic capacitance or an externally connected capacitor connected in parallel; a first winding connected to the first full-bridge circuit; a transformer having a second winding connected to the second full-bridge circuit and magnetically coupled to the first winding; an inductance component connected in series to the first winding or the second winding; and a control circuit for soft-switching control of each switching element of the first full-bridge circuit and the second full-bridge circuit. The control circuit is switchable between an initial charging operation of gradually increasing an output voltage value to a target voltage value and a normal operation of stably outputting power at the predetermined target voltage value. In the initial charging operation, the control circuit only executes an output angle modulation mode for adjusting a voltage output period of each of the first full-bridge circuit and the second full-bridge circuit. In the normal operation, the control circuit selects and executes one of the plurality of control modes according to a value of the output power from the plurality of control modes including the output angle modulation mode.

[0010] The second invention of the present application is the DC-DC converter of the first invention, wherein the plurality of control modes selected in the normal operation include at least the output angle modulation mode and a frequency modulation mode for adjusting a switching frequency. In the normal operation, when the output power is less than a threshold power, the control circuit executes the output angle modulation mode, and when the output power is greater than or equal to the threshold power, the control circuit executes the frequency modulation mode.

[0011] The third invention of the present application is the DC-DC converter of the first invention, wherein the plurality of control modes selected in the normal operation include at least the output angle modulation mode and a phase control mode for changing the phase difference between the voltage of the first full-bridge circuit and the voltage of the second full-bridge circuit. In the normal operation, when the output power is less than the threshold power, the control circuit executes the output angle modulation mode, and when the output power is greater than or equal to the threshold power, the control circuit executes the phase control mode.

[0012] The fourth invention of the present application is the DC-DC converter of the first invention, wherein the plurality of control modes selected in the normal operation include at least the output angle modulation mode, a frequency modulation mode for adjusting the switching frequency, and a phase control mode for changing the phase difference between the voltage of the first full-bridge circuit and the voltage of the second full-bridge circuit. In the normal operation, when the output power is less than the first threshold power, the control circuit executes the output angle modulation mode, when the output power is greater than or equal to the first threshold power and less than the second threshold power, the control circuit executes the frequency modulation mode, and when the output power is greater than or equal to the second threshold power, the control circuit executes the phase control mode.

[0013] The fifth invention of the present application is the DC-DC converter according to any one of the first to fourth inventions, wherein in any of the control modes, at the switching timing of turning on and off the switching element, the inductor current flowing through the equivalent inductor of the transformer and the inductance component is greater than or equal to the threshold current.

[0014] The sixth invention of the present application is the DC-DC converter of the fifth invention, wherein the threshold current is set such that the energy stored in the equivalent inductor is greater than or equal to the energy stored in the two capacitors of the leg where the switching element for performing switching is arranged.

[0015] The seventh invention of the present application is the DC-DC converter of the fifth invention, wherein the threshold current is I ref, when the input voltage of the first full-bridge circuit is Vx, the capacitance of the capacitor is C, the inductance of the equivalent inductor is L, and the correction coefficient is represented by α, I ref = α·Vx√(2C / L) is satisfied.

[0016] The eighth invention of the present application is a DC-DC converter according to any one of the first to seventh inventions, wherein in the initial charging operation, the control circuit sets the first voltage output period of the first full-bridge circuit to only the second half of the calculated voltage output period.

[0017] The ninth invention of the present application is a DC-DC converter according to any one of the first to eighth inventions, wherein in each of the plurality of control modes, the control circuit performs arithmetic processing using the output voltage, and when the output voltage is less than a predetermined threshold value, the control circuit performs the arithmetic processing by applying the value of the threshold value to the output voltage.

Advantages of the Invention

[0018] According to the first to ninth inventions of the present application, in normal operation, by switching the control mode according to the output power command value, switching loss can be reduced with simple control. On the other hand, in the initial charging operation, the control mode is fixed to suppress the situation where ZVS cannot be achieved due to transformer flux imbalance associated with the switching of the control mode. That is, according to the first to eighth inventions of the present application, ZVS operation can be performed with simple control in both the initial charging operation and the normal operation, and switching loss can be reduced.

[0019] In particular, according to the eighth invention, an expected inductor current can be obtained even at the start of the initial charging operation. That is, ZVS can be performed from the second switching even at the start of the initial charging operation.

[0020] In particular, according to the ninth invention, appropriate arithmetic processing can be performed even when the output voltage is low.

Brief Description of the Drawings

[0021]

Figure 1

Figure 2

Figure 3

Figure 4

Figure 5

Figure 6

Figure 7

Figure 8

Figure 9

Figure 10

Figure 11

Mode for Carrying Out the Invention

[0022] Hereinafter, embodiments of the present invention will be described with reference to the drawings. Hereinafter, regarding the "DC-DC converter" of the present invention, a dual active bridge (DAB) converter (hereinafter referred to as a DC-DC converter) will be described as an example.

[0023] <1.Circuit Configuration of DC-DC Converter> FIG. 1 is a circuit diagram of a DC-DC converter 1 according to the present embodiment.

[0024] The DC-DC converter 1 includes a pair of input / output terminals IO11 and input / output terminal IO12, and a pair of input / output terminals IO21 and input / output terminal IO22. A DC power supply E1 is connected to the pair of input / output terminals IO11 and IO12. A capacitor Cy and a load are connected to the pair of input / output terminals IO21 and IO22.

[0025] The DC-DC converter 1 transforms the power supply voltage of the DC power supply E1 input from the input / output terminals IO11 and IO12 and outputs it from the input / output terminals IO21 and IO22. Also, when a DC power supply is connected to the input / output terminals IO21 and IO22, the DC-DC converter 1 can transform the power supply voltage of the DC power supply input from the input / output terminals IO21 and IO22 and output it from the input / output terminals IO11 and IO12. That is, the DC-DC converter 1 is a converter capable of bidirectional power transmission.

[0026] The DC-DC converter 1 includes a first full-bridge circuit 10, a second full-bridge circuit 20, and a transformer T.

[0027] The transformer T includes a first winding n1 and a second winding n2. The first winding n1 and the second winding n2 are magnetically coupled. The first winding n1 is connected to the input / output terminals IO11 and IO12 via the first full-bridge circuit 10. The second winding n2 is connected to the input / output terminals IO21 and IO22 via the second full-bridge circuit 20.

[0028] The first full-bridge circuit 10 has a first leg in which the switching element Q11 and the switching element Q12 are connected in series, and a second leg in which the switching element Q13 and the switching element Q14 are connected in series. Diodes D11, D12, D13, D14, and capacitors C11, C12, C13, C14 are connected in parallel to the switching elements Q11, Q12, Q13, Q14. The switching elements Q11 to Q14 are MOS-FETs. However, the switching elements Q11 to Q14 may be IGBTs, JFETs, or the like. The diodes D11 to D14 may be actual elements or parasitic diodes. Also, the capacitors C11 to C14 may be actual elements, parasitic capacitances, or combinations of parasitic capacitances and actual elements.

[0029] Both ends of the first winding n1 of the transformer T are connected to the midpoints of the first leg and the second leg, respectively. An inductor L1 is provided between the first winding n1 of the transformer T and the midpoint of the first leg. However, the inductor L1 only needs to be connected in series to the first winding n1 or the second winding n2, and its placement location can be changed as appropriate. For example, the inductor L1 may be provided between the first winding n1 and the midpoint of the second leg. Also, the inductor L1 may be an actual element, the leakage inductance of the transformer T, or a combination of an actual element and the leakage inductance.

[0030] The second full-bridge circuit 20 has a third leg in which switching elements Q21 and Q22 are connected in series, and a fourth leg in which switching elements Q23 and Q24 are connected in series. Diodes D21, D22, D23, D24 and capacitors C21, C22, C23, C24 are connected in parallel to the switching elements Q21, Q22, Q23, Q24. The switching elements Q21 to Q24 are MOS-FETs. However, the switching elements Q21 to Q24 may be IGBTs, JFETs, or the like. The diodes D21 to D24 may be actual elements or parasitic diodes. Also, the capacitors C21 to C24 may be actual elements, parasitic capacitances, or combinations of parasitic capacitances and actual elements.

[0031] Both ends of the second winding n2 of the transformer T are connected to the midpoints of the third leg and the fourth leg, respectively. The inductor L1 may be provided between the second winding n2 and the midpoint of the third leg or the fourth leg.

[0032] The gate terminals of the switching elements Q11 to Q14 and the switching elements Q21 to Q24 are connected to the control circuit 30.

[0033] The control circuit 30 performs switching control on each of the switching elements Q11 to Q14 and Q21 to Q24 based on a target voltage value input from the outside. Specifically, the control circuit 30 includes a voltage controller 31, a control value calculation unit 32, and a switching control unit 33. The voltage controller 31 outputs a current command value and a power command value based on the target voltage value and delivers them to the control value calculation unit 32. The control value calculation unit 32 calculates control values such as the voltage output periods τ 1 , τ 2 , the switching frequency f, the phase difference δ, etc. of the full-bridge circuits 10 and 20 described later. Then, the switching control unit 33 inputs switching signals for turning on and off each of the switching elements Q11 to Q14 and Q21 to Q24 based on the control values calculated by the control value calculation unit 32.

[0034] In this way, the control circuit 30 performs switching control on each of the switching elements Q11 to Q14 and Q21 to Q24 so that the output voltage of the DC-DC converter 1 becomes the target voltage value set. In the present embodiment, the control circuit 30 soft-switches each of the switching elements Q11 to Q14 and Q21 to Q24 in order to reduce switching losses.

[0035] <2. Regarding the soft-switching operation> The soft-switching operations of each of the switching elements Q11 to Q14 and Q21 to Q24 will be described below. In the present embodiment, 3-LEVEL type DAB control is adopted.

[0036] The DC-DC converter 1 performs power transmission from one of the input / output terminals IO11 and IO12 and the input / output terminals IO21 and IO22 to the other, or from the other to one. Hereinafter, the input / output terminals IO11 and IO12 will be described as the input side (primary side), and the input / output terminals IO21 and IO22 will be described as the output side (secondary side).

[0037] FIG. 2 is a diagram showing a timing chart of the DC-DC converter 1. FIGS. 3, 4, 5, 6, 7, and 8 are diagrams for explaining the current path in the DC-DC converter 1. In FIGS. 3 to 8, the illustration of the second full-bridge circuit 20 is simplified, and the inductor L1 and the transformer T in FIG. 1 are represented by an equivalent inductor L.

[0038] In FIG. 2, the timing chart is shown only for each switching element Q11 to Q14 of the first full-bridge circuit 10. Also, V1 in FIG. 2 is the voltage between the midpoint of the switching element Q11 and the switching element Q12 and the midpoint of the switching element Q13 and the switching element Q14 shown in FIG. 1 (the voltage of the first full-bridge circuit). V2 is the voltage between the midpoint of the switching element Q21 and the switching element Q22 and the midpoint of the switching element Q23 and the switching element Q24 (the voltage of the second full-bridge circuit). In this example, the case where the absolute values of V1 and V2 are the same during each voltage output period τ 1 , τ 2 is described. I L is the inductor current flowing through the inductor L (see FIGS. 3 to 8).

[0039] The control circuit 30 performs switching control on the first full-bridge circuit 10 and the second full-bridge circuit 20 with a phase difference. Hereinafter, the phase difference between the first full-bridge circuit 10 and the second full-bridge circuit 20 is represented by δ. The control circuit 30 performs phase-shift PWM control at the switching frequency f (period 1 / f) in each of the first full-bridge circuit 10 and the second full-bridge circuit 20.

[0040] Also, hereinafter, the switching control for each switching element Q11 to Q14 of the first full-bridge circuit 10 will be described. For the second full-bridge circuit 20, switching control is performed so that the voltage V2 has the waveform shown in FIG. 2, and the description can be the same as that of the first full-bridge circuit 10. Therefore, in FIGS. 3 to 8, for simplicity of explanation, only the current path on the first full-bridge circuit 10 side is shown. In each figure, each switching element is shown by a simplified circuit symbol.

[0041] (t0 to t1) During the period from t0 to t1, the switching elements Q11 and Q14 are on, and the switching elements Q12 and Q13 are off.

[0042] In this case, as shown in FIG. 3, current flows in the order of the DC power supply E1, the switching element Q11, the inductor L, the second full-bridge circuit 20, the switching element Q14, and the DC power supply E1. The voltage V1 during this period is Hi.

[0043] At timing t1, after the switching element Q11 is turned off, the switching element Q12 is turned on with a dead time in between. During this dead time, both the switching elements Q11 and Q12 are off. At this time, due to the nature of the inductor L, inductor current I L continues to flow. Therefore, as shown in FIG. 4, current flows from each of the capacitor C11 and the capacitor C12 to the inductor L. That is, the capacitor C11 is charged and the capacitor C12 is discharged. When the capacitor C12 is discharged, the drain-source voltage of the switching element Q12 becomes zero. At this time, when the switching element Q12 is turned on, ZVS occurs.

[0044] (t1~t2) During the period from t1 to t2, the switching elements Q12 and Q14 are on, and the switching elements Q11 and Q13 are off. In this case, as shown in FIG. 5, current flows through the path of the inductor L from the switching element Q14 and the switching element Q12. The voltage V1 at this time is zero.

[0045] At timing t2, after the switching element Q14 is turned off, the switching element Q13 is turned on with a dead time in between. During this dead time, similar to the explanation in FIG. 4, the capacitor C14 is charged and the capacitor C13 is discharged. When the capacitor C13 is discharged, the drain-source voltage of the switching element Q13 becomes zero. At this time, when the switching element Q13 is turned on, ZVS occurs.

[0046] (t2~t3) During the period from t2 to t3, the switching elements Q12 and Q13 are on, and the switching elements Q11 and Q14 are off. Immediately after turning on the switching element Q13 at timing t2, as shown in FIG. 6, a current flows through the path of the DC power supply E1, the switching element Q12, the inductor L, the second full-bridge circuit 20, the switching element Q13, and the DC power supply E1. This current will flow backward to the DC power supply E1. Then, when the current flowing through the inductor L becomes 0 A, as shown in FIG. 7, a current starts to flow through the path of the DC power supply E1, the switching element Q13, the second full-bridge circuit 20, the inductor L, the switching element Q12, and the DC power supply E1. The voltage V1 during this period has the opposite polarity to that during the period from t0 to t1.

[0047] Also, at timing t3, after the switching element Q12 is turned off, the switching element Q11 is turned on with a dead time in between. Then, similar to the explanation in FIG. 4, the capacitor C12 is charged and the capacitor C11 is discharged. By discharging the capacitor C11, the drain-source voltage of the switching element Q11 becomes zero. When the switching element Q11 is turned on at this time, zero-voltage switching (ZVS) occurs.

[0048] (t3~t0) During the period from t3 to t0, the switching elements Q11 and Q13 are on, and the switching elements Q12 and Q14 are off. In this case, as shown in FIG. 8, a current flows through the path of the inductor L, the switching element Q11, and the switching element Q13. The voltage V1 at this time is zero.

[0049] At timing t0, after the switching element Q13 is turned off, a dead time is provided and the switching element Q14 is turned on. Then, similar to the explanation in FIG. 4, the capacitor C13 is charged and the capacitor C14 is discharged. By discharging the capacitor C14, the drain-source voltage of the switching element Q14 becomes zero. When the switching element Q14 is turned on at this time, zero-voltage switching (ZVS) occurs. Then, the state transitions to the state shown in FIG. 3.

[0050] By performing switching control as described above, the voltage V1 transitions as shown in the waveform of FIG. 2. Also, when the control circuit 30 performs switching control on the second full-bridge circuit 20, the voltage V2 transitions as shown in the waveform of FIG. 2. As described above, since the first full-bridge circuit 10 and the second full-bridge circuit 20 are switching-controlled with a phase difference δ, the phase difference between the rise of the voltage V1 and the rise of the voltage V2 is δ.

[0051] <3. For each control mode> Hereinafter, as examples of specific control modes, three control modes, namely, the output angle modulation mode, the frequency modulation mode, and the phase control mode, will be described.

[0052] <3-1. About the output angle modulation mode> First, the output angle modulation mode will be described. Here, a polarity inversion period during which the voltage V1 and the voltage V2 have opposite polarities to each other is represented by τ c The voltage output period of the first full-bridge circuit 10 is represented by τ 1 The voltage output period of the second full-bridge circuit 20 is represented by τ 2 τ 1 τ 2 τ c (all referring to FIG. 2) are expressed in terms of time as angles (radians). In this example, τ 1 = τ 2 and τ 1 = τ 2 = τ.

[0053] When the DC-DC converter 1 outputs low power, the control circuit 30 performs output control of the DC-DC converter 1 in the output angle modulation mode. In the output angle modulation mode, the control circuit 30 fixes the switching frequency f and the polarity inversion period τ c and changes τ 1 and τ 2 so that the output power from the DC-DC converter 1 becomes the target value.

[0054] τ 1It is changed by controlling the phase of each switching element of the first full-bridge circuit 10. Also, τ 2 is changed by controlling the on-phase of each switching element of the second full-bridge circuit 20.

[0055] τ which is a fixed value c is set so that each switching element can achieve ZVS. For this purpose, τ c needs to satisfy the conditions of the following formula (1).

Equation

[0056] Also, I ref is the current value of the inductor current I L required to achieve ZVS. As described above, for example, at the dead time (Figure 4) of timing t1, if the drain-source voltage of the switching element Q12 becomes zero after the capacitor C11 is charged and the capacitor C12 is discharged, the turn-on of the switching element Q12 becomes ZVS. That is, if the energy of the inductor L is at least more than the energy stored in each of the capacitors C11 and C12, the switching element Q12 can be ZVS. For this purpose, the following formula (2) needs to hold.

Equation

[0057] In formula (2), I L is the inductor current flowing through the inductor L. C is the capacitance of each of the capacitors C11 to C14. And formula (2) is converted into the following formula (3). Note that α in formula (3) is a correction coefficient, and an appropriate value is set as needed. Here, α = 1 is assumed.

Equation

[0058] Inductor current I L When it is equal to or greater than α·Vx√(2C / L) in Equation (3), zero-voltage switching (ZVS) of the switching element Q12 becomes possible. That is, the threshold current I ref can be expressed as α·Vx√(2C / L). That is, the threshold current I ref is set so as to be equal to or greater than the energy stored in the two capacitors C11 and C12 of the leg where the switching element Q12 for performing switching is arranged. And at each timing of turning on the switching element, if the condition of |I L |≧|I ref | is satisfied, ZVS of each switching element becomes possible.

[0059] Next, representing the output power of the DC-DC converter 1 as P and τ 1 =τ 2 =τ, the power P can be expressed by the following Equation (4).

Equation

[0060] In Equation (4), Vx and Vy are known. τ c is represented by the above Equation (1). Also, the power P is the target value to be output from the DC-DC converter 1 and is known. Therefore, when the target value of the power to be output from the DC-DC converter 1 is given, τ(=τ 1 =τ 2 ) can be calculated from the inverse function of Equation (4).

[0061] The control circuit 30 performs switching control on each of the first full-bridge circuit 10 and the second full-bridge circuit 20 so that τ 1 , τ 2 become τ obtained from Equation (4). Thereby, the target power P is output from the DC-DC converter 1.

[0062] As described above, in the output angle modulation mode, since the switching frequency f is constant, it is possible to suppress the improvement of the switching loss due to the increase in the switching frequency. Furthermore, since the switching frequency does not increase, it is possible to reduce the heat generation of the elements, particularly the inductor L.

[0063] Also, by flowing the threshold current I through the inductor L ref of the above inductor current I L ZVS of each switching element of each of the first full bridge circuit 10 and the second full bridge circuit 20 can be realized.

[0064] <3-2. About the frequency modulation mode> Next, the frequency modulation mode will be described.

[0065] The above equation (4) is in the form of a quadratic function with respect to τ. Therefore, τ has two solutions for the power P. Thus, one τ is specified by a predetermined algorithm. For example, the solution may be obtained near the vertex of the quadratic function. In this case, the frequency can be suppressed and the heat generation can be suppressed. Alternatively, the solution may be obtained at a position away from the vicinity of the vertex of the quadratic function. In particular, the following equation (5) describes a suitable calculation formula for τ that can be applied to the above-described algorithm.

Equation

[0066] In the frequency modulation mode, the output power of the DC-DC converter 1 is represented by P, and τ 1 and τ 2 are equal, and τ (=τ 1 =τ 2 ) is represented, and using tc which represents τ c in terms of time, the power P can be represented by the following equation (6).

Equation

[0067] In Equation (6), Vx and Vy are known, and τ c is represented by the above Equation (1), and τ fix is τ represented by Equation (5). Also, the power P is the target value to be output from the DC-DC converter 1 and is known. Therefore, ω can be calculated by the inverse function of Equation (6). And from ω, the switching frequency f can be calculated.

[0068] The control circuit 30 performs switching control on each switching element of the first full-bridge circuit 10 and the second full-bridge circuit 20 at the switching frequency f obtained from Equation (6). Thereby, the target power P is output from the DC-DC converter 1.

[0069] In the frequency modulation mode, when the switching frequency f is decreased, the output power increases. That is, when trying to increase the output power in the frequency modulation mode, the switching frequency f does not increase. For this reason, it is possible to suppress the improvement of switching loss due to the increase in the switching frequency. Furthermore, since the switching frequency does not increase, heat generation of elements, particularly the inductor L, can be reduced.

[0070] Also, similar to the output angle modulation mode, by flowing a current of the inductor L equal to or greater than the threshold current I ref in the inductor current I L ZVS of each switching element of the first full-bridge circuit 10 and the second full-bridge circuit 20 can be realized.

[0071] As described above, in this embodiment, it is possible to suppress the improvement of switching loss due to the increase in the switching frequency. Furthermore, since the switching frequency does not increase, heat generation of elements, particularly the inductor L, can be reduced. Also, since ZVS of each switching element can be realized, highly efficient power conversion can be realized. This control does not require complex control, so ZVS operation can be performed with simple control, and switching loss can be reduced.

[0072] <Regarding the Phase Control Mode> Next, the phase control mode will be described.

[0073] In the phase control mode, while keeping the voltage output period τ and the switching frequency f of the full-bridge circuits 10 and 20 constant, the phases of the voltage on the first winding n1 side and the voltage on the second winding n2 side are changed. That is, the phase difference δ between V1 and V2 is changed. The change in the phase difference δ can be achieved by changing the phase differences between the switching elements of each of the first full-bridge circuit 10 and the switching elements of each of the second full-bridge circuit 20.

[0074] In the case of the phase control mode, the output power P is expressed by the following formula (7). n in formula (7) is the winding ratio between the first winding n1 and the second winding n2.

Equation

[0075] As can be seen from formula (7), by changing the phase difference δ between the first full-bridge circuit 10 and the second full-bridge circuit 20, the output power P can be controlled.

[0076] When attempting to implement the phase control mode in a light load region with a small output power, it is necessary to increase the switching frequency, which is not suitable. In a heavy load region with a large output power, when attempting to implement the output angle modulation mode or the frequency modulation mode, problems such as a decrease in frequency or an increase in reactive current occur, while such efficiency deterioration does not occur in the phase control mode.

[0077] <4. Selection of Control Mode According to Output Power of DC-DC Converter> When the control circuit 30 controls the output power of the DC-DC converter 1, the control circuit 30 can be switched between an initial charging operation and a normal operation.

[0078] <4-1. Normal Operation> Normal operation is performed when stably outputting a predetermined target voltage. In normal operation, the control circuit 30 selects which control mode to execute from a plurality of control modes. If each of the above-described control modes is switched according to the output power (power command value), switching loss and heat generation of the inductor L can be efficiently reduced.

[0079] The selectable control modes include a plurality of control modes including the above-described output angle modulation mode. In the following embodiments, the selectable control modes include, in addition to the output angle modulation mode, a frequency modulation mode, a phase control mode, or both the frequency modulation mode and the phase control mode.

[0080] FIG. 9 is a diagram showing changes in the voltage output period τ (τ 1 =τ 2 =τ) and the switching frequency f with respect to the output power P when switching between the output angle modulation mode and the frequency modulation mode according to the output power P. In FIG. 9, the voltage output period τ is indicated by a solid line. Also, the switching frequency f is indicated by a dashed line. In the example of FIG. 9, the control circuit 30 executes the output angle modulation mode when the output power is less than the threshold power P T , and executes the frequency modulation mode when the output power P is greater than or equal to the threshold power P T .

[0081] As described above, as shown in FIG. 9, in the output angle modulation mode, the switching frequency f is constant, and as τ increases, the output power P increases. On the other hand, in the frequency modulation mode, the voltage output period τ is constant, and as the switching frequency f decreases, the output power P increases.

[0082] Therefore, during normal operation, the control circuit 30 divides the regions into three regions: a light load region, a medium load region, and a heavy load region according to the output power P, and switches the control mode. Specifically, the control circuit 30 calculates the output power P to be output based on the input target power value. When the output power P is less than a predetermined first threshold power, it is the light load region. When the output power P is equal to or greater than the predetermined first threshold power and less than the predetermined second threshold power, it is the medium load region. When the output power P is equal to or greater than the second threshold power, it is the heavy load region. Note that the second threshold power is a value greater than the first threshold power.

[0083] In the light load region, the control circuit 30 sets the switching frequency f so as not to increase, and executes the output angle modulation mode. Thereby, it is possible to suppress an increase in switching loss. Also, in the medium load region, the control circuit 30 executes a frequency modulation mode in which the switching frequency f can be made smaller as the target power increases. Further, in the heavy load region, the control circuit 30 executes a phase difference control mode in which the phase difference δ between V1 and V2 is changed while keeping the voltage output period τ and the switching frequency f constant.

[0084] In this way, by switching the control mode according to the target power, the control circuit 30 can perform appropriate control to suppress switching loss and heat generation of the inductor L.

[0085] Note that during normal operation, the control circuit 30 may divide the regions into two regions, a light load region and a heavy load region, according to the value of the output power P, and switch the control mode. In that case, when the output power P is less than a predetermined threshold power, it is the light load region, and when the output power P is equal to or greater than the threshold power, it is the heavy load region.

[0086] Then, the control circuit 30 executes the output angle modulation mode in the light load region and the frequency modulation mode in the heavy load region. Alternatively, the control circuit 30 executes the output angle modulation mode in the light load region and the phase difference control mode in the heavy load region.

[0087] <4-2. Initial charging operation> The initial charging operation is performed when gradually increasing the output voltage value to the target voltage value. For example, when the power supply voltage Vx of the DC power supply E1 is maintained at a predetermined voltage value (e.g., 500V), and when charging the output voltage Vy from approximately 0V to a predetermined voltage value (e.g., 400V), the output voltage value is gradually increased from about 0V close to the current Vy to the final target voltage value of 400V. Then, after the output voltage value reaches the target voltage value, it shifts to the normal operation.

[0088] In the initial charging operation, since the target voltage value and the output power P change significantly, if the load region and the control mode are made to correspond as in the normal operation, the control mode will change midway. In that case, flux imbalance occurs at the timing when the control mode is switched, and the inductor current I at the switching timing of the switching element L does not reach the threshold current I ref and ZVS does not occur. This may cause failures of the switching element, generation of noise due to hard switching, malfunction of the control circuit 30, etc.

[0089] Therefore, in the initial charging operation, the control circuit 30 executes the above-described output angle modulation mode without changing the control mode regardless of the values of the target voltage value and the output power P. Specifically, since the output power range during initial charging operates within the output angle modulation mode, when the power command value becomes a command value exceeding the region of the output angle modulation mode, the operation is limited at the point of the maximum output power in the output angle modulation mode.

[0090] Also, in this embodiment, at the start of the initial charging operation, the first voltage output period of the first full-bridge circuit 10 is set to half of the calculated voltage output period τ 1 Figure 10 shows V1, V2, and the inductor current I when the first voltage output period of the first full-bridge circuit 10 is the calculated voltage output period τ 1 in the case where it is set as such, and LIt is a diagram showing an example. FIG. 11 shows the first voltage output period of the first full-bridge circuit 10 as the calculated voltage output period τ at the start of the initial charging operation 1 when it is set to half of L the example of V1, V2 and the inductor current I

[0091] In FIGS. 10 and 11, although the switching of the second full-bridge circuit 20 is being performed because the output voltage Vy is 0V, V2 remains almost 0V. For this reason, the inductor current I L increases or decreases during the voltage output period of the first full-bridge circuit 10

[0092] Assuming a continuous state as shown in FIG. 2, the control circuit 30 calculates the voltage output period τ 1 of the first full-bridge circuit 10 and the voltage output period τ 2 of the second full-bridge circuit 20. For this reason, even at the start of the initial charging operation, as shown by the dashed-dotted line in FIG. 10, at the start of the first voltage output period of the first full-bridge circuit 10, assuming a state where the inductor current I L starts from a state smaller than -I ref , the voltage output period τ 1 is calculated

[0093] However, in reality, since the inductor current I L starts from 0 at the start of the first voltage output period of the first full-bridge circuit 10, as shown by the solid line in FIG. 10, the inductor current I L greatly exceeds I ref at t1 and t2, and the inductor current I L becomes approximately 0 at t3, and the absolute value of the inductor current I L at the time of switching at t3 is smaller than I ref , and zero voltage switching (ZVS) does not occur

[0094] Therefore, in this embodiment, as described above, at the start of the initial charging operation, the voltage output period in the first half of the first switching period of the first full-bridge circuit 10 is set to only half of the second half of the calculated voltage output period τ. 1 That is, when the calculated voltage output period τ 1 is from t0 to t1 and the intermediate time between t0 and t1 is t1', the control circuit 30 does not output power from the first full-bridge circuit 10 during the first half of t0 to t1', and only outputs power during the second half of t1' to t1. Note that during the voltage output period (t2 to t3) in the second half of the first switching period, the voltage is output as calculated.

[0095] As a result, as shown in FIG. 11, even when the inductor current I L is 0 at the start of the first voltage output period of the first full-bridge circuit 10, the inductor current I L increases to approximately the target value during the first period of t0 to t1. As a result, even at the start of the initial charging operation, the inductor current I L can be set as expected. That is, even at the start of the initial charging operation, the desired output power can be obtained. Also, ZVS can be performed starting from the second switching even at the start of the initial charging operation.

[0096] When power is transmitted from the second full-bridge circuit 20 side to the first full-bridge circuit 10 side, at the start of the initial charging operation, the voltage output period in the first half of the first switching period of the second full-bridge 20 may be set to only half of the second half of the calculated voltage output period. That is, at the start of the initial charging operation, for the first full-bridge circuit that performs switching first among the two full-bridge circuits 10 and 20, if the voltage output period in the first half of the first switching period is set to only half of the second half of the calculated voltage output period, ZVS can be performed starting from the second switching even at the start of the initial charging operation.

[0097] Also, at the start of the initial charging operation, the output voltage Vy (the voltage Vy between the input / output terminals IO21 and IO22) may be 0V. In such a case, if the calculations of the above parameters are performed with Vy = 0, abnormal values may be calculated when Vy is in the denominator, etc., and there is a risk that the DC-DC converter 1 may not operate properly.

[0098] Therefore, in this embodiment, when the output voltage Vy is smaller than a predetermined lower limit threshold value Vy' (for example, 1V), the value of the predetermined lower limit threshold value Vy' is applied to the output voltage Vy, and it is applied to the calculation process of the above parameters with Vy = Vy'. Note that the lower limit threshold value Vy' is a positive value greater than 0. Thereby, the values of the respective parameters can be set within an appropriate range.

[0099] <5. Modification Example> As described above, an embodiment of the present invention has been described, but the present invention is not limited to the above-described embodiment.

[0100] In the above embodiment, in Equation (4), τ = τ 1 = τ 2 is a condition, but τ 1 and τ 2 may be different. In this case, using the following Equation (8), τ 1 and τ 2 for outputting the target power P can be calculated.

Equation

[0101] Also, in the above embodiment, the input / output terminals IO11 and IO12 are described as the input side, and the input / output terminals IO21 and IO22 are described as the output side. However, the DC-DC converter 1 can transmit power bidirectionally. Therefore, it is possible to use the input / output terminals IO11 and IO12 as the output side and the input / output terminals IO21 and IO22 as the input / output side. In this case, since it can be described in the same manner as the above embodiment, the description thereof is omitted. Note that the DC-DC converter 1 does not have to be a bidirectional type.

[0102] Also, in the above embodiment, the polarity inversion period τ c is set as a fixed value, but it may be a variable value. In the above embodiment, if it is equal to or greater than the above fixed value, ZVS operation can be realized.

[0103] Each element appearing in the above embodiment or modification example may be appropriately combined within a range where no contradiction occurs.

Explanation of Reference Numerals

[0104] 1 DC-DC converter 10 First full-bridge circuit 20 Second full-bridge circuit 30 Control circuit C11, C12, C13, C14 Capacitor C21, C22, C23, C24 Capacitor I L Inductor current I ref Threshold current L Inductor L1 Inductor P Output power P T Threshold power Q11, Q12, Q13, Q14 Switching element Q21, Q22, Q23, Q24 Switching element T Transformer V1 Voltage of the first full-bridge circuit V2 Voltage of the second full-bridge circuit Vx Input voltage Vy Output voltage f Switching frequency n1 First winding n2 Second winding δ Phase difference τ Voltage output period τ1 Voltage output period (on the side of the first full-bridge circuit) τ2 Voltage output period (on the side of the second full-bridge circuit)

Claims

1. A first full-bridge circuit having four switching elements, including a capacitor that is a parasitic capacitance or an externally connected capacitor connected in parallel; A second full-bridge circuit having four switching elements, including a capacitor that is a parasitic capacitance or an externally connected capacitor connected in parallel; A transformer having a first winding connected to the first full-bridge circuit and a second winding connected to the second full-bridge circuit and magnetically coupled to the first winding; An inductance component connected in series to the first winding or the second winding; A control circuit for soft-switching control of each switching element of the first full-bridge circuit and the second full-bridge circuit; Comprising: The control circuit: An initial charging operation of gradually increasing an output voltage value to a target voltage value; A normal operation of stably outputting power at a predetermined target voltage value; Is switchable to, In the initial charging operation, the control circuit only executes an output angle modulation mode for adjusting a voltage output period of each of the first full-bridge circuit and the second full-bridge circuit; In the normal operation, the control circuit selects and executes one of the control modes from a plurality of control modes including the output angle modulation mode according to a value of the output power, a DC-DC converter.

2. The DC-DC converter according to claim 1, The plurality of control modes selected in the normal operation include at least: The output angle modulation mode; A frequency modulation mode for adjusting a switching frequency; Including, The control circuit, in the normal operation, When the output power is less than a threshold power, executes the output angle modulation mode; When the output power is greater than or equal to the threshold power, executes the frequency modulation mode, a DC-DC converter.

3. The DC-DC converter according to claim 1, The plurality of control modes selected in the normal operation include at least: The output angle modulation mode; A phase control mode for changing a phase difference between the voltage of the first full-bridge circuit and the voltage of the second full-bridge circuit; Including, The control circuit, in the normal operation, When the output power is less than a threshold power, executes the output angle modulation mode; When the output power is greater than or equal to the threshold power, executes the phase control mode, a DC-DC converter.

4. The DC-DC converter according to claim 1, wherein the plurality of control modes selected in the normal operation include at least the output angle modulation mode, a frequency modulation mode for adjusting the switching frequency, a phase control mode for changing the phase difference between the voltage of the first full-bridge circuit and the voltage of the second full-bridge circuit, and the control circuit, in the normal operation, when the output power is less than a first threshold power, executes the output angle modulation mode, when the output power is equal to or greater than the first threshold power and less than a second threshold power, executes the frequency modulation mode, when the output power is equal to or greater than the second threshold power, executes the phase control mode. A DC-DC converter.

5. The DC-DC converter according to any one of claims 1 to 4, wherein in any of the control modes, at the switching timing between the turn-on and turn-off of the switching element, the inductor current flowing through the equivalent inductor of the transformer and the inductance component is equal to or greater than a threshold current. A DC-DC converter.

6. The DC-DC converter according to claim 5, wherein the threshold current is set such that the energy stored in the equivalent inductor is equal to or greater than the energy stored in the two capacitors of the leg where the switching element for performing switching is arranged. A DC-DC converter.

7. The DC-DC converter according to claim 5, wherein Let the threshold current be I ref , when the input voltage of the first full-bridge circuit is Vx, the capacitance of the capacitor is C, the inductance of the equivalent inductor is L, and the correction coefficient is represented by α I ref = α · Vx√(2C / L), satisfies. A DC-DC converter.

8. The DC-DC converter according to any one of claims 1 to 4, wherein in the initial charging operation, the control circuit sets the first voltage output period of the first full-bridge circuit to only the second half of the calculated voltage output period. A DC-DC converter.

9. The DC-DC converter according to any one of claims 1 to 4, wherein in each of the plurality of control modes, the control circuit performs arithmetic processing using the output voltage, and when the output voltage is less than a lower limit threshold value which is a predetermined positive value, the control circuit applies the value of the lower limit threshold value to the output voltage and performs the arithmetic processing. A DC-DC converter.

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