Motor control device

Abstract

To provide a motor control device capable of stabilizing motor drive.SOLUTION: A motor control device includes: a PWM signal generation part for generating a PWM signal so as to determine a rotor position from phase current and to track the rotor position; and a current detection part for detecting the phase current from a signal generated at a current detection element connected to the DC side of an inverter circuit and the PWM signal. The PWM signal generation part includes: a function part capable of executing first to third output methods so that the current detection part can detect two-phase current at two points of timing within a carrier wave period and for estimating interlinkage flux of armature winding based on the phase current and an output voltage command; and a function part for estimating a revolving magnetic field angle and speed from the interlinkage flux, and outputting a changeover command for causing the PWM signal generation part to execute the first output method if a modulation ratio is less than a first threshold value, the second output method if the modulation ratio is the first threshold value or more and less than a second threshold value, and the third output method if the modulation ratio is the second threshold value or more.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] An embodiment of the present invention relates to a control device that controls a motor via an inverter circuit by PWM controlling a plurality of switching elements connected in a three-phase bridge. [Background technology]

[0002] When detecting the currents of the U, V, and W phases to control a motor, one technology uses a single shunt resistor inserted in the DC section of the inverter circuit to detect the currents.To detect all three phases of current using this method, it is necessary to generate a three-phase PWM (Pulse Width Modulation) signal so that the currents of two or more phases can be detected within one cycle of the PWM carrier wave.

[0003] For this reason, Patent Document 1 proposes a technology that can constantly detect two or more phase currents, even in a region where the modulation rate of the voltage applied to the motor is low, without increasing noise, by shifting the phase of the PWM signal within one cycle. Meanwhile, Non-Patent Document 1, for example, proposes a magnetic flux observer as a method for estimating the motor speed and angle from estimated magnetic flux. The magnetic flux observer method estimates the α-axis component Ψα and the β-axis component Ψβ of the flux linkage of the motor windings based on two-phase currents Iα and Iβ, two-phase voltages Vα and Vβ, and motor winding resistance R, for example, obtained from a current sensor, and also estimates the motor's rotating magnetic field angle, and ultimately the rotor phase angle and generated torque T.

[0004] In addition, in Patent Application No. 2022-130097, the applicant has proposed a technology to suppress estimation errors in the interlinkage magnetic flux that can occur from startup to low speed ranges when a flux observer method that estimates the phase angle and speed of the motor's rotating magnetic field from the estimated interlinkage magnetic flux is combined with a one-shunt current output method. [Prior art documents] [Patent documents]

[0005] [Patent Document 1] Patent No. 5178799 [Non-patent literature]

[0006] [Non-Patent Document 1] Inoue, et al., 3 others, Inoue, et al., 5 others, "Verification of a magnetic flux estimation method to expand the operating range of direct torque control in PMSM," Proceedings of the 2021 National Convention of the Institute of Electrical Engineers of Japan, Institute of Electrical Engineers of Japan, March 1, 2021, 5-095 Summary of the Invention [Problem to be solved by the invention]

[0007] In the magnetic flux observer described in Non-Patent Document 1, it is assumed that a current sensor such as a CT or a three-shunt current output method is used to detect the current used to estimate the magnetic flux. However, in home appliances, a one-shunt current output method is often used to reduce the cost of inverters. When a one-shunt current output method is used, if the modulation rate of the voltage applied to the motor is low from startup to low speed, the current cannot be detected, which may result in errors in the magnetic flux estimation. In particular, when the currents of the α-axis and β-axis, which vary sinusoidally, are used to calculate the magnetic flux estimation, using the previous value when the current is not detected results in a problem of large errors in the estimation of the interlinkage magnetic flux.

[0008] Furthermore, the above application provides a motor control device that, when combining a flux observer system that estimates the phase angle and speed of the motor's rotating magnetic field from estimated flux linkage with a one-shunt current output system, shifts the phase of the PWM signal within one cycle using a three-phase modulation system, suppressing estimation errors in flux linkage that may occur from startup to low-speed ranges, thereby enabling stable motor drive. However, when the phase of the PWM signal is shifted using three-phase modulation in the medium-speed range, the current detection rate drops sharply, resulting in a problem of increased estimation errors in flux linkage.

[0009] Therefore, a motor control device using a magnetic flux observer system is provided that suppresses estimation errors in the interlinkage magnetic flux over the entire motor drive range from startup to high-speed operation, thereby enabling stable motor drive. [Means for solving the problem]

[0010] The motor control device of the embodiment drives a motor via an inverter circuit that converts direct current into three-phase alternating current by controlling the on / off of a plurality of switching elements connected in a three-phase bridge in accordance with a PWM signal, a current detection element connected to a DC side of the inverter circuit and generating a signal corresponding to a current value; a PWM signal generating unit that determines a rotor position based on at least a phase current of the motor and generates a PWM signal so as to follow the rotor position; a current detection unit that detects a phase current of the motor based on a signal generated in the current detection element and the PWM signal, a first output method in which the PWM signal generation unit outputs a three-phase phase-shifted PWM signal so that the current detection unit can detect two-phase currents at two timing points within a carrier wave period of the PWM signal, and causes the current detection unit to detect currents at fixed timings; a second output method in which a two-phase phase-shifted PWM signal is output to cause the current detection unit to detect the current at a fixed timing; a third output method in which a three-phase or two-phase symmetrical PWM signal is output and the current detection unit detects the current at a fixed or variable timing; a magnetic flux estimator that estimates a flux linkage of an armature winding of the motor based on a phase current of the motor and an output voltage command; a rotating magnetic field angle and a speed of the motor based on the interlinkage magnetic flux, and causes the PWM signal generating unit to execute the first output mode if a modulation rate of the voltage applied to the motor is less than a first threshold value; If the modulation rate is equal to or greater than the first threshold and less than the second threshold, the second output method is executed; a signal switching output unit that outputs a switching command to execute the third output method if the modulation rate is equal to or greater than the second threshold value; The motor control device further includes an angle correction unit that, when the motor current cannot be detected within one electrical angle cycle, uses the speed estimated in the previous control cycle and generates an angle calculated based on the speed estimated in the previous control cycle.

[0011] A "symmetric PWM signal" refers to a PWM signal in which the direction in which the pulse width of the PWM signal is increased or decreased is the same for each phase, based on an arbitrary phase of the carrier wave period. A "phase-shifted PWM signal" refers to a PWM signal in which the direction in which the pulse width of the PWM signal is increased or decreased is different for each phase, based on an arbitrary phase of the carrier wave period. [Brief explanation of the drawings]

[0012] [Figure 1] FIG. 1 is a functional block diagram illustrating a configuration of a motor control device according to a first embodiment. [Figure 2] Diagram showing vector control block using magnetic flux observer [Figure 3] Functional block diagram showing the detailed configuration of the position estimation control unit [Figure 4] Functional block diagram showing the configuration of the integrator used in the magnetic flux estimation section [Figure 5] Diagram showing two-phase PWM signals with phase shifts [Figure 6] Flowchart showing angle correction processing in the single-shunt current detection method [Figure 7] FIG. 10 is a diagram showing an example of current detection rate according to each current detection method. [Figure 8] A flowchart showing a process for switching the current detection method depending on the modulation rate. [Figure 9] Diagram showing an example of switching current detection methods [Figure 10] FIG. 2 is a functional block diagram illustrating the configuration of a motor control device according to a second embodiment. [Figure 11] FIG. 10 is a functional block diagram showing the configuration of an integrator used in a magnetic flux estimation unit according to a third embodiment. [Figure 12]FIG. 10 is a diagram showing the relationship between the actual angle and the estimated angle in the low speed range in the first embodiment. [Figure 13] FIG. 11 is a diagram showing the relationship between the actual angle and the estimated angle in the low speed range in the third embodiment. [Figure 14] 10 is a flowchart illustrating a process performed when a motor is started according to a fourth embodiment. [Figure 15] Diagram showing each signal waveform [Figure 16] FIG. 13 is an enlarged view of a part of FIG. 12. [Figure 17] FIG. 10 is a diagram illustrating the principle of correcting the phase θ of a rotating magnetic field according to a fifth embodiment. [Figure 18] FIG. 10 is a functional block diagram illustrating the configuration of a motor control device according to a sixth embodiment. [Figure 19] A flowchart showing the process of switching the carrier frequency and current detection method for PWM control depending on the modulation rate. DETAILED DESCRIPTION OF THE INVENTION

[0013] (First embodiment) FIG. 1 is a functional block diagram showing the configuration of a motor control device according to this embodiment, which is obtained by adding several functional blocks to FIG. 1 of Patent Document 1. The DC power supply unit 1 is represented by a DC power supply symbol, but if DC power is generated from a commercial AC power supply, it may include a rectifier circuit, a smoothing capacitor, and other components. An inverter circuit 3 is connected to the DC power supply unit 1 via a positive bus 2a and a negative bus 2b, with a shunt resistor 4, which serves as a current detection element, inserted on the negative bus 2b side. The inverter circuit 3 is configured by connecting switching elements, such as N-channel power MOSFETs 5 (U+, V+, W+, U-, V-, W-), in a three-phase bridge configuration, with the output terminals of each phase connected to the respective phase windings of a motor 6, such as a brushless DC motor.

[0014] The terminal voltage of the shunt resistor 4 is detected by a current detection unit 7. The current detection unit 7 detects currents Iu, Iv, and Iw of the U, V, and W phases based on the terminal voltage and the three-phase PWM signals output to the inverter circuit 3. The phase currents detected by the current detection unit 7 are provided to a duty generation unit 8, which performs A / D conversion and reads them, and then performs calculations based on the control conditions of the motor 6, etc. As a result, duty ratios U_DUTY, V_DUTY, and W_DUTY for generating PWM signals for each phase are determined.

[0015] For example, in the case of vector control, when a rotational speed command ωref of the motor 6 is provided to the duty generation unit 8 from a microcomputer or the like that sets the control conditions, the duty generation unit 8 generates a torque current command Iqref based on the difference between the estimated rotational speed and the actual rotational speed of the motor 6. A rotor position θ of the motor 6 is determined from the phase currents Iu, Iv, and Iw of the motor 6, and the torque current Iq and excitation current Id are calculated by vector control calculation using the rotor position θ. A PI control calculation, for example, is performed on the difference between the torque current command Iqref and the torque current Iq to generate a voltage command Vq. The excitation current Id is also processed in the same way to generate a voltage command Vd, and the voltage commands Vq and Vd are converted to three-phase voltages Vu, Vv, and Vw using the rotor position θ. Phase duty ratios U, V, and W_DUTY are then determined based on these three-phase voltages Vu, Vv, and Vw.

[0016] The phase duty ratios U, V, and W_DUTY are provided to a PWM signal generator 9, which compares the levels with the carrier wave to generate a three-phase PWM signal. A lower-arm signal is also generated by inverting the three-phase PWM signal, and after adding dead time as necessary, the signal is output to a drive circuit 10. In accordance with the provided PWM signal, the drive circuit 10 outputs gate signals to the gates of the six power MOSFETs 5 (U+, V+, W+, U-, V-, and W-) that make up the inverter circuit 3. The upper-arm signal is output at a potential boosted by the required level.

[0017] The DC voltage detection unit 11 detects the voltage of the DC power supply 1 and outputs the detection result to the motor applied voltage modulation factor calculation unit 12. The motor applied voltage modulation factor calculation unit 12 calculates the modulation factor of the voltage to be applied to the motor 6 via the inverter circuit 3 based on the duty ratio information and the like input from the duty generation unit 8. The calculated modulation factor is output to the PWM output method selection unit 13. The PWM output method selection unit 13, which is a signal switching output unit, outputs a switching signal that switches the output method of the PWM signal by the PWM signal generation unit 9 according to the input modulation factor.

[0018] Fig. 2 shows a vector control block using a magnetic flux observer. In Fig. 2, each phase current detected by a current detection unit 7 is converted into Iα and Iβ, which are the α-axis component and β-axis component of the motor current, in an abc / αβ conversion unit 21 of a duty generation unit 8. The currents Iα and Iβ obtained by this conversion are provided to a position estimation control unit 23. The position estimation control unit 23 estimates the magnetic flux based on the currents Iα and Iβ and voltage commands Vα and Vβ input from a dq / αβ conversion unit 26, which will be described later.

[0019] The position estimation control unit 23 includes a magnetic flux estimation unit 23a and a speed / position estimation unit 23b. The magnetic flux estimation unit 23a estimates the α-axis and β-axis components φα and φβ of the interlinkage magnetic flux according to the following equations (1) and (2). Mutual inductance is used for L. Self-inductance, d-axis inductance Ld, and q-axis inductance Lq may also be used instead. φα=∫(Vα-R×Iα)dt-LIα …(1) φβ=∫(Vβ-R×Iβ)dt-LIβ …(2) The voltages of the α-axis and β-axis components used in the calculations of equations (1) and (2) may be values ​​calculated previously.

[0020] First, the speed and position estimation unit 23b estimates the phase θ of the rotating magnetic field relative to the α-axis and the torque T based on the estimated magnetic fluxes φα and φβ according to the following equations (3) and (4), respectively. θ=ATAN(φβ / φα) …(3) T = 3 / 2 × (number of pole pairs) × (φα × Iβ - φβ × Iα) … (4)

[0021] The integrators on the right-hand sides of equations (1) and (2) integrate the magnetic flux as shown in Fig. 4 using an incomplete integration method using a low pass filter (LPF) with a cutoff angular frequency ωc as shown in the transfer function of equation (5). "s" is a differential operator. G(S)=1 / (s+ωc) …(5) When the frequency of the magnetic flux is sufficiently greater than the cutoff angular frequency ωc, good estimation results can be obtained. ω is estimated by differentiating θ estimated using equation (3). In addition to a general LPF, an IIR (Infinite Impulse Response) filter, an FIR (Finite Impulse Response) filter, etc. may also be used as the LPF.

[0022] FIG. 3 is a functional block diagram showing the internal configuration of the position estimation control unit 23 in more detail, corresponding to the above calculation. The integrator 29 of the magnetic flux estimation unit 23a shown in FIG. 3 is actually configured as a combination of an integrator 29a and a low-pass filter (LPF) 29b, as shown in FIG. 4, and employs a so-called imperfect integration method. The output signal of the integrator 29a contains an offset. The output signal is filtered by the LPF 29b to extract the offset component, and the offset component is then subtracted by a subsequent subtractor to cancel the offset component. The LPF 29b and the subsequent subtractor may be configured with an HPF (High Pass Filter). Speed ​​control requires information about the rotor speed of the motor 6. When using a magnetic flux observer in a vector control configuration, the fact that the rotating magnetic field speed of the motor 6 and the rotor speed are steadily the same is utilized.

[0023] Referring again to Figure 2, a rotational speed command ωref for the motor 6 is provided by a host control device such as a microcomputer that sets control conditions. A speed control unit 24 generates a torque current command Iqref based on the difference between the rotational speed command ωref and the rotational speed ω estimated by the position estimation unit 23. An αβ / dq conversion unit 22 calculates a torque current Iq and an excitation current Id from the currents Iα and Iβ by vector control calculation using the rotor position θ.

[0024] In the current control unit 25, for example, a PI control calculation is performed on the difference between the torque current command Iqref and the torque current Iq to generate a voltage command Vq. A similar process is performed on the excitation current Id side to generate a voltage command Vd. A space vector generation unit 27 converts the voltage commands Vq and Vd into three-phase voltages Vu, Vv, and Vw using the rotor position θ. Then, duty ratios U_DUTY, V_DUTY, and W_DUTY for generating PWM signals for each phase are determined based on the three-phase voltages Vu, Vv, and Vw.

[0025] The phase duty ratios U, V, and W_DUTY are provided to a PWM generator 28, which compares the levels with the carrier to generate a two-phase or three-phase PWM signal. A lower-arm signal is also generated by inverting the two-phase or three-phase PWM signal, and after adding dead time as necessary, the signal is output to the drive circuit 10. The method by which the PWM generator 28 generates the phase-shifted three-phase PWM signals is, for example, the method of the fourth embodiment disclosed in Patent Document 1. In the method by which the PWM generator 28 generates the phase-shifted two-phase PWM signals, as shown in FIG. 5, for example, an inverted triangular wave is used as the comparison carrier for the U phase, an inverted sawtooth wave is used as the comparison carrier for the V phase, and a sawtooth wave is used as the comparison carrier for the W phase, for the duty during two-phase modulation.

[0026] The motor applied voltage modulation rate calculation unit 12 shown in FIG. 1 calculates the modulation rate of the motor applied voltage for each carrier period based on Vα and Vβ calculated by the duty generation unit 8, as shown in equation (6). (Modulation rate) = 100 × Vdc / (√3 × √(Vq2 + Vd2)) … (6) The calculation result is output to the PWM output method selection unit 13. Based on this information, the PWM output method selection unit 13 outputs a signal to the PWM signal generation unit 9 to switch the PWM output signal. The PWM signal generation unit 9 also outputs a current detection timing signal to the current detection unit 7. Note that the modulation rate of the motor applied voltage may simply be substituted by the motor rotation speed or the like.

[0027] Since the currents Iα and Iβ vary sinusoidally over time, if the phase currents of the motor 6 cannot be detected in the one-shunt current output method, estimating the magnetic flux using the currents Iα and Iβ estimated in the previous control cycle may result in poor estimation accuracy. Furthermore, since the angle varies in a sawtooth waveform, if the phase currents of the motor 6 cannot be detected, using the angle estimated in the previous control cycle will result in errors in the calculations of the vector control system. In this embodiment, the control cycle is equal to the carrier wave cycle.

[0028] FIG. 6 shows a flowchart of the angle correction process in the one-shunt current output method using a flux observer. The currents and voltages of the α-axis and β-axis are calculated (S1), and flux observer control is performed (S2). If the phase current can be detected (S3; OK), normal control is performed, and calculations are sequentially performed to estimate the angle θ, load torque T, and speed ω (S4 to S6). On the other hand, if the phase current cannot be detected (S3; NG), the previously estimated speed ω is used (S7), and the angle θ calculated by integrating that speed ω is used (S8). This process is performed by the flux estimation unit 23a, which also serves as the angle correction unit.

[0029] FIG. 7 shows an example of the current detection rate for each current detection method. As the motor rotation speed and load torque increase, the modulation rate of the motor-applied voltage approaches 100%. The PWM output method selector 13 switches the PWM signal output method used by the PWM signal generator 9 and the phase current detection method used by the current detector 7 depending on the modulation rate. When the modulation rate is in a low range, the PWM signal generator 9 generates a three-phase PWM signal by shifting the output phase of the PWM signal pulse for each phase using a method different from the conventional method described in Patent Document 1. This is referred to as the first output method. When the modulation rate is in an intermediate range, around 50% to 70%, the PWM signal generator 9 generates a PWM signal by shifting the output phase of two-phase PWM signal pulses, as shown in FIG. 5. This is referred to as the second output method. When the modulation rate is in a high range, the selector outputs a switching command to generate two-phase or three-phase PWM signals, each of which is a pulse signal symmetrical with respect to the midpoint of the PWM period, as shown in FIG. 7 of Patent Document 1. This is referred to as the third output method.

[0030] Hereinafter, the method of causing the current detection unit 7 to detect two-phase currents at fixed timing, corresponding to the above-mentioned first output method, will be referred to as the first detection method. Also, the method of causing the current detection unit 7 to detect two-phase currents at fixed timing, corresponding to the second output method, will be referred to as the second detection method. Furthermore, the method of causing the current detection unit 7 to detect two-phase currents at fixed or variable timing, corresponding to the third output method, will be referred to as the third detection method. When driving the motor 6 from startup to the high-speed region, the PWM output method selector 13 switches between the first detection method, the second detection method, and the third detection method in that order depending on the modulation rate.

[0031] Conversely, when the speed of the motor 6 is shifted from the high-speed region to the low-speed region, the detection methods are switched in the order of the third detection method, the second detection method, and the first detection method. In this case, by adding hysteresis to the threshold value for switching the current detection method, frequent switching of the current detection method is suppressed even when the speed of the motor 6 increases or decreases.

[0032] FIG. 8 shows a flowchart for switching the current detection method. After the motor 6 starts, the process branches to case 1 (S9) and selects the first or second detection method. The modulation rate is compared with the current detection method switching threshold (S10), and if the modulation rate is less than, for example, 50%, the first detection method (S11) is selected, and the process branches to case 1 again (S12). If the modulation rate is 50% or more, the second detection method is selected (S13), and the process moves to case 2 (S14), where the process branches and selects one of the first to third detection methods. A modulation rate of 50% corresponds to the first threshold.

[0033] For example, if the modulation rate is less than 45% (S15), the first detection method is selected (S16), and the process returns to case 1 (S17). If the modulation rate is 45% or higher (S18), the current detection method switching threshold is again compared with the modulation rate. If the modulation rate is less than 60%, for example (S18), the second detection method is continued (S19). If the modulation rate is 60% or higher (S18), the third detection method is selected (S21), and the process transitions to case 3 (S22). A modulation rate of 60% corresponds to the second threshold. After transitioning to case 3, if the modulation rate is less than 55%, for example (S23), the second detection method is selected (S24), and the process transitions to case 2 (S25). If the modulation rate is 55% or higher (S23), the third detection method is selected (S26), and the process returns to case 3 (S27). A modulation rate of 55% corresponds to the third threshold. The threshold value may be changed as appropriate, taking into account current ripple, A / D conversion time, etc.

[0034] Figure 9 shows an example in which the one-shunt current detection method using a magnetic flux observer is switched to the third detection method, the second detection method, and the first detection method to drive the motor 6. If the drive frequency is increased after starting the motor 6, the detection methods are switched in the order of the first, second, and third detection methods. If the drive frequency is decreased after switching to the third detection method, the detection methods are switched in the order of the third, second, and first detection methods.

[0035] As described above, according to this embodiment, the PWM signal generation unit 9 determines the rotor position based on at least the phase currents of the motor 6 and generates a PWM signal to track the rotor position. The current detection unit 7 detects the phase currents of the motor 6 based on the signal generated in the shunt resistor 4 and the PWM signal. The PWM signal generation unit 9 outputs PWM signals using first to third output methods so that the current detection unit 7 can detect two-phase currents at two timing points within the carrier wave period according to the first to third detection methods. In the first output method, the duty ratio of one of the three phases is increased or decreased in both directions, lagging and leading, based on an arbitrary phase of the carrier wave period, while the duty ratio of the other phase is increased or decreased in one direction, lagging or leading, and the duty ratio of the remaining phase is increased or decreased in the opposite direction.

[0036] The magnetic flux estimation unit 23a estimates the armature winding of the motor 6 based on the phase current of the motor 6 and the output voltage command. The magnetic flux linkage of the wire is estimated, and the angle of the rotating magnetic field and the speed of the motor 6 are estimated based on the magnetic flux linkage. The PWM output method selection unit 13 outputs a switching command to the PWM signal generation unit 9 to execute the first output method when the modulation rate of the voltage applied to the motor is in a low range, the second output method when it is in an intermediate range, and the first output method when it is in a high range. Accordingly, the current detection unit 7 switches to one of the first to third detection methods. When the motor current cannot be detected, the magnetic flux estimation unit 23a uses the previously estimated speed and generates an angle calculated based on the previously estimated speed.

[0037] Here, the condition that the motor current cannot be detected is when the duration of the PWM signal for which the current is to be detected within one electrical angle cycle is shorter than the time within which the current can be detected, taking into account factors such as current ripple and A / D conversion time, for example, 5 to 10 μsec. Therefore, even if a PWM signal is generated so that two-phase currents can be detected at two timing points, the detection rate does not always reach 100%; the actual detection rate is generally in the range of 70% to 100%. Furthermore, the detection rate drops sharply when the modulation rate exceeds approximately 50 to 60%, which may cause unstable motor drive depending on the application.

[0038] Therefore, by adopting the second detection method in the region where the modulation rate exceeds approximately 50 to 60%, the current detection rate can be increased compared to the first detection method. With this configuration, even when the single-shunt current detection method is applied, the three-phase currents Iu, Iv, and Iw can be detected with a high current detection rate from low to high modulation rates of the motor applied voltage, and magnetic flux can be estimated based on the α-axis current, β-axis current, and voltage command vector. Additionally, even when the phase currents of motor 6 cannot be detected, the previously estimated speed value can be used to prevent deterioration in position estimation accuracy.

[0039] (Second embodiment) In the following, the same parts as in the first embodiment are denoted by the same reference numerals and their explanations are omitted, and only the differences are explained. In the first embodiment, the angle θ and speed ω of the motor 6 are estimated based on the estimated magnetic flux, and the estimation is applied to vector control. In the second embodiment, a case where magnetic flux observer control is applied to direct torque control is shown.

[0040] As shown in Fig. 10, direct torque control using a flux observer uses a UVW / αβ conversion unit 31, a torque calculation unit 32 which is a direct torque control execution unit, a binary level output unit 33, and a switching table 34 instead of the αβ / dq conversion unit 22 and the speed estimator 24 to space vector generator 27. A target torque command Tref and a target magnetic flux command φref are input from a higher-level control device instead of the rotational speed command ωref. Then, a three-phase PWM signal is generated by referring to the switching table 34. Direct torque control is a known technique, so a detailed description will be omitted. Note that the motor can be applied to a synchronous reluctance motor or an induction motor in addition to a permanent magnet motor.

[0041] (Third embodiment) In the first embodiment, an inexact integration method is used for the integrators on the right-hand sides of equations (1) and (2). In the second embodiment, a second-order generalized integration method for the transfer function shown in FIG. 11 and equation (7) is used to integrate the magnetic flux in the same case. G(S)=kω' / (s2+ Kω'S+ω'2) …(7) ω' is the natural angular frequency of the second-order filter, and k is a coefficient that determines the attenuation.

[0042] The incomplete integration method can estimate the magnetic flux well when the magnetic flux frequency is sufficiently higher than ωc, but the use of LPF 29b causes the accuracy to decrease in the low speed range indicated by the double-headed arrow in Fig. 12. In contrast, the second-order generalized integration method shown in Fig. 13 improves the frequency characteristics in the low speed range, making it possible to expand the operable range.

[0043] As described above, according to the third embodiment, when the modulation rate of the voltage applied to the motor is in a low range, the output phase of the PWM signal pulse for each phase is shifted by the method of Patent Document 1 to generate a three-phase PWM signal for sensorless operation. By integrating the magnetic flux using the second-order generalized integral method, the estimation accuracy of the motor magnetic flux can be maintained even in slower speed ranges, making sensorless control possible.

[0044] (Fourth embodiment) According to the third embodiment, the estimation accuracy of the motor magnetic flux can be maintained even at low speeds. However, when starting the motor 6, the motor magnetic flux cannot be estimated with sufficient accuracy. For this reason, when starting the motor 6, it is possible to perform forced commutation, in which a d-axis current is applied at an angle corresponding to the command speed, to increase the rotation speed of the motor 6 and then switch to sensorless operation. However, if the load torque of the motor 6 is excessive, there is a risk of loss of synchronization. Applying a sufficient d-axis current during forced commutation can handle the load at startup, but this increases the power required at startup.

[0045] Therefore, in the fourth embodiment, when the motor 6 is started, forced commutation is performed to apply a d-axis current according to the estimated motor magnetic flux angle. In addition, a torque current command Iqref is also applied based on the difference between the command rotation speed and the estimated rotation speed of the motor 6. As a result, when the motor load is small during forced commutation at start-up, a d-axis current is applied, and when the motor load is large, a q-axis current is applied. This makes it possible to change the motor current according to the load even during forced commutation at start-up, and start the motor 6 without wasting power.

[0046] In the start-up control sequence shown in FIG. 14, positioning control is first performed (S31). Here, the excitation current command Idref is set to a predetermined value, the torque current command Iqref is set to zero, and the angle is set to a target angle. Next, "forced commutation control 1" is performed (S32). The current commands Idref and Iqref, as well as the angle, are set to the same as in step S31. Then, the rotation speed command ωref is increased, and if the rotation speed command ωref exceeds the rotation speed threshold 1 in step S33 (S33; YES), the process proceeds to "forced commutation control 2" (S34).

[0047] In "forced commutation control 2", the excitation current command Idref is set to, for example, about half the predetermined value in "forced commutation control 1". The torque current command Iqref uses the result of speed control in vector control, and the angle is set to a value estimated by the magnetic flux observer. If the rotation speed command ωref exceeds rotation speed threshold 2 in the following step S35 (YES), the process transitions to sensorless control (S36). Here, the excitation current command Idref is set to zero. Then, a similar determination is made as in step S35 (S37), and if the rotation speed command ωref exceeds rotation speed threshold 2, the startup process is terminated, or if it is equal to or less than rotation speed threshold 2, the process returns to step S34.

[0048] Fig. 15 shows the operating waveforms of the motor when forced commutation is performed according to the estimated motor magnetic flux angle, and Fig. 16 shows an enlarged view of the rectangular area within the load application section of Fig. 15. The torque current component Iq also increases when the load increases during forced commutation. This shows that the motor output torque can be varied according to the load torque, even during forced commutation.

[0049] As described above, according to the fourth embodiment, when starting the motor 6, forced commutation is performed to apply a d-axis current in accordance with the estimated motor magnetic flux angle, and a torque current command Iqref is also applied based on the difference between the estimated rotational speed of the motor 6 and the command rotational speed, so that the output torque of the motor 6 can be varied in accordance with the load torque even during forced commutation. This allows the motor 6 to be started without wasting power.

[0050] (Fifth embodiment) In the first and second embodiments, the phase θ of the rotating magnetic field is estimated using equation (3). As the load torque of the motor increases, the error between the phase θ of the rotating magnetic field and the phase of the magnet increases. Therefore, the phase θ of the rotating magnetic field is corrected using equation (9) using the torque angle δ calculated using equation (8) from the q-axis inductance Lq, d-axis inductance Ld, q-axis current Iq, d-axis current Id, and magnet magnetic flux Φ. Figure 17 is a diagram showing the principle of correction. δ=tan -1 {(Lq × Iq) / (Ld × Id + Φ)} …(8) θ=θs-δ …(9)

[0051] As described above, according to the fifth embodiment, it is possible to reduce the position estimation error between the phase θ of the rotating magnetic field and the phase of the magnet that occurs when the load torque of the motor 6 increases when starting the motor 6. This allows the motor 6 to be driven stably even when the load torque of the motor 6 increases.

[0052] (Sixth embodiment) When shifting the output phase of the PWM signal pulse for each phase as in the above-described embodiments, if the carrier frequency is within the human audible range, such as 4 kHz, noise may become a problem. On the other hand, if the carrier frequency is increased across the entire motor drive range, there is a concern that the switching loss of the inverter circuit 3 will increase, resulting in a decrease in overall efficiency. Therefore, in the sixth embodiment, when the PWM frequency output method selector 13 causes the current detection unit 7 to switch between the first and second current detection methods and the third current detection method depending on the modulation rate, the PWM frequency changer 41 shown in FIG. 18 changes the carrier frequency.

[0053] 19, if the modulation rate is less than 50% in step S10, the carrier frequency is set to, for example, 8k to 16kHz or more (S27), and the first detection method is executed (S11). Similarly, if the modulation rate is less than 55% in step S23, the carrier frequency is set to, for example, 8k to 16kHz or more (S30), and the second detection method is executed (S24).

[0054] On the other hand, if the modulation rate is 60% or higher in step S18, the carrier frequency is set low, for example, to 4 kHz (S29), and the third detection method is executed (S21). With this detection method, noise caused by the carrier frequency is reduced compared to when the output phase is shifted. It is not important to change the current detection method or the carrier frequency in any order. The carrier frequency may be changed in stages or all at once. In addition, the modulation rate of the motor applied voltage may be simply substituted by the motor rotation speed, etc.

[0055] (Other embodiments) The first to third embodiments of Patent Document 1 may be applied as a method for expanding the pulse width of each phase PWM signal. The magnetic fluxes φα and φβ may be estimated using the calculations of equations (10) and (11). φα=∫(Vα-R×Iα)dt …(10) φβ=∫(Vβ-R×Iβ)dt …(11) The control period does not necessarily have to match the carrier wave period, but may be set to more than twice the carrier wave period, or half the carrier wave period, for example.

[0056] Although several embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be embodied in various other forms, and various omissions, substitutions, and modifications can be made without departing from the spirit of the invention. These embodiments and their modifications are included within the scope and spirit of the invention, and are also included in the scope of the invention and its equivalents as defined in the claims. [Explanation of symbols]

[0057] In the drawing, 3 indicates an inverter circuit, 4 indicates a shunt resistor, 5 indicates a power MOSFET, 6 indicates a motor, 7 indicates a current detection unit, 8 indicates a duty generation unit, 9 indicates a PWM signal generation unit, 12 indicates a motor applied voltage modulation rate calculation unit, 13 indicates a PWM output method selection unit, and 23 indicates a position estimation control unit.

Claims

1. A motor control device drives a motor via an inverter circuit that converts direct current into three-phase alternating current by controlling the on / off of a plurality of switching elements connected in a three-phase bridge in accordance with a PWM signal, a current detection element connected to a DC side of the inverter circuit and generating a signal corresponding to a current value; a PWM signal generating unit that determines a rotor position based on at least a phase current of the motor and generates a PWM signal so as to follow the rotor position; a current detection unit that detects a phase current of the motor based on a signal generated in the current detection element and the PWM signal, a first output method in which the PWM signal generation unit outputs a three-phase phase-shifted PWM signal so that the current detection unit can detect two-phase currents at two timing points within a carrier wave period of the PWM signal, and causes the current detection unit to detect currents at fixed timings; a second output method in which a two-phase phase-shifted PWM signal is output to cause the current detection unit to detect a current at a fixed timing; a third output method in which a three-phase or two-phase symmetrical PWM signal is output and the current detection unit detects the current at a fixed or variable timing; a magnetic flux estimator that estimates a flux linkage of an armature winding of the motor based on a phase current of the motor and an output voltage command; a rotating magnetic field angle and a speed of the motor based on the interlinkage magnetic flux, and causes the PWM signal generating unit to execute the first output method if a modulation rate of the voltage applied to the motor is less than a first threshold value; If the modulation rate is equal to or greater than the first threshold and less than the second threshold, the second output method is executed; a signal switching output unit that outputs a switching command to execute the third output method when the modulation rate is equal to or greater than the second threshold value; and an angle correction unit that, when the current detection unit cannot detect the phase current within one electrical angle cycle, uses the speed estimated in a previous control cycle and generates an angle calculated based on the speed estimated in the previous control cycle.

2. 2. The motor control device according to claim 1, wherein the signal switching output unit outputs a switching command to execute the third output method, and then outputs a switching command to execute the second output method when the modulation rate falls below a third threshold that is set to be equal to or greater than the first threshold and less than the second threshold.

3. In the first output method, the PWM signal generation unit One of the three-phase PWM signals has a pulse width that is increased or decreased in both directions, that is, in the delay and advance directions, based on an arbitrary phase of the carrier wave period. The other phase increases or decreases the pulse width in one direction, either the delay side or the advance side, based on an arbitrary phase of the carrier wave period, 2. The motor control device according to claim 1, wherein the remaining one phase is generated by increasing or decreasing the pulse width in a direction opposite to the direction with reference to an arbitrary phase of the carrier wave period.

4. 2. The motor control device according to claim 1, wherein the magnetic flux estimator estimates the flux linkage by performing time integration based on a motor current value obtained by converting the three-phase AC motor current into a two-phase AC motor current, an output voltage command obtained by converting a DC component output voltage command into a two-phase AC component, and a winding resistance value of the motor.

5. 5. The motor control device according to claim 4, wherein the magnetic flux estimation unit estimates the flux linkage by time-integrating, using a double integrator, the output voltage command converted into two phases, a value calculated from the two-phase AC current of the motor, and a winding resistance value of the motor.

6. 6. The motor control device according to claim 1, wherein the magnetic flux estimator corrects a position estimation error that increases or decreases due to a load torque generated at a motor angle calculated from the estimated magnetic flux, using a torque angle calculated from a q-axis inductance Lq, a d-axis inductance Ld, a q-axis current Iq, a d-axis current Id, and a magnet magnetic flux Φ.

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