receiver

Abstract

To provide a receiver.SOLUTION: A receiver according to the present invention includes: an I branch including an I branch mixer that down-converts an RF (radio frequency) signal and outputs an I component and an I branch channelization filter that separates a baseband signal from the output signal of the I branch mixer; a Q branch including a Q branch mixer that down-converts the RF signal and outputs a Q component and a Q branch channelization filter that separates the baseband signal from the output signal of the Q branch mixer; a frequency-dependent mismatch estimator that calculates frequency-dependent mismatch between the I branch and the Q branch; and a frequency-dependent mismatch compensation unit that compensates for the frequency-dependent mismatch between the I branch and the Q branch on the basis of the calculation result of the frequency-dependent mismatch estimator.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present technology generally relates to receivers. [Background technology]

[0002] A direct conversion receiver uses quadrature mixing to simultaneously downgrade an RF (radio frequency) band signal to baseband (BB). Quadrature mixing uses two mixers to multiply the received signal by two local oscillator (LO) signals that have a 90-degree phase difference. The I and Q signals, which are split into baseband I and Q branches, are passed through a low-pass filter (LPF) and amplifier before being converted to discrete-time signal samples by an ADC for processing. Summary of the Invention [Problem to be solved by the invention]

[0003] An ideal receiver operates under three assumptions: a 90-degree phase difference between the two LO signals, the same gain, and the same frequency characteristics of the LPF, amplifier, and DAC in the two I / Q branches. Actual semiconductor processes create component-level mismatches that are unrelated to the design, which causes these three assumptions to not be met, resulting in LO phase / gain mismatch and baseband filter mismatch.

[0004] The above-mentioned receiver mismatch has both frequency-independent and frequency-dependent characteristics, and such IQ mismatch of the receiver acts by adding a conjugate image to the original signal to be received, degrading the signal-to-noise ratio (SNR) of the received signal like an interferer and increasing the bit error rate (BER).

[0005] One of the problems to be solved by this technology is to provide a technology that can compensate for the two mismatches that are the drawbacks of the above-mentioned conventional technology. [Means for solving the problem]

[0006] This embodiment is a receiver, and the receiver includes: an I branch including an I branch mixer that downconverts an RF (radio frequency) signal to output an I component and an I branch channelization filter that separates a baseband signal from the output signal of the I branch mixer; a Q branch including a Q branch mixer that downconverts the RF signal to output a Q component and a Q branch channelization filter that separates the baseband signal from the output signal of the Q branch mixer; a frequency-dependent mismatch estimator that calculates frequency-dependent mismatch between the I branch and the Q branch; and a frequency-dependent mismatch compensator that compensates for the frequency-dependent mismatch between the I branch and the Q branch based on the calculation result of the frequency-dependent mismatch estimator.

[0007] In one aspect of this embodiment, the frequency-dependent mismatch compensator further includes a compensation filter in an I path having a transfer function corresponding to the transfer function of the Q-branch channelization filter divided by the transfer function of the I-branch channelization filter.

[0008] In one aspect of this embodiment, the frequency dependent mismatch compensator further includes a delay line in a Q path corresponding to a delay of the compensation filter.

[0009] In one aspect of this embodiment, the frequency-dependent mismatch estimation unit calculates a transfer function of a compensation filter that compensates for the frequency-dependent mismatch from a value obtained by normalizing a time average of a tone-image correlation of a signal output from the channelization filter by an average power of the signal output from the channelization filter.

[0010] In one aspect of this embodiment, when the value is m(k), the frequency dependent mismatch estimator

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[0011] In one aspect of this embodiment, the frequency-dependent mismatch estimator calculates a time average of a tone-image correlation using a pilot signal including an SSB multitone, normalized by the average power of the signal output from the channelization filter.

[0012] This embodiment is a receiver, and the receiver includes: an I branch including an I branch mixer that down-converts an RF (radio frequency) signal to output an I component and an I branch channelization filter that separates a baseband signal from the output signal of the I branch mixer; a Q branch including a Q branch mixer that down-converts the RF signal to output a Q component and a Q branch channelization filter that separates the baseband signal from the output signal of the Q branch mixer; a frequency-independent mismatch estimator that calculates frequency-independent mismatch between the I branch and the Q branch; and a frequency-independent mismatch compensator that compensates for frequency-dependent mismatch between the I branch and the Q branch based on a calculation result of the frequency-independent mismatch estimator.

[0013] In one aspect of this embodiment, the independent mismatch estimation unit calculates the gain mismatch and phase mismatch of the I branch mixer and the Q branch mixer from a value obtained by normalizing the time average of the squared output signals of the I branch mixer and the Q branch mixer by the time average of the power of the output signals of the I branch mixer and the Q branch mixer.

[0014] In one aspect of this embodiment, the non-dependence inconsistency estimation unit

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[0015] In one aspect of this embodiment, the frequency-independent mismatch compensation unit includes a first multiplier connected to the I branch, a second multiplier connected to the Q branch, a third multiplier that amplifies an output signal of the first multiplier, and an adder that adds the output of the second multiplier and the output of the third multiplier.

[0016] In one aspect of this embodiment, the gain of the first multiplier is gRX and the gain of the second multiplier is cos(1 / θ RX ), and the gain of the third multiplier is tan(θ RX ) corresponds to [Effects of the Invention]

[0017] This embodiment provides the advantage that it is possible to compensate for frequency dependent and frequency independent mismatches in the receiver. [Brief explanation of the drawings]

[0018] [Figure 1] FIG. 1 shows a schematic diagram of a practical receiver with mismatch. [Figure 2] FIG. 10 is a diagram illustrating an overview of a frequency-dependent mismatch compensator. [Figure 3] FIG. 10 is a diagram illustrating an overview of a frequency-independent mismatch compensator. DETAILED DESCRIPTION OF THE INVENTION

[0019] The present embodiment will be described below with reference to the accompanying drawings. Fig. 1 is a diagram showing an overview of a practical receiver 10 with mismatch. Referring to Fig. 1, the receiver 10 according to this embodiment includes an I branch 100 including an I branch mixer 110 that downconverts an RF signal (radio frequency signal, r(t)) and outputs an I component, and a Q branch 200 including a Q branch mixer 210 that downconverts the RF signal r(t) and outputs a Q component.

[0020] In one embodiment, the I-branch 100 may further include an I-branch channelization filter 120 that separates a baseband signal from the output signal of the I-branch mixer 110, and the Q-branch 200 may further include a Q-branch channelization filter 220 that separates a baseband signal from the output signal of the Q-branch mixer 210.

[0021] The receiver 10 can include a frequency-dependent mismatch estimator (FD est) 310 that calculates the frequency-dependent mismatch between the I branch 100 and the Q branch 200, and a frequency-dependent mismatch compensator 320 that compensates for the frequency-dependent mismatch between the I branch 100 and the Q branch 200 based on the calculation result of the frequency-dependent mismatch estimator 310.

[0022] In addition, in one embodiment of the receiver 10, the receiver 10 may include a frequency-independent mismatch estimator 410 that calculates the frequency-independent mismatch between the I branch 100 and the Q branch 200, and a frequency-independent mismatch compensator 420 that compensates for the frequency-independent mismatch between the I branch 100 and the Q branch 200 based on the calculation result of the frequency-independent mismatch estimator 410.

[0023] In the illustrated embodiment, receiver 10 is illustrated as including both a frequency-dependent mismatch estimator 310 and a frequency-dependent mismatch compensator 320, and a frequency-independent mismatch estimator 410 and a frequency-independent mismatch compensator 420, but receiver embodiments not shown may include only one of a frequency-dependent mismatch estimator and a frequency-dependent mismatch compensator, or a frequency-independent mismatch estimator and a frequency-independent mismatch compensator.

[0024] 1, a received signal r(t) is branched and input to an I branch 100 and a Q branch 200. The signal r(t) is down-converted by a mixer 110 in the I branch 100 into a signal cosω RX The signal r(t) is mixed with the down-converted signal in the mixer 210 of the Q branch 200 and is then down-converted.

[0025] The signal x downconverted by the mixer 110 in the I branch I (t) is input to the I-branch channelization filter 120, which converts the baseband signal y IThe impulse response of the I-branch channelization filter 120 in the time domain is h I RX The signal x down-converted by the mixer 210 in the Q branch can be expressed as Q (t) is input to the Q-branch channelization filter 220. The Q-branch channelization filter 220 、 baseband signal y Q The Q-branch channelization filter 220 outputs the impulse response in the time domain as h Q RX It can be displayed as (t).

[0026] In one embodiment, the I-branch channelization filter 120 and the Q-branch channelization filter 220 may be low pass filters (LPFs). In one embodiment, the outputs of the I-branch channelization filter 120 and the Q-branch channelization filter 220 may be provided to an analog-to-digital converter (ADC). In one embodiment, the I-branch 100 may further include a delay line 123 that delays the signal by a corresponding delay time to match the delay of the Q-branch 200. The outputs of the I-branch channelization filter 120 and the Q-branch channelization filter 220 may be provided to the analog-to-digital converter (ADC).

[0027] Ideally, the signals provided to each mixer by a local oscillator (not shown) for downconversion have the same amplitude, and the signal provided to the mixer 110 in the I branch and the signal provided to the mixer 210 in the Q branch have a phase difference of 90 degrees.

[0028] However, there is a difference in the magnitude of the signals provided by the actual local oscillator (LO), and the phase difference of the provided signals is not exactly 90 degrees. In such a case, the mismatch in signal magnitude is called gain mismatch (g). RX and the phase mismatch is expressed as θ RXHowever, the gain and phase mismatch of the mixer during downconversion cannot be distinguished from the LO mismatch and do not need to be displayed separately. As will be explained later, the gain and phase mismatch are independent of time (and frequency).

[0029] L is the mismatched signal provided to the mixer RX (t) is expressed by the following formula:

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[0030] The output signal x(t) down-converted to the mismatched LO signal is expressed by the following equation:

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[0031] Furthermore, formula 4 can be summarized in the frequency domain as formula 5 below.

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[0032] In the ideal case where there is no effect of mismatch, g RX is 1 and θ RX is 0, so the β value is 0 and the transmission frequency ω TX and receiving frequency ω RX Since and are the same, the baseband signal r BBHowever, if gain mismatch and phase mismatch occur, the β value is not 0, and the IQ mismatch results in an unintended r BB (t) is the conjugate image of the signal, r BB * It can be seen that the (t) component is further produced.

[0033] The conjugate image thus generated is r BB * The (t) component acts as an interference signal on the desired signal, degrading the signal-to-noise ratio and reception performance of the receiver. Also, as shown in Equation 5, when viewed from the frequency domain, the desired signal r BB The coefficients α and β of (t) and its conjugate image are not functions of frequency but are constants, and in this sense this can be said to be frequency-independent IQ mismatch.

[0034] Baseband I-branch 100 and Q-branch 200 filters h RX I (t), h RX Q The mismatch in (t) includes the difference in poles and zeros due to the circuit elements on the semiconductor process that implement the analog baseband anti-aliasing low-pass filter, and the difference in propagation delay up to just before the ADC sample and hold.

[0035] When there is an IQ mismatch in the baseband filter, the input and output of the filter stage satisfy the following equation 6.

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[0036] (x i (t): Output signal of I branch mixer, x Q (t): Output signal of Q-branch mixer, h i RX (t): Impulse response of the I-branch channelization filter, h Q RX (t): Impulse response of the Q-branch channelization filter,

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[0037] In Equation 6, the target signal x(t) is not transmitted as is, but is transmitted as a conjugate image x * (t) is added. Looking at this from the frequency domain, the coefficients H e RX (f), H o RX Since (f) is a function of frequency, and the image interference is also a function of frequency, this can be described as a frequency-dependent IQ mismatch.

[0038] When the IQ mismatch of the LO, mixer, and baseband branch are all calculated, the demodulated signal y(t) can be expressed by the following equation 7. Due to the mismatch between the I branch 100 and the Q branch 200 of the receiving end 10, a complex conjugate term is generated in the output signal as expressed in equation 6, and an image is formed in the frequency domain from this.

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[0039] In Equation 7, H I RX (f) is taken out as a common D RX (f)=H Q RX (f) / H I RX (f), which is expressed as the ratio of the transfer function of the Q-branch channelization filter to the transfer function of the I-branch channelization filter. Therefore, the Q-branch channelization filter 220 is a filter (h) that shows the change of the I-branch channelization filter 120 and the Q-branch channelization filter 222 relative to the I-branch channelization filter 120 in the time domain, as shown in FIG. D RX (t)) 223. The above-mentioned Equation 7 can be summarized as follows:

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[0040] H I RX (f) is placed in common on the I branch and the Q branch, and H is placed only on the Q branch. D RX (f)=H Q RX (f) / H I RX We can consider a system model that includes a frequency-independent mismatch compensation unit 410 that includes a frequency-dependent mismatch compensation filter 322 with a transfer function of (f). D RX (n) can be expressed as shown in Figure 2. D RX (t) can be summarized as the following Equation 9, from which the mismatch between the baseband filters of the I and Q branches can be expressed.

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[0041] The mismatch between the I and Q branch baseband filters is D RX (f), and to compensate for this, H D RX To achieve this, add h to the I branch after AD sampling. D RX (t) discrete time version of h D RX (n) to compensate. h D RX Since (n) is an IIR (Infinite Impulse Response), FIR (Finite Impulse Response) approximation is applied for practical realization. In this case, the Q branch includes a tapped delay line 325, and h D RX(n) and balance it with the integer group delay D. This allows the receiver frequency dependent IQ mismatch to be compensated for.

[0042] The complex envelope transformation equations of the gain mismatch and phase mismatch induced by Equation 4 and Equation 5 can be obtained as Equation 10 below, and the matrix-form transformation equation of the IQ vector can be obtained as Equation 11. (z(t)=r BB (t))

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[0043] When Equation 10 is compiled into a determinant and its inverse matrix is ​​found, Equation 11 is obtained, and the gain mismatch and phase mismatch of the receiving LO can be compensated for.

[0044] Fig. 3 is a diagram showing an overview of the frequency-independent mismatch compensation unit 420. Referring to Fig. 3, the frequency-independent mismatch compensation unit 420 includes a first multiplier 421 connected to the I branch, a second multiplier 422 connected to the Q branch, a third multiplier 423 that multiplies the output signal of the first multiplier, and an adder 424 that adds the output of the second multiplier 422 and the output of the third multiplier 423 together.

[0045] The frequency-independent mismatch compensator 420 shown in FIG. 3 realizes Equation 11, and the gain of the first multiplier 421 included in the frequency-independent mismatch compensator 420 is g RX The gain of the second multiplier 422 corresponds to cos(1 / θ RX ) In one embodiment, it corresponds to cos(1 / θ RX ) using the Taylor series to obtain 1+(θ RX 2 ) / 2, which can be used as the gain of the second multiplier 422.

[0046] The gain of the third multiplier is tan(θ RX ) In one embodiment, tan(θ RX) using the Taylor series θ RX +(θ RX 2 ) / 3, which can be used as the gain of the third multiplier 423. By compensating for the frequency-independent mismatch in this way, the mismatch-compensated received signal r I and r Q can be obtained.

[0047] To compensate for the frequency-independent IQ mismatch, the IQ gain mismatch of the LO and mixer circuit, g RX , θ is the IQ phase mismatch RX In Equation 4, when the N-point normalized DFT of the discrete-time signal x(n) obtained by sampling the signal x(t) is X(k), (-N / 2≦k≦N / 2−1), the k-th frequency component X(k) contains not only the k-th frequency component R(k) of the original signal but also the complex conjugate R of the −k-th frequency component, which is the conjugate image. * (-k) is included.

[0048] X(k)=αR(k)+βR * (-k), X(-k)=αR(-k)+βR * Since (k), the time average of the product of X(k) and X(-k) is calculated as shown in the following formula 12.

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[0049] In Equation 12, for thermal noise and general signals, R(k) and R(-k) are uncorrelated and E[R(k)]=0, so Equation 12 can be summarized as Equation 13.

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[0050] The relationship between the expected values ​​of the power of the front and rear signals affected by the IQ mismatch is given by the following Equation 14.

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[0051] From the above two equations, the receiver LO, IQ signal gain mismatch and phase mismatch of the mixer, g RX , θ RX It is possible to obtain an index that estimates the

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[0052] That is, the gain mismatch component of the mixer, g RX and the phase mismatch component θ RX is the time average E[|xn| 2 ] is the time average of the square of the mixer output signal E[xn 2 ] can be obtained by normalizing the estimated index with any received signal r BB (t) and does not require a separate pilot signal, so it can be called blind estimation.

[0053] If the above-mentioned X(k) is an ergodic process, the time average E[X(k)X(-k)] must be the same as the ensemble average, and if the ensemble average calculation result of X(k)X(-k) is the same as the time average, as in Equation 16 below, it indicates that X(k) is an ergodic process. Therefore, the ensemble average can be used as an estimation index for the time average.

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[0054] The above-mentioned estimated index is an ensemble average of tone image correlations. However, the ensemble average of tone image correlations can be summarized as follows:

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[0055] The mean square of the signal provides an estimation index as shown in Equation 18 below.

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[0056] The power of the signals before and after contamination due to IQ mismatch satisfies the following formula 19.

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[0057] From the two equations, the time domain signal x after contamination by IQ mismatch is n g is an index that estimates IQ gain mismatch and phase mismatch using only RX , θ RX It is possible to obtain an index that estimates the

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[0058] As mentioned above, to compensate for the frequency dependent IQ mismatch, h D RX h, which is the discrete signal version of (t) D RX (n) must be estimated. The output signal y(t) is affected by the conjugate image in the channelization filter, and the D RX For discrete signal samples of a signal y(t), let y(n) be the signal y(t) and let Y(k) be the N-point normalized DFT of y(n). Y(k) satisfies the relationship in Equation 21, according to the continuous-time version of the equation above.

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[0059] As a pilot signal, if an SSB (Single-Side Band) multitone signal is input that fills half of the positive numbers in the frequency bin of the N-point DFT, R BB (-k)=R BB * (-k)=0, and the equation can be summarized as Equation 22.

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[0060] The time average of the tone image correlation Y(k)Y(-k) normalized by the average power of Y(k) is defined as m(k) as follows:

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[0061] In Equation 23, h D Since (n) is a real filter, the conjugate symmetry theorem H D (-k)=H D * By applying (k), H is obtained as shown in the following formula 24. D (kg RX e -jθRX can be calculated.

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[0062] Since we input a positive single-sideband (SSB) multitone, the m(k) calculation is only valid in the interval 1 ≤ k ≤ (N / 2-1) (E[|Y(k)| 2 ]=0 for -N / 2≦k≦0). However, the conjugate symmetry theorem H D (-k)=H D * (k) By H D Since the argument (arg) of (0) is 0, θ RX =-arg{Γ(0)}. However, since Γ(0) cannot be calculated directly, it is estimated by extrapolation of arg{Γ(k)}.

[0063] Also, the LO gain mismatch g RX is included in HD(k) and does not need to be calculated separately, but if there is a mismatch in the low-pass filter, |H D (0)|=1, so g RX =|Γ(0)|. Similarly, |Γ(k)| is extrapolated and estimated. H in the interval -N / 2≦k≦0 D (k) can be filled in using the conjugate symmetry theorem.

[0064] In this way, for the entire interval -N / 2≦k≦N / 2-1, H D Since (k) is estimated, we can take the N-point normalized IDFT and apply an appropriate window to obtain the compensation FIR filter hD(n), which can be expressed as Equation 25 below.

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[0065] That is, by calculating the ratio of the time average of the correlation calculation between the signal and the image to the average power of the signal for each discrete frequency position, it is possible to check the mismatch and phase mismatch in the filter for each frequency position. [Explanation of symbols]

[0066] 10 Receivers 100 I Branch 110 I Branch Mixer 120 I-Branch Channelizing Filter 123 Delay Element 200 Q Branch 210 Q Branch Mixer 220 Q-branch channelizing filter 223 Difference Filter 310 Frequency-dependent mismatch estimator 320 Frequency dependent mismatch compensation section 410 Frequency-independent mismatch estimator 420 Frequency-independent mismatch compensation section

Claims

1. an I branch including an I branch mixer that down-converts an RF signal (radio frequency signal) and outputs an I component, and an I branch channelization filter that separates a baseband signal from the output signal of the I branch mixer; a Q branch including a Q branch mixer that downconverts the RF signal and outputs a Q component, and a Q branch channelization filter that separates a baseband signal from the output signal of the Q branch mixer; a frequency-dependent mismatch estimator for calculating the frequency-dependent mismatch between the I branch and the Q branch; and a frequency-dependent mismatch compensator that compensates for frequency-dependent mismatch between the I branch and the Q branch based on the calculation result of the frequency-dependent mismatch estimator.

2. The frequency dependent mismatch compensator comprises:

2. The receiver of claim 1, further comprising a compensation filter in an I path having a transfer function corresponding to the transfer function of the Q branch channelization filter divided by the transfer function of the I branch channelization filter.

3. 3. The receiver of claim 2, wherein the frequency dependent mismatch compensator further includes a delay line in a Q path corresponding to a delay of the compensation filter.

4. the frequency dependent mismatch estimator, 2. The receiver according to claim 1, wherein a transfer function of a compensation filter that compensates for frequency-dependent mismatch is determined from a value obtained by normalizing a time average of a tone-image correlation of a signal output from the channelization filter by an average power of the signal output from the channelization filter.

5. When the value is m(k), The frequency dependent mismatch estimator [Equation 1] Satisfying the relationship, The argument and absolute value of Γ(k) are estimated by extrapolation. Ｄ (k) is found, The H Ｄ (k), a normalized N-point inverse DFT is performed, and a window is applied to obtain the impulse response h Ｄ 5. The receiver of claim 4, further comprising: (n) determining means for determining whether the first parameter is a first parameter or a second parameter.

6. the frequency dependent mismatch estimator, 5. The receiver of claim 4, wherein a time average of tone-image correlation is calculated using a pilot signal including an SSB multitone signal, normalized by the average power of the signal output from the channelization filter.

7. an I branch including an I branch mixer that down-converts an RF signal (radio frequency signal) and outputs an I component, and an I branch channelization filter that separates a baseband signal from the output signal of the I branch mixer; a Q branch including a Q branch mixer that downconverts the RF signal and outputs a Q component, and a Q branch channelization filter that separates a baseband signal from the output signal of the Q branch mixer; a frequency independent mismatch estimator for calculating a frequency independent mismatch between the I branch and the Q branch; and a frequency-independent mismatch compensator that compensates for frequency-dependent mismatch between the I branch and the Q branch based on the calculation result of the frequency-independent mismatch estimator.

8. The non-dependent inconsistency estimation unit 8. The receiver according to claim 7, wherein the gain mismatch and phase mismatch of the I branch mixer and the Q branch mixer are calculated from values ​​obtained by normalizing the time average of the power of the output signals of the I branch mixer and the Q branch mixer by the time average of the squares of the output signals of the I branch mixer and the Q branch mixer.

9. The non-dependent inconsistency estimation unit [Equation 2] (where X(k): output of the mixer in the discrete time domain, R(k): input of the mixer in the discrete time domain) to calculate the gain mismatch (g ＲＸ ), phase mismatch (θ ＲＸ 8. The receiver of claim 7, wherein the receiver estimates

10. The frequency independent mismatch compensation unit a first multiplier coupled to the I branch; a second multiplier coupled to the Q branch; a third multiplier for amplifying the output signal of the first multiplier; 10. The receiver of claim 9, further comprising: an adder that adds an output of the second multiplier and an output of the third multiplier.

11. The gain of the first multiplier is g ＲＸ Corresponding to, The gain of the second multiplier is cos(1 / θ ＲＸ ) corresponds to The gain of the third multiplier is tan(θ ＲＸ 11. The receiver of claim 10, wherein the receiver corresponds to

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