Transient thermal resistance measuring apparatus

Abstract

To improve accuracy of measurement of transient thermal resistance in a very short period of time immediately after stop of a heating current.SOLUTION: A transient thermal resistance measuring device 100 includes a heating power supply device 1 for supplying a heating current Ih to a semiconductor device 8 to be measured, a measuring power supply device 2 for supplying a measuring current Im to the semiconductor device 8 to be measured, a voltage measuring device 3 for measuring a voltage Vm between both ends of the semiconductor device 8 to be measured, and a constant current circuit 30 provided in a flow path of the measuring current Im. The constant current circuit 30 includes a transistor, a shunt resistor, a predetermined operational amplifier, and a predetermined voltage source. The predetermined operational amplifier controls the current value of the measurement current Im flowing between the main terminals of the transistor via the control terminal of the transistor so that the voltage across the shunt resistor becomes equal to the output voltage of the predetermined voltage source. The predetermined operational amplifier has a unity gain frequency greater than or equal to 4MHz and a slew rate greater than or equal to 2V / μ s.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present invention relates to a transient thermal resistance measuring device. [Background technology]

[0002] When carrying out the thermal design of a device incorporating semiconductor devices, it is important to understand the heat generation state of the semiconductor device. In particular, power semiconductor devices can get hot during use, and there is a possibility that they may exceed the guaranteed operating temperature range of the product, so accurate temperature control is required. It is extremely important to understand the temperature rise process of power semiconductor elements during very short-term operation due to pulse drive. In other words, it is necessary to accurately grasp the thermal characteristics that change over time during switching, and this is done by transient thermal resistance measurement.

[0003] In this measuring device, a predetermined minute current is first supplied to the semiconductor device to be measured, and the voltage across the semiconductor device before heat generation is measured. A predetermined current for generating heat is then supplied to the semiconductor device for a predetermined period of time, causing the semiconductor device to generate heat and its temperature to rise. A predetermined minute current is then supplied, and the voltage across the semiconductor device is measured. The amount of voltage change due to heat generation is then converted into an amount of temperature rise using the temperature dependency of the voltage of the semiconductor device, and the thermal resistance of the semiconductor device is calculated from the amount of temperature rise and the power required to generate heat. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] Japanese Patent Application Laid-Open No. 2013-113649 (see especially FIGS. 11 and 15 and paragraphs

[0066] to

[0068] of the specification) Summary of the Invention [Problem to be solved by the invention]

[0005] However, the above-mentioned conventional thermal resistance measuring device still has room for improvement in the following respects. Specifically, the transient thermal resistance characteristics of a semiconductor device can be measured during both the heat absorption process (temperature rise process) and the heat dissipation process (temperature fall process) of the semiconductor device. However, the change in thermal resistance in the initial temperature change of the semiconductor device can be measured more accurately when measured during the heat dissipation process of the semiconductor device than when measured during the heat absorption process. However, the above-mentioned conventional thermal resistance measuring device has a problem in that it measures the thermal resistance of the semiconductor device during the heat absorption process. On the other hand, when measuring the heat dissipation process of a semiconductor device, there is a problem in that, due to a sudden stop of the supply of a predetermined heating current (hereinafter sometimes simply referred to as "stop"), the transient thermal resistance cannot be accurately measured for a very short period of time immediately after the heating current is stopped, as will be described later.

[0006] The present invention has been made to solve the above-mentioned problems, and aims to provide a transient thermal resistance measuring device that can improve the measurement accuracy of transient thermal resistance during the very short period immediately after the heating current is stopped. [Means for solving the problem]

[0007] In order to achieve the above object, a transient thermal resistance measuring apparatus according to an aspect of the present disclosure includes a heating power supply that supplies a heating current to a semiconductor device under test for heating the device under test, a measuring power supply that supplies a minute DC measuring current to the semiconductor device under test, a voltage measuring apparatus that measures a voltage across the semiconductor device under test, and a constant current circuit provided in a flow path of the measuring current, the constant current circuit having a first main terminal, a second main terminal, and a control terminal, the measuring current flows between the first main terminal and the second main terminal, and a transistor whose current value changes in accordance with a potential difference of the control terminal with respect to a potential of the second main terminal, and one end of the transistor and a predetermined operational amplifier having an inverting input terminal to which the voltage at one end of the shunt resistor is input and a non-inverting input terminal to which the voltage at the positive terminal of a predetermined voltage source that outputs a current value setting voltage is input, and whose output is input to a control terminal of the transistor, wherein the first main terminal of the transistor and the other end of the shunt resistor are electrically connected to the flow path of the measurement current so that the constant current circuit is inserted in the flow path of the measurement current, and the negative terminal of the predetermined voltage source is electrically connected to the other end of the shunt resistor, and the predetermined operational amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

[0008] Furthermore, a transient thermal resistance measuring apparatus according to another aspect of the present disclosure includes a heating power supply that supplies a heating current to a semiconductor device under test to heat the device under test, a measuring power supply that is a constant current source that supplies a minute DC measuring current to the semiconductor device under test, a voltage measuring apparatus that measures the voltage across the semiconductor device under test, and a resistive element provided in the flow path of the measuring current. [Effects of the Invention]

[0009] The present invention has an effect of providing a transient thermal resistance measuring device that can improve the accuracy of measuring the transient thermal resistance in the very short time period immediately after the heating current is stopped. [Brief explanation of the drawings]

[0010] [Figure 1] FIG. 1 is a functional block diagram showing an example of the configuration of a transient thermal resistance measurement device according to the first embodiment of the present disclosure. [Figure 2] FIG. 2 is a circuit diagram showing an example of the configuration of the constant current circuit of FIG. [Figure 3] FIG. 3 is a circuit diagram showing an example of a circuit configuration in which the transistors of the constant current circuit of FIG. 2 are N-channel MOSFETs. [Figure 4] FIG. 4 is a circuit diagram showing an example of a circuit configuration in which the transistors of the constant current circuit of FIG. 2 are P-channel MOSFETs. [Figure 5A] FIG. 5A is a circuit diagram showing a method for measuring Vsd when the semiconductor device under test includes a MOSFET as a semiconductor element whose heat generation state is to be grasped. [Figure 5B] FIG. 5B is a circuit diagram showing a method for measuring Vds when the semiconductor device under test includes a MOSFET as a semiconductor element whose heat generation state is to be grasped. [Figure 5C] FIG. 5C is a circuit diagram showing a method for measuring Vth when the semiconductor device under test includes a MOSFET as a semiconductor element whose heat generation state is to be grasped. [Figure 5D] FIG. 5D is a circuit diagram showing a method for measuring Vce when the semiconductor device under test includes an IGBT as a semiconductor element whose heat generation state is to be grasped. [Figure 5E] FIG. 5E is a circuit diagram showing a method for measuring Vth when the semiconductor device under test includes an IGBT as a semiconductor element whose heat generation state is to be grasped. [Figure 5F] FIG. 5F is a circuit diagram showing a method for measuring the voltage across the terminals when the semiconductor device under test is a TEG including a semiconductor element whose heat generation state is to be grasped. [Figure 6] FIG. 6 is a flowchart showing the control contents of the control device and voltage measuring device of FIG. [Figure 7] FIG. 7 is a graph schematically showing the effect of the comparative example. [Figure 8] FIG. 8 is a graph schematically showing the effect of the transient thermal resistance measuring device of the first embodiment. [Figure 9] FIG. 9 is a waveform diagram showing an example of actual measurements of the effect of the comparative example. [Figure 10] FIG. 10 is a graph showing the transition of the bonding temperature obtained in the comparative example. [Figure 11] FIG. 11 is a waveform diagram showing an example of measurements of the effect of the transient thermal resistance measuring device of the first embodiment. [Figure 12] FIG. 12 is a graph showing the transition of the junction temperature obtained by the transient thermal resistance measurement device of the first embodiment. [Figure 13] FIG. 13 is a graph showing the transient thermal resistance characteristics obtained by the transient thermal resistance measurement devices of the comparative example and the first embodiment. [Figure 14] FIG. 14 is a circuit diagram showing an example of the configuration of an improved constant current circuit together with a circuit for simulation. [Figure 15] FIG. 15 is a waveform diagram showing the results of a simulation of the operation of a transient thermal resistance measuring device using an improved constant current circuit. [Figure 16] FIG. 16 is a graph showing the results of a study of the response speed required for a given operational amplifier. [Figure 17A] FIG. 17A is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 1 in FIG. [Figure 17B] FIG. 17B is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of the transient thermal resistance measurement device using the sample operational amplifier numbered 2 in FIG. [Figure 17C] FIG. 17C is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of the transient thermal resistance measurement device using the sample operational amplifier numbered 3 in FIG. [Figure 17D] FIG. 17D is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of the transient thermal resistance measurement device using the sample operational amplifier numbered 4 in FIG. [Figure 17E] FIG. 17E is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of the transient thermal resistance measurement device using the sample operational amplifier numbered 5 in FIG. [Figure 18] FIG. 18 is a diagram showing a resistance element in a transient thermal resistance measurement apparatus according to the second embodiment of the present disclosure. DETAILED DESCRIPTION OF THE INVENTION

[0011] (Findings that led to this disclosure) The present disclosure has been conceived based on the following findings.

[0012] <First finding> The specifications of semiconductor devices are generally defined when the temperature rise due to heat generation is in equilibrium. However, in a transient state of temperature rise due to heat generation, the semiconductor device itself absorbs the heat generated by the heat generation. Therefore, when a semiconductor device is used in a transient state of temperature rise due to heat generation, a large current can flow through the semiconductor device. This is the first finding.

[0013] <Second finding> Therefore, the transient thermal resistance of a semiconductor device that is used to pass a large current by utilizing such a transient state cannot be accurately measured unless the semiconductor device is heated with a large current (e.g., 20 A to 1000 A) that is within the allowable limit of the semiconductor device. Therefore, the present inventors attempted to measure the transient thermal resistance of a semiconductor device with such a large heating current. Measurement of transient thermal resistance is regulated, for example, by the standard JEDEC51-14 (JESD51-14). However, the following problems arose.

[0014] In applications where a large current is passed through a transient state as described above (hereinafter referred to as "transient state applications"), the voltage across the semiconductor device (junction temperature) must be accurately measured during a very short period of time immediately after the heating current is stopped (hereinafter referred to as the "specific transient thermal resistance measurement period"). This specific transient thermal resistance measurement period is, for example, within 10 μs. However, during this specific transient thermal resistance measurement period, the measurement current oscillates due to the sudden decrease in the measurement current caused by the stopping of the heating current. This causes oscillations in the voltage across the semiconductor device, making it difficult to accurately measure the transient thermal resistance. This oscillation in the measurement current was negligible when the heating current was small. This is the second finding.

[0015] <Third finding> The inventors presumed that this oscillation in the measurement current was caused by the parasitic impedance of the wiring supplying the heating current to the semiconductor device, and came up with the idea of ​​inserting a specified resistor in the measurement current supply path to dampen the oscillation in the measurement current. Furthermore, they presumed that this oscillation in the measurement current was also caused by the constant current control of the current source supplying the measurement current being unable to keep up with (respond) to a sudden decrease in the measurement current, and as a countermeasure, they configured the specified resistor in a constant current circuit. As a result, they were able to accurately measure transient thermal resistance during the specified transient thermal resistance measurement period. This is their third finding.

[0016] The present disclosure has been made based on the above first to third findings. Specific contents of the present disclosure will be described in detail below.

[0017] A transient thermal resistance measuring apparatus according to an aspect of the present disclosure includes a heating power supply that supplies a heating current to a semiconductor device under test for heating the device under test, a measuring power supply that supplies a minute DC measuring current to the semiconductor device under test, a voltage measuring apparatus that measures a voltage across the semiconductor device under test, and a constant current circuit provided in a flow path of the measuring current, the constant current circuit having a first main terminal, a second main terminal, and a control terminal, the measuring current flows between the first main terminal and the second main terminal, and the current value of the measuring current changes depending on the potential difference of the control terminal with respect to the potential of the second main terminal, and a transistor having one end connected to the second terminal of the transistor. a predetermined operational amplifier having an inverting input terminal to which the voltage at one end of the shunt resistor is input and a non-inverting input terminal to which the voltage at the positive terminal of a predetermined voltage source that outputs a current value setting voltage is input, and whose output is input to a control terminal of the transistor, wherein the first main terminal of the transistor and the other end of the shunt resistor are electrically connected to the flow path of the measurement current so that the constant current circuit is inserted in the flow path of the measurement current, and the negative terminal of the predetermined voltage source is electrically connected to the other end of the shunt resistor, and the predetermined operational amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

[0018] Here, if the transistor is a MOSFET, the first main terminal, the second main terminal, and the control terminal are the drain, the source, and the gate, respectively. If the transistor is a bipolar transistor, the first main terminal, the second main terminal, and the control terminal are the collector, the emitter, and the base, respectively. If the transistor is an IGBT, the first main terminal, the second main terminal, and the control terminal are the collector, the emitter, and the gate, respectively.

[0019] With this configuration, depending on the type of transistor (NPN type, PNP type, N-channel type, or P-channel type), the measurement current flows sequentially through the first main terminal of the transistor, the second main terminal of the transistor, and the shunt resistor, or through the shunt resistor, the second main terminal of the transistor, and the first main terminal of the transistor. If the transistor is an NPN or N-channel type, an increase in the measurement current increases the absolute value of the voltage (positive voltage) across the shunt resistor. As a result, if the voltage across one end of the shunt resistor exceeds the voltage at the positive terminal of the predetermined voltage source, the predetermined operational amplifier lowers the potential at the control terminal of the transistor. This decreases the measurement current, decreasing the absolute value of the voltage across the shunt resistor. When the voltage across one end of the shunt resistor becomes equal to the voltage at the positive terminal of the predetermined voltage source, an equilibrium state is reached. On the other hand, if the transistor is a PNP or P-channel type, an increase in the measurement current increases the absolute value of the voltage (negative voltage) across the shunt resistor. As a result, when the voltage at one end of the shunt resistor falls below the voltage at the positive terminal of the specified voltage source, the specified operational amplifier increases the potential at the control terminal of the transistor. This decreases the measurement current, decreasing the absolute value of the voltage across the shunt resistor. When the voltage at one end of the shunt resistor equals the voltage at the positive terminal of the specified voltage source, an equilibrium state is reached. In this equilibrium state, the measurement current equals the absolute value of the current setting voltage divided by the resistance of the shunt resistor (hereinafter referred to as the set current value). This ensures that the measurement current is always controlled to the set current value. Furthermore, because the specified operational amplifier has a unity-gain frequency of 4 MHz or higher and a slew rate of 2 V / μs or higher, the constant current circuit effectively follows the sudden decrease in the measurement current caused by the cessation of the heating current, effectively suppressing oscillations in the measurement current associated with the sudden decrease. As a result, the accuracy of transient thermal resistance measurements can be improved for the very short period immediately after the heating current is stopped.

[0020] The predetermined operational amplifier may have a unity gain frequency of 50 MHz or more and a slew rate of 20 V / μs or more.

[0021] With this configuration, the constant current circuit more effectively follows the sudden decrease in the measurement current caused by the cessation of the heating current, and oscillation of the measurement current caused by the sudden decrease in the measurement current is more effectively suppressed.

[0022] The constant current circuit may include a main operational amplifier that is the specified operational amplifier, and a differential amplifier circuit that includes a sub-operational amplifier, wherein the differential amplifier circuit amplifies the difference between the voltage at one end of the shunt resistor and the voltage at the other end of the shunt resistor using the sub-operational amplifier, and the output of the sub-op amplifier is input to the inverting input terminal of the main operational amplifier as the voltage at one end of the shunt resistor relative to the voltage at the other end of the shunt resistor, and the sub-op amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

[0023] According to this configuration, the voltage across the shunt resistor is accurately detected, and the measurement current is therefore controlled to an accurate current value.

[0024] The main operational amplifier and the auxiliary operational amplifier may each have a unity gain frequency of 50 MHz or more and a slew rate of 20 V / μs or more.

[0025] According to this configuration, the measurement current is controlled to an accurate current value, and oscillation of the measurement current due to a sudden decrease in the measurement current is more suitably suppressed.

[0026] Furthermore, a transient thermal resistance measuring apparatus according to another aspect of the present disclosure includes a heating power supply that supplies a heating current to a semiconductor device under test to heat the device under test, a measuring power supply that is a constant current source that supplies a minute DC measuring current to the semiconductor device under test, a voltage measuring apparatus that measures the voltage across the semiconductor device under test, and a resistive element provided in the flow path of the measuring current.

[0027] With this configuration, the resistance of the resistive element present in the measurement current flow path functions as a damper against sudden fluctuations in the measurement current, mitigating the sudden changes in the measurement current, thereby improving the accuracy of measuring the transient thermal resistance in the very short period immediately after the heating current is stopped.

[0028] The resistive element may have a resistance value of 2Ω or more and 200Ω or less.

[0029] This configuration effectively alleviates sudden changes in the measurement current, thereby effectively improving the accuracy of measuring the transient thermal resistance in the very short period immediately after the heating current is stopped.

[0030] Specific embodiments of the present disclosure will be described below with reference to the drawings. Note that, hereinafter, identical or corresponding elements throughout the drawings will be designated by the same reference numerals, and redundant description thereof will be omitted. Furthermore, since the following drawings are for explaining the present disclosure, elements unrelated to the present disclosure may be omitted, dimensions may be inaccurate due to exaggeration or the like, or may be simplified, and the shapes of corresponding elements may not match in multiple drawings. In the specification and claims of this application, "connecting" electrically or mechanically includes both "directly connecting" and "indirectly connecting." Furthermore, the present disclosure is not limited to the following embodiments.

[0031] <Definition> In this specification and claims, "measuring transient thermal resistance characteristics" includes both "directly measuring transient thermal resistance characteristics" and "indirectly measuring transient thermal resistance characteristics." Therefore, measuring the voltage Vm across a semiconductor device under test while supplying a minute measurement current Im to the semiconductor device under test is nothing other than "measuring transient thermal resistance characteristics."

[0032] For convenience, the voltage across the semiconductor device under test (DUT) when only the measurement current Im is supplied is called the "voltage across Vm," and the voltage across the semiconductor device under test (DUT) when the heating current Ih (and the measurement current Im) are supplied is called the "forward voltage VF."

[0033] Ignoring the minute measurement current Im, the "heating current Ih" is essentially the forward current IF for heating the semiconductor device under test (DUT). Since the term "forward current IF" is usually used when calculating transient thermal resistance characteristics, the "heating current Ih" is sometimes called the "forward current IF" when calculating transient thermal resistance characteristics.

[0034] (Embodiment 1)

[0035] [composition] 1 is a functional block diagram showing an example of the configuration of a transient thermal resistance measuring device 100 according to the first embodiment of the present disclosure. Referring to FIG. 1, the transient thermal resistance measuring device 100 includes a heating power supply device 1, a measurement power supply device 2, a voltage measuring device 3, a constant current circuit 30 as a resistor, and a control device 5.

[0036] The heating power supply 1 supplies a heating current Ih to a semiconductor device under test (hereinafter sometimes referred to as DUT) 8 to heat the DUT 8. The measurement power supply 2 supplies a measurement current Im of a predetermined minute DC current value to the DUT 8. The voltage measurement device 3 measures the voltage Vm across the DUT 8. The control device 5 controls the operations of the heating power supply 1, the measurement power supply 2, and the voltage measurement device 3, and calculates the junction temperature transition and transient thermal resistance characteristics. The constant current circuit 30 supplies the measurement current Im to the DUT 8. Im It is provided in the flow path of the.

[0037] The positive terminal of the heating power supply device 1 is connected to the positive terminal of the DUT 8 by a wire W1, and the negative terminal of the heating power supply device 1 is connected to the negative terminal of the DUT 8 by a wire W2. A backflow blocking diode 6 is provided in the middle of the wire W1. The heating power supply device 1 supplies a heating current Ih to the DUT 8 through the wires W1 and W2.

[0038] The positive terminal of measurement power supply device 2 is connected to the positive terminal of DUT 8 by wiring W3, and the negative terminal of measurement power supply device 2 is connected to the negative terminal of DUT 8 by wiring W4. A constant current circuit 30 is provided in the middle of wiring W3. A reverse current blocking diode 7 is also provided in the middle of wiring W3.

[0039] The measurement power supply device 2 passes a minute measurement current Im through the wires W3 and W4 to the DUT 8. Note that the negative terminal of the wire W3 and the positive terminal of the wire W4 may be connected to the wires W1 and W2 for the heating current Ih, respectively. However, this is not preferable because it increases the influence of the wiring impedance of the wires W1 and W2, thereby increasing the oscillation of the measurement current Im when the heating current Ih is stopped.

[0040] The positive input terminal of the voltage measuring device 3 is connected to the positive terminal of the DUT 8 by a wiring W5, and the negative input terminal of the voltage measuring device 3 is connected to the negative terminal of the DUT 8 by a wiring W6. The voltage measuring device 3 measures the voltage across the DUT 8 (hereinafter, sometimes simply referred to as "voltage across the DUT") through the wiring W5 and the wiring W6. VM The voltage measuring device 3 measures the voltage across the terminals, creates a data file D of the measured data of the voltage across the terminals, and sends this to the control device 5. The voltage measuring device 3 also controls the on / off of the switch section 1b of the heating power supply device 1 by a control signal Sc4.

[0041] The control device 5 uses control signals Sc1 to Sc3 to control the operations of the heating power supply device 1, the measurement power supply device 2, and the voltage measurement device 3. The control device 5 also calculates the junction temperature transition and transient thermal resistance characteristics using a data file D of the end-to-end voltage measurement data from the voltage measurement device 3.

[0042] Each of these components will be described in detail below.

[0043] <Heating power supply device 1> The heating power supply device 1 includes a heating power supply unit 1a and a switch unit 1b.

[0044] The heating power supply unit 1a is not particularly limited as long as it can output a predetermined large current (for example, 20 A to 1000 A) as the heating current Ih. The heating current Ih is a direct current. An example of the heating power supply unit 1a is a constant current source. The constant current source is, for example, configured as a switching power supply, but may also be configured as a linear power supply.

[0045] The switch unit 1b includes, for example, a switching element 11 connected to the positive output terminal of the heating power supply unit 1a and a diode 12 having an anode connected to the positive output terminal and a cathode connected to the negative output terminal of the heating power supply unit 1a. The switching element 11 is turned on and off by a control signal Sc4 from the voltage measurement device 3. When the switching element 11 is turned off while the switching element 11 and the heating power supply unit 1a are on and a heating current Ih is being output from the heating power supply device 1, the output current of the heating power supply unit 1a flows through the diode 12, thereby instantly stopping the output of the heating current Ih. Since the transient thermal resistance measurement device 100 measures the heat dissipation characteristics of the DUT 8, it is important that the output of the heating current Ih can be instantly stopped at the desired timing. Note that the switch unit 1b may be omitted from the heating power supply device 1. In this case, the supply of the heating current Ih is started or stopped by starting or stopping the heating power supply unit 1a, respectively.

[0046] <Measurement power supply device 2> The measurement power supply device 2 is not particularly limited as long as it can output a minute DC measurement current Im. The measurement power supply device 2 is configured, for example, as a constant voltage source or a constant current source, and in this example, the measurement power supply device 2 is configured as a constant voltage source that outputs a predetermined voltage Vo.

[0047] <Voltage measuring device 3> The voltage measuring device 3 measures the "voltage at both ends" VMand generates a data file D of voltage measurement data across the voltage measured at both ends, and turns on and off the switching element 11 of the switch unit 1b of the heating power supply device 1 using a control signal Sc4. The voltage measuring device 3 is configured, for example, by a computing unit having a processor and memory. A predetermined program for executing the functions of the voltage measuring device 3 is stored in the memory of this computing unit, and the voltage measuring device 3 is realized as a functional block by the processor reading and executing this predetermined program. In this case, the computing unit operates as the voltage measuring device 3. This computing unit can be configured, for example, by a computer, a personal computer, a microcontroller, an MPU, an FPGA (Field Programmable Gate Array), a PLC (Programmable Logic Controller), etc.

[0048] <Control device 5> The control device 5 controls the operation of the heating power supply device 1, the measurement power supply device 2, and the voltage measurement device 3 using control signals Sc1 to Sc3, respectively, and calculates the junction temperature transition and transient thermal resistance characteristics. The control device 5 is configured, for example, by a computing unit having a processor and memory. A predetermined program for executing the functions of the control device 5 is stored in the memory of this computing unit, and the control device 5 is realized as a functional block by the processor reading and executing this predetermined program. In this case, the computing unit operates as the control device 5. The computing unit can be configured, for example, by a computer, a personal computer, a microcontroller, an MPU, an FPGA (Field Programmable Gate Array), a PLC (Programmable Logic Controller), etc.

[0049] It should be noted that the functions of the elements disclosed herein can be performed using circuits or processing circuits, including general-purpose processors, special-purpose processors, integrated circuits, ASICs (Application Specific Integrated Circuits), conventional circuits, and / or combinations thereof, configured or programmed to perform the disclosed functions. A processor is considered a processing circuit or circuit because it includes transistors and other circuitry. In this disclosure, a "circuit" or "unit" is hardware that performs the recited functions or hardware that is programmed to perform the recited functions. The hardware may be hardware disclosed herein or other known hardware that is programmed or configured to perform the recited functions. Where hardware is a processor, which is considered a type of circuit, the "circuit" or "unit" is a combination of hardware and software, and software is used to configure the hardware and / or processor.

[0050] <Constant current circuit 30> FIG. 2 is a circuit diagram showing an example of the configuration of the constant current circuit 30 of FIG. 1. Referring to FIG. 2, the basic constant current circuit 30 includes a transistor Q as a variable resistance element, a shunt resistor SH, a predetermined operational amplifier OA, and a predetermined voltage source 10. The transistor Q includes a first main terminal Tf, a second main terminal Ts, and a control terminal Tc. A measurement current Im flows between the first main terminal Tf and the second main terminal Ts, and the magnitude of the measurement current Im varies depending on the potential difference between the second main terminal Ts and the control terminal Tc. The predetermined operational amplifier OA has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more. The basis for these numerical limitations will be explained in detail later. The predetermined voltage source 10 outputs a current value setting voltage Vsig.

[0051] One end of the shunt resistor SH is connected to the second main terminal Ts of the transistor Q. One end of the shunt resistor SH is connected to the inverting input terminal of the specified operational amplifier OA. As a result, the voltage Vsh at one end of the shunt resistor SH is input to the inverting input terminal of the specified operational amplifier OA. One end of the shunt resistor SH may be directly connected to the inverting input terminal of the specified operational amplifier OA, or may be indirectly connected (for example, via a preceding circuit) to the inverting input terminal of the specified operational amplifier OA. The positive terminal of the specified voltage source 10 is connected to the non-inverting input terminal of the specified operational amplifier OA. The positive terminal of the specified voltage source 10 may be directly connected to the non-inverting input terminal of the specified operational amplifier OA, or may be indirectly connected (for example, via an input resistor) to the non-inverting input terminal of the specified operational amplifier OA.

[0052] The first main terminal Tf of the transistor Q constitutes the first terminal T1 of the constant current circuit 30, the other end of the shunt resistor SH constitutes the second terminal T2 of the constant current circuit 30, and the negative terminal of the predetermined voltage source 10 constitutes the third terminal T3 of the constant current circuit 30. The constant current circuit 30 is inserted into the wiring W3 (the flow path of the measurement current Im) so that the first terminal T1 and the second terminal T2 are connected to the wiring W3. T3 is connected to a reference potential point (ground as a negative reference potential point or Vcc as a positive reference potential point). The negative terminal of the predetermined voltage source 10 (the third terminal T3 of the constant current circuit 30) is electrically connected to the other end of the shunt resistor SH (the second terminal T2 of the constant current circuit 30). Specifically, for example, as shown in FIG. 14, the negative terminal of the predetermined voltage source 10 is connected to ground, and the other end of the shunt resistor SH is essentially connected to ground by a differential amplifier circuit (measurement circuit 31). In the predetermined operational amplifier OA, the non-inverting input terminal and the inverting input terminal are imaginarily shorted. Therefore, the shunt resistor SH and the predetermined voltage source 10 are essentially connected in parallel with each other.

[0053] Here, the transistor Q is a type of variable resistance element, an active resistance element whose resistance value can be controlled. If the transistor Q is a MOSFET, the first main terminal Tf, the second main terminal Ts, and the control terminal Tc are the drain, source, and gate, respectively. If the transistor Q is a bipolar transistor, the first main terminal Tf, the second main terminal Ts, and the control terminal Tc are the collector, emitter, and base, respectively. If the transistor Q is an IGBT, the first main terminal Tf, the second main terminal Ts, and the control terminal Tc are the collector, emitter, and gate, respectively.

[0054] 3 is a circuit diagram showing an example of a circuit configuration in which the transistor Q of the constant current circuit 30 of FIG. 2 is an N-channel MOSFET. Referring to FIGS. 1 and 3, the first terminal T1 of the constant current circuit 30 is connected to the downstream end of the measurement current Im in the measurement power supply device 2 side portion of the wiring W3, and the second terminal T2 of the constant current circuit 30 is connected to the upstream end of the measurement current Im in the DUT 8 side portion of the wiring W3. In addition, the negative terminal of the predetermined voltage source 10 is electrically connected to the other end of the shunt resistor SH as described above. With this constant current circuit 30, the measurement current Im flows through the drain of the transistor Q. D (first main terminal Tf), source of transistor Q S The current flows through the second main terminal Ts and the shunt resistor SH in this order. If the measurement current Im increases, the absolute value of the voltage (positive voltage) across the shunt resistor SH increases. As a result, when the voltage at one end of the shunt resistor SH exceeds the voltage at the positive end of the predetermined voltage source 10, the predetermined operational amplifier OA lowers the potential at the gate G (control terminal Tc) of the transistor Q. Then, the measurement current Im decreases, and the absolute value of the voltage across the shunt resistor SH decreases. When the voltage at one end of the shunt resistor SH becomes equal to the voltage at the positive end of the predetermined voltage source 10, an equilibrium state is reached. In this equilibrium state, the measurement current Im increases in magnitude by the absolute value of the current setting voltage Vsig across the shunt resistor SH. SH Resistance value of In RshThe current value of the measurement current Im is always controlled to the set current value (hereinafter referred to as the set current value). As a result, the current value of the measurement current Im is always controlled to the set current value. The current value setting voltage Vsig is selected so that the current value obtained by dividing the current value setting voltage Vsig by the resistance value Rsh of the shunt resistor SH becomes the desired set current value.

[0055] The response speed of the constant current circuit 30 is essentially determined by the response speed of the specified operational amplifier OA. Regarding the unity gain frequency and slew rate, which are indicators of the operational amplifier's response speed, this specified operational amplifier OA has a unity gain frequency of 4 MHz or higher and a slew rate of 2 V / μs or higher. Therefore, the constant current circuit 30 can effectively follow the sudden decrease in the measurement current Im caused by the cessation of the heating current Ih, and effectively suppress oscillation of the measurement current Im caused by the sudden decrease. As a result, the accuracy of transient thermal resistance measurement can be improved for the very short period immediately after the cessation of the heating current Ih. The effects of NPN-type bipolar transistors and IGBTs are similar to those described above.

[0056] 4 is a circuit diagram showing an example of a circuit configuration in which the transistor Q of the constant current circuit 30 of FIG. 2 is a P-channel MOSFET. Referring to FIGS. 1 and 4, the second terminal T2 of the constant current circuit 30 is connected to the downstream end of the measurement current Im in the measurement power supply device 2 side portion of the wiring W3, and the first terminal T1 of the constant current circuit 30 is connected to the upstream end of the measurement current Im in the DUT 8 side portion of the wiring W3. In addition, the negative terminal of the predetermined voltage source 10 is electrically connected to the other end of the shunt resistor SH as described above. With this constant current circuit 30, the measurement current Im flows through the shunt resistor SH, the source of the transistor Q, and the D (second main terminal Ts), and the drain of transistor Q D(first main terminal Tf). If the measurement current Im increases, the absolute value of the voltage (negative voltage) across the shunt resistor SH increases. As a result, if the voltage at one end of the shunt resistor SH falls below the voltage at the positive end of the predetermined voltage source 10, the predetermined operational amplifier OA raises the potential of the gate G (control terminal Tc) of the transistor Q. Then, the measurement current Im decreases, and the absolute value of the voltage across the shunt resistor SH decreases. When the voltage across the shunt resistor SH becomes equal to the voltage at the positive end of the predetermined voltage source 10, an equilibrium state is reached. In this equilibrium state, the measurement current Im increases the absolute value of the current setting voltage Vsig across the shunt resistor. SH Resistance value of Rsh The measurement current Im is a current value (hereinafter referred to as the set current value) obtained by dividing the current value Vsig by the resistance value Rsh of the shunt resistor SH. This ensures that the current value of the measurement current Im is always controlled to the set current value. The current value setting voltage Vsig is selected so that the current value obtained by dividing the current value setting voltage Vsig by the resistance value Rsh of the shunt resistor SH becomes the desired set current value. Furthermore, since the predetermined operational amplifier OA has a unity gain frequency of 4 MHz or higher and a slew rate of 2 V / μs or higher, the constant current circuit 30 effectively follows the sudden decrease in the measurement current Im caused by the cessation of the heating current Ih, effectively suppressing oscillation of the measurement current Im due to the sudden decrease. As a result, the accuracy of transient thermal resistance measurement can be improved for the very short period immediately after the cessation of the heating current Ih. The effects of PNP-type bipolar transistors and IGBTs are similar to those described above.

[0057] <Semiconductor device under test (DUT) 8> DUT8 isThe DUT8 includes one or more semiconductor elements whose thermal state is to be grasped. Examples of such semiconductor elements include a diode, a MOSFET, an IGBT, and a bipolar transistor. In the first embodiment, the transient thermal resistance characteristics of the DUT8 during the heat dissipation process are measured. To grasp the heat generation state of these semiconductor elements, the DUT8 is heated by a heating current Ih (forward current IF), and then the heating current Ih is stopped. Thereafter, a minute measurement current Im is supplied to the DUT8 during the heat dissipation process of the DUT8, while measuring the voltage Vm across the DUT8. This voltage Vm across the DUT8 is converted into a junction temperature Tj based on its dependency on the junction temperature, and then converted into a transient thermal resistance θth(jc). Furthermore, the DUT8 is placed on, for example, a water-cooled heat sink for heat dissipation.

[0058] {Method of measuring various types of voltage Vm} The voltage Vm across the DUT 8 varies depending on the type and location of the semiconductor element whose heat generation state is to be determined, and these will be described below in order with reference to FIGS. 1 and 5A-5F.

[0059] <Diode forward voltage> Referring to FIG. 1, if the DUT 8 includes a diode as a semiconductor element whose heat generation state is to be monitored, a forward heating current Ih is supplied to the diode to heat it up, and a forward measurement current Im is supplied to the diode to measure the forward voltage of the diode as a voltage Vm across both ends.

[0060] <MOSFET Vsd> 5A is a circuit diagram showing a method for measuring Vsd when the DUT 8 includes a MOSFET as a semiconductor element for which the heat generation state is to be grasped. In FIG. 5A, an N-channel MOSFET is shown as an example of the MOSFET. Referring to FIG. 5A, the N-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the source and a negative voltage is applied to the drain. 2Then, a gate voltage negative with respect to the source is applied to the gate, turning the N-channel MOSFET off. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the body diode 8a of the N-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the source-drain voltage Vsd of the N-channel MOSFET, which is the forward voltage of the body diode 8a due to the measurement current Im, is measured as the voltage Vm across both ends.

[0061] On the other hand, when the MOSFET is a P-channel MOSFET (not shown), the P-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the drain and a negative voltage is applied to the source. 2 Then, a gate voltage that is positive with respect to the source is applied to the gate, turning the P-channel MOSFET off. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the body diode of the P-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the source-drain voltage Vsd of the P-channel MOSFET, which is the forward voltage of the body diode due to the measurement current Im, is measured as the voltage Vm across both ends.

[0062] MOSFET Vds 5B is a circuit diagram showing a method for measuring Vds when the DUT 8 includes a MOSFET as a semiconductor element for which the heat generation state is to be grasped. In FIG. 5B, an N-channel MOSFET is shown as an example of the MOSFET. Referring to FIG. 5B, the N-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the drain and a negative voltage is applied to the source. 2Then, a gate voltage that is positive with respect to the source is applied to the gate, turning the N-channel MOSFET on. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the N-channel (inversion layer) of the N-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the drain-source voltage Vds of the N-channel MOSFET due to the measurement current Im is measured as the voltage Vm across both ends.

[0063] On the other hand, when the MOSFET is a P-channel MOSFET (not shown), the P-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the source and a negative voltage is applied to the drain. 2 Then, a gate voltage that is negative with respect to the source is applied to the gate, turning the P-channel MOSFET on. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the P-channel (inversion layer) of the P-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the drain-source voltage Vds of the P-channel MOSFET due to the measurement current Im is measured as the voltage Vm across both ends.

[0064] <MOSFET Vth> 5C is a circuit diagram showing a method for measuring Vth when the DUT 8 includes a MOSFET as a semiconductor element for which the heat generation state is to be grasped. In FIG. 5C, an N-channel MOSFET is shown as an example of the MOSFET. Referring to FIG. 5C, the N-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the drain and a negative voltage is applied to the source. 2Then, the gate and drain are short-circuited, placing the N-channel MOSFET in a so-called "diode-connected" state. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the N-channel (inversion layer) of the N-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the threshold voltage Vth of the N-channel MOSFET due to the measurement current Im is measured as the voltage Vm across both ends.

[0065] On the other hand, when the MOSFET is a P-channel MOSFET (not shown), the P-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the source and a negative voltage is applied to the drain. 2 Then, the gate and drain are short-circuited, placing the P-channel MOSFET in a so-called "diode-connected" state. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the P-channel (inversion layer) of the P-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the threshold voltage Vth of the P-channel MOSFET due to the measurement current Im is measured as the voltage Vm across both ends.

[0066] <IGBT Vce> 5D is a circuit diagram showing a method for measuring Vce when the DUT 8 includes an IGBT as the semiconductor element whose heat generation state is to be grasped. In FIG. 5D, an N-channel IGBT is shown as an example of the IGBT. Referring to FIG. 5D, the N-channel IGBT is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the collector and a negative voltage is applied to the emitter. 2Then, a gate voltage that is positive with respect to the emitter is applied to the gate, turning the N-channel IGBT on. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the N-channel IGBT, heating the DUT 8. Next, the heating current Ih is stopped, and the collector-emitter voltage Vce of the N-channel IGBT due to the measurement current Im is measured as the voltage Vm across both ends.

[0067] On the other hand, when the IGBT is a P-channel IGBT (not shown), the P-channel IGBT is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the emitter and a negative voltage is applied to the collector. 2 Then, a gate voltage negative with respect to the emitter is applied to the gate, turning the P-channel IGBT on. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the P-channel IGBT, heating the DUT 8. Next, the heating current Ih is stopped, and the collector-emitter voltage Vce of the P-channel IGBT due to the measurement current Im is measured as the voltage Vm across both ends.

[0068] <IGBT Vth> 5E is a circuit diagram showing a method for measuring Vth when the semiconductor device under test includes an IGBT as a semiconductor element for which the heat generation state is to be grasped. In FIG. 5E, an N-channel IGBT is exemplified as the IGBT. Referring to FIG. 5E, the N-channel IGBT is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the collector and a negative voltage is applied to the emitter. 2Then, the gate and collector are short-circuited, placing the N-channel IGBT in a so-called "diode-connected" state. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the N-channel (inversion layer) of the N-channel IGBT, heating the DUT 8. Next, the heating current Ih is stopped, and the threshold voltage Vth of the N-channel IGBT due to the measurement current Im is measured as the voltage Vm across both ends.

[0069] On the other hand, when the IGBT is a P-channel IGBT (not shown), the P-channel IGBT is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the emitter and a negative voltage is applied to the collector. 2 Then, the gate and collector are short-circuited, placing the P-channel IGBT in a so-called "diode-connected" state. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the P-channel (inversion layer) of the P-channel IGBT, heating the DUT 8. Next, the heating current Ih is stopped, and the threshold voltage Vth of the P-channel IGBT due to the measurement current Im is measured as the voltage Vm across both ends.

[0070] <<Voltage across TEG>> 5F is a circuit diagram showing a method for measuring the voltage across the DUT8 when the DUT8 is a TEG (Test Elementary Group) including a semiconductor element whose heat generation state is to be ascertained. Referring to FIG. 5F, the TEG constituting the DUT8 includes an IGBT 8c, which is a semiconductor element whose heat generation state is to be ascertained, and a diode 8d is disposed near the IGBT. In this case, a heating current Ih is supplied to the IGBT 8c and a measurement current Im is supplied to the diode 8d to heat the DUT8. After the heating current Ih is stopped, the forward voltage of the diode 8d due to the measurement current Im is measured as the voltage across the DUT8. The same applies when the semiconductor element whose heat generation state is to be ascertained is a MOSFET.

[0071] In the first embodiment, the heat radiation characteristics (temperature drop characteristics) of the DUT 8 are measured to measure the transient thermal resistance, so the DUT 8 is placed on a heat sink.

[0072] {Junction temperature Tj of DUT8} As is clear from the above explanation, the voltage Vm across the DUT8, which is the voltage drop due to the measurement current Im of the semiconductor element whose heat generation state is to be grasped, includes Vsd, which is the forward voltage of the diode and the forward voltage of the body diode of the MOSFET, as well as forward drop voltages of the pn junction such as the threshold voltage Vth of the MOSFET and IGBT, and drop voltages of the channel (inversion layer) such as Vds, which is the drain-source voltage of the MOSFET, and Vce, which is the collector-emitter voltage of the IGBT. Therefore, hereinafter, for convenience, both the temperature obtained by converting the forward drop voltage of the pn junction into the temperature of the pn junction and the temperature obtained by converting the drop voltage of the channel (inversion layer) into the temperature of the channel (inversion layer) will be referred to as the "junction temperature Tj of the DUT8."

[0073] {Relationship to this disclosure} As is clear from the above explanation, the heating power supply 1 and measurement power supply 2 of the transient thermal resistance measuring device 100 of the present disclosure only serve to supply the heating current Ih and measurement current Im to the DUT 8, respectively, and therefore the present disclosure is not limited in any way by the form of the DUT 8.

[0074] [Operation] Next, the operation of the transient thermal resistance measuring device 100 configured as above will be described with reference to Fig. 1 and Fig. 6. This operation is performed by the control device 5 and the voltage measuring device 3 working together to control the transient thermal resistance measuring device 100. Fig. 6 is a flowchart showing the control contents of the control device 5 and the voltage measuring device 3 in Fig. 1.

[0075] 1 and 6, first, the control device 5 causes the heating power supply unit 1a of the heating power supply device 1 to start operating, and causes the measurement power supply device 2 to start supplying the measurement current Im (step S1). Specifically, the control device 5 outputs a control signal Sc1, which is an ON signal, to the heating power supply unit 1a of the heating power supply device 1. The heating power supply unit 1a then outputs a heating current to the switch unit 1b. This heating current flows through the diode 12 in the switch unit 1b and returns to the heating power supply unit 1a. The control device 5 also outputs a control signal Sc2, which is an ON signal, to the measurement power supply device 2. This causes the measurement power supply device 2 to start supplying the measurement current Im to the DUT 8. At this time, the constant current circuit 30 controls the measurement current Im to a desired set current value.

[0076] Next, the voltage measuring device 3 causes the switch unit 1b of the heating power supply device 1 to start supplying a heating current Ih (forward current IF), and also causes the voltage measuring device 3 to start measuring the forward voltage VF of the DUT 8 (step S2). Specifically, the voltage measuring device 3 outputs an ON signal control signal Sc4 to the switch unit 1b of the heating power supply device 1. This turns on the switching element 11 of the switch unit 1b, and the heating current from the heating power supply unit 1a flows split between the diode 12 and the switching element 11. The heating current Ih output from the switching element 11 is supplied to the DUT 8 via the wiring W1 and wiring W2. This heating current Ih is detected by a heating current sensor (not shown) and is feedback-controlled to a predetermined current value by the heating power supply unit 1a. In addition, the control device 5 outputs an ON signal control signal Sc3 to the voltage measuring device 3. This causes the voltage measuring device 3 to start measuring the forward voltage VF of the DUT 8.

[0077] Next, when a predetermined heating period has elapsed, the voltage measuring device 3 causes the switch unit 1b of the heating power supply device 1 to stop supplying the heating current Ih, and causes the control device 5 to stop the heating power supply unit 1a of the heating power supply device 1 (step S3). The "predetermined heating period" is determined as appropriate, taking into consideration the structure of the DUT 8, the measurement conditions for the transient thermal resistance characteristics, and the like. The "predetermined heating period" is also determined to be a time sufficient for the temperature of the DUT 8 to saturate. Specifically, when the predetermined heating period has elapsed, the voltage measuring device 3 outputs a control signal Sc4, which is an off signal, to the switch unit 1b of the heating power supply device 1, and notifies the control device 5 of this. This turns off the switching element 11 of the switch unit 1b, and stops the output of the heating current Ih. As a result, all of the heating current from the heating power supply unit 1a flows through the diode 12. Meanwhile, when the control device 5 receives the above notification from the voltage measuring device 3, it outputs a control signal Sc1, which is an off signal, to the heating power supply unit 1a of the heating power supply device 1. This causes the heating power supply unit 1a to stop outputting the heating current.

[0078] When the supply of the heating current Ih is stopped, a voltage drop occurs across the DUT 8 due to only the measurement current Im, and the voltage measuring device 3 begins to measure this voltage drop as the voltage Vm across the DUT 8 (step S4).

[0079] Next, when a predetermined measurement time has elapsed since the heating current Ih was stopped, the control device 5 causes the voltage measuring device 3 to acquire measurement data (step S5). Specifically, the control device 5 outputs a control signal Sc3, which is an OFF signal, to the voltage measuring device 3. As a result, the voltage measuring device 3 stops measuring the voltage Vm across the DUT 8 and creates a data file D of the forward voltage VF, the voltage Vm across the DUT 8, the heating current Ih (forward current IF), and the measurement current Im, and transmits this data file D to the control device 5. The control device 5 also outputs a control signal Sc2, which is an OFF signal, to the measurement power supply device 2. As a result, the measurement power supply device 2 stops supplying the measurement current Im. Note that the order in which the measurement of the voltage Vm across the DUT 8 and the supply of the measurement current Im are stopped is not particularly limited. The control device 5 then calculates the transient thermal resistance characteristics of the DUT 8 based on this data file D, in accordance with the user's operation of a user interface (not shown). A specific calculation procedure for the transient thermal resistance characteristics will be described later.

[0080] [Action and effect] The effects of the transient thermal resistance measuring apparatus 100 of the first embodiment will be described below.

[0081] <Transient thermal resistance characteristic measurement procedure> First, the procedure for measuring the transient thermal resistance characteristics will be briefly explained. As mentioned above, the measurement of the transient thermal resistance θth(jc) is specified in the standard: JEDEC51-14 (JESD51-14).

[0082] First, the change characteristic of the voltage Vm across the DUT 8 with respect to the junction temperature Tj of the DUT 8 (hereinafter, sometimes referred to as the dependency of the voltage Vm across the DUT 8 on the junction temperature Tj) is measured. This measurement is performed, for example, by placing the DUT 8 in a thermostatic chamber and changing the temperature while supplying a measurement current Im to the DUT 8 to change the voltage Vm across the DUT 8. As is widely known, the voltage Vm across the DUT 8 decreases almost linearly as the junction temperature Tj increases.

[0083] Next, the voltage Vm across the DUT8 is measured while changing the junction temperature Tj of the DUT8. In this embodiment, the voltage Vm across the DUT8 is measured while lowering the junction temperature Tj of the DUT8 by heat dissipation through cooling using a water-cooled heat sink. To achieve this, the DUT8 is first heated by a forward current IF (heating current Ih) for a "predetermined heating time." This heats the DUT8 so that its junction temperature Tj reaches a saturated state. Furthermore, the forward voltage VF of the DUT8 is measured during the "predetermined heating time." After the "predetermined heating time" has elapsed, heating of the DUT8 is stopped. Then, from the point at which the heating is stopped, the voltage Vm across the DUT8 is measured while the DUT8 is cooled by the heat sink.

[0084] Next, using the above-mentioned dependency of the voltage Vm across the DUT 8 on the junction temperature Tj, the measured time-series data of the voltage Vm across the DUT 8 is converted into the junction temperature Tj of the time-series data of the DUT 8. In this case, since the voltage Vm across the DUT 8 immediately after the forward current IF (heating current Ih) is stopped cannot be accurately measured due to the recovery characteristics of the DUT 8, the junction temperature Tj(0) at the time the forward current IF (heating current Ih) is stopped is estimated.

[0085] Next, the transient thermal resistance characteristics of the time-series data are calculated by dividing the junction temperature change (Tj(0)-Tj(t)) obtained by subtracting the junction temperature Tj of the time-series data of DUT 8 from this junction temperature Tj(0) by the power P required to heat DUT 8. The power P required to heat DUT 8 is calculated by integrating the heating current Ih (set current value) x forward voltage VF at every moment during the "predetermined heating time."

[0086] Through the above measurements and calculations, the transient thermal resistance characteristics of the DUT 8 are measured.

[0087] <Example> An example was carried out to confirm the effect of the transient thermal resistance measurement apparatus 100 of the first embodiment. The example is the transient thermal resistance measurement apparatus 100 described above. A comparative example was also carried out to compare with the example. The configuration of the comparative example is the same as that of the transient thermal resistance measurement apparatus 100, except that the constant current circuit 30 is not present.

[0088] {overview} FIG. 7 is a graph schematically illustrating the effect of the comparative example. FIG. 8 is a graph schematically illustrating the effect of the transient thermal resistance measurement apparatus 100 of the first embodiment. In FIGS. 7 and 8, the top graph shows a waveform diagram of the heating current Ih (forward current IF), the second graph from the top shows a waveform diagram of the measurement current Im, the third graph from the top shows waveform diagrams of the forward voltage VF and the voltage Vm across the DUT 8, the fourth graph from the top shows the transition of the junction temperature Tj, and the fifth graph from the top shows the transient thermal resistance characteristic (θth(jc)). Note that Tj(0) in the fourth graph from the top represents the estimated junction temperature immediately after the heating current is turned off. In each graph, the horizontal axis represents time. Furthermore, the vertical dashed line indicates the time when the heating current Ih is stopped.

[0089] 7, in the comparative example, when the heating current Ih is stopped, as shown in the area surrounded by the dotted circle, the measurement current Im oscillates and vibrates significantly immediately after the heating current Ih is stopped, which causes the voltage Vm across the DUT 8 to vibrate significantly. As a result, the portion of the transition of the junction temperature Tj immediately after the start of measurement fluctuates, making it impossible to measure this portion accurately, and making it impossible to measure the transient thermal resistance θth(jc) during the "specific transient thermal resistance measurement period," which is the very short period immediately after the heating current Ih is stopped.

[0090] 8, in the transient thermal resistance measurement apparatus 100 of the first embodiment, when the heating current Ih is stopped, the oscillation of the measurement current Im is suppressed immediately after the heating current Ih is stopped, as shown in the area surrounded by the dotted circle, and therefore the oscillation of the voltage Vm across the DUT 8 is also suppressed. This is because the oscillation of the measurement current Im is suppressed by the constant current circuit 30. As a result, the fluctuation in the portion immediately after the start of measurement of the transition of the junction temperature Tj is improved, improving the measurement accuracy of that portion, and making it possible to measure the transient thermal resistance θth(jc) up to a time region very close to the start of measurement during the "specific transient thermal resistance measurement period."

[0091] {Measurement example} In this measurement example, a fast response electronic load was used as the constant current circuit 30. A commercially available multi-function electronic load in constant current mode was used as this fast response electronic load.

[0092] 9 is a waveform diagram showing an example of actual measurements of the effect of the comparative example. Tj 11 is a waveform diagram showing an example of measurements of the effect of the transient thermal resistance measurement apparatus 100 of the first embodiment. FIG. 12 is a graph showing the transition of the junction temperature obtained by the transient thermal resistance measurement apparatus 100 of the first embodiment. Tj 13 is a graph showing the transition of the thermal resistance. FIG. 13 is a graph showing the transient thermal resistance characteristics obtained by the transient thermal resistance measurement apparatus 100 of the comparative example and the first embodiment. This transient thermal resistance characteristic includes contact thermal resistance. FIGS. 9 and 11 were created by tracing the image on the display screen of an oscilloscope, which is a display device for displaying the measurement results. FIGS. 10, 12, and 13 were created by tracing the image data of the graphs showing the measurement results.

[0093] 9 and 11, the upper waveform shows the waveform of the voltage Vm across the DUT 8, the middle waveform shows the waveform of the heating current Ih, and the lower waveform shows the waveform of the measurement current Im. Also, "T" in the figures indicates the time when the heating current Ih is stopped.

[0094] Referring to Figure 9, in the comparative example, as shown in the area surrounded by the dotted circle, immediately after the heating current Ih was stopped, the measurement current Im jumped out by about 160 mA, and this change did not converge for about 30 μs.

[0095] On the other hand, referring to FIG. 11, in the transient thermal resistance measuring device 100 of this embodiment 1, as shown in the area surrounded by the dotted circle, immediately after the heating current Ih was stopped, the measurement current Im jumped out by about 60 mA, but this change converged within about 10 μs.

[0096] Referring to FIG. 10, in the comparative example, the transition of the junction temperature Tj fluctuated in the region enclosed by the dotted circle, up to 1 ms after the start of measurement, and this region could not be measured accurately.

[0097] On the other hand, referring to FIG. 12, in the transient thermal resistance measuring apparatus 100 of the present embodiment 1, as shown in the area surrounded by the dotted circle, the transition of the junction temperature Tj hardly fluctuates even in the portion up to 1 ms after the start of measurement, and this portion was able to be measured accurately.

[0098] 13, the dashed line indicates the measurement results of the comparative example, and the solid line indicates the measurement results of the transient thermal resistance measurement apparatus 100 of the present embodiment 1. Referring to Fig. 13, as shown in the area surrounded by the dotted ellipse, in the comparative example, the measurement results were inaccurate during the "specific transient thermal resistance measurement period," and the transient thermal resistance θth(jc) could not be measured accurately, whereas in the transient thermal resistance measurement apparatus 100 of the present embodiment 1, the transient thermal resistance θth(jc) could be measured accurately during the "specific transient thermal resistance measurement period."

[0099] From the above-mentioned actual measurement examples, the transient thermal resistance measuring device 100 of the first embodiment can measure the transient thermal resistance during a specific transient thermal resistance measurement period (a very short period immediately after the heating current Ih is stopped). θth(jc) It has been demonstrated that the measurement accuracy can be improved.

[0100] [Improved constant current circuit] FIG. 14 is a circuit diagram showing an example of the configuration of an improved constant current circuit 30A obtained by improving the constant current circuit 30 of FIGS. 2 and 3, together with a circuit for simulation.

[0101] 14, an improved constant current circuit 30A includes a transistor Q, a measurement circuit 31, and a control circuit 32. The transistor Q is the same as the transistor Q in the constant current circuit 30 of FIGS. 2 and 3, and therefore a description thereof will be omitted.

[0102] The measurement circuit 31 is composed of a differential amplifier circuit including an auxiliary operational amplifier OA2. Specifically, in the measurement circuit 31, a shunt resistor SH has a resistance value of, for example, 2.5 Ω. One end of the shunt resistor SH is connected to the non-inverting input terminal of the auxiliary operational amplifier OA2 via a resistor element R8. Resistor elements R8 and R9 form a resistor voltage divider circuit, and the negative terminal of resistor element R9 is connected to ground. The other end of the shunt resistor SH is connected to the inverting input terminal of the auxiliary operational amplifier OA2 via resistor element R7, which is an input resistor. Resistor element R7 is connected to resistor element R10, which is a negative feedback resistor. Resistor elements R7 to R10 each have a resistance value of, for example, 1 kΩ. Therefore, the amplification factor of this differential amplifier circuit is 1. In other words, this differential amplifier circuit serves to accurately measure the voltage across the shunt resistor SH relative to the ground potential, and therefore to accurately measure the measurement current Im flowing through the shunt resistor SH. Resistor element R10 and resistor element R9 are connected in parallel with capacitors C2 and C3, respectively. Therefore, the parallel circuit of resistor element R10 and capacitor C2, and the parallel circuit of resistor element R9 and capacitor C3, each form a low-pass filter. Therefore, this differential amplifier circuit also serves to remove unwanted signals from the voltage across shunt resistor SH. This differential amplifier circuit also removes common-mode noise.

[0103] The portion of the measurement current Im flow path (wires W3 and W4) from the other end of shunt resistor SH to one output terminal (here, the negative terminal) of measurement power supply device 2 is connected to a reference potential point (here, ground (see FIG. 14)). Therefore, this differential amplifier circuit cancels the voltage drop due to reverse current blocking diode 7 and DUT 8 in the measurement current Im flow path, and as a result, as described above, this differential amplifier circuit plays a role in accurately measuring the voltage across shunt resistor SH with respect to the ground potential.

[0104] The control circuit 32 includes a main operational amplifier OA1, which is the predetermined operational amplifier OA in FIGS. 2 and 3. The control circuit 32 performs the same function as the predetermined operational amplifier OA in FIGS. 2 and 3. The positive terminal of a predetermined voltage source 10, which outputs a current value setting voltage Vsig, is connected to the non-inverting input terminal of the main operational amplifier OA1 via a resistor element R5, which serves as an input resistor. The negative terminal of the predetermined voltage source 10 is connected to ground. The output terminal of the auxiliary operational amplifier OA2 is connected to the inverting input terminal of the main operational amplifier OA1 via a resistor element R6, which serves as an input resistor. The output terminal of the main operational amplifier OA1 is connected to the gate G of an N-channel MOSFET transistor Q via a resistor element R2, which serves as a gate resistor. A snubber circuit consisting of a series circuit of resistor element R3 and capacitor C1 is provided between the inverting input terminal and output terminal of the main operational amplifier OA1. The power supply terminals of the main operational amplifier OA1 and the auxiliary operational amplifier OA2 are connected to a pair of constant voltage sources 33 and 34, respectively.

[0105] The gate G of the N-channel MOSFET is connected to the source S via a resistive element R1, which is a gate-source resistor. The main operational amplifier OA1 and the sub operational amplifier OA2 are composed of high-speed operational amplifiers. For example, a commercially available high-speed operational amplifier can be used as this high-speed operational amplifier.

[0106] In this improved constant current circuit 30A, the main operational amplifier OA1 receives the current setting voltage Vsig and the accurately measured voltage Vsh across the shunt resistor SH. The main operational amplifier OA1 operates to create an imaginary short circuit, resulting in Vsig-Vsh = 0. Then, a voltage obtained by multiplying Vsig-Vsh by the amplification factor of the main operational amplifier OA1 is output from the output terminal of the main operational amplifier OA1 and applied to the gate of the transistor Q. The transistor Q then generates a drain current (measurement current Im) corresponding to the output characteristics of the transistor Q. This allows the voltage Vsh across the shunt resistor SH to be accurately detected, thereby controlling the measurement current Im to an accurate current value. Furthermore, because the main operational amplifier OA1 and the auxiliary operational amplifier OA2 are high-speed operational amplifiers, the response speed of the improved constant current circuit 30A is improved, and sudden changes in the measurement current Im are more effectively suppressed. As a result, transient thermal resistance characteristics can be measured more accurately. The reason why the response speed of the improved constant current circuit 30A is improved by using a high-speed operational amplifier will be described later.

[0107] 14 shows a circuit configuration for simulating a transient thermal resistance measurement device 100 using an improved constant current circuit 30A. Therefore, in this circuit configuration, the switch unit 1b of the heating power supply 1 is omitted. Lt1 and Lt2 are assumed as the wiring impedances of the wiring through which the heating current Ih flows. In this simulation, the heating current Ih was 200 A, and the time from when the heating current Ih started to stop until it was completely stopped was 5 μs. The measurement current Im was 1 A.

[0108] FIG. 15 is a waveform diagram showing the results of a simulation of the operation of the transient thermal resistance measurement apparatus 100 using the improved constant current circuit 30A. FIG. 15 was created by tracing an image on the display screen of an oscilloscope, which is a device used to display the measurement results. In FIG. 15, the horizontal axis represents time, and the vertical axis represents the current values ​​of the heating current Ih and the measurement current Im. In FIG. 15, the upper waveform diagram represents the waveform of the heating current Ih, and the lower waveform diagram represents the waveform of the measurement current Im.

[0109] 15, in this simulation, the measurement current Im fluctuated as the heating current Ih was stopped. However, the fluctuations converged within 5 μs, from the start to the end of the heating current Ih stopping, and only an overshoot of approximately 1.04 A was observed. In other words, when the heating current Ih was stopped, the measurement current Im changed to a single pulse with a peak value of approximately 0.04 A (40 mA) and did not oscillate. Therefore, the fluctuations in the measurement current Im were drastically improved compared to when the improved constant current circuit 30A was not present.

[0110] Therefore, the above simulations confirmed that the transient thermal resistance measuring device 100 using the improved constant current circuit 30A can more effectively suppress sudden changes in the measurement current Im, and as a result, can measure transient thermal resistance characteristics more accurately.

[0111] In the above, an N-channel MOSFET is used as the transistor Q in the improved constant current circuit 30A, but a P-channel MOSFET, a bipolar transistor, or an IGBT may also be used as the transistor Q. Since creating the improved constant current circuit 30A in this manner is self-evident from the above description and Figures 2 to 4, a description thereof will be omitted.

[0112] [Consideration of the response speed required for a given op-amp OA] The response speed of the constant current circuit 30 is substantially determined by the response speed of the operational amplifier. This also applies to the improved constant current circuit 30A. Therefore, the inventors have studied the response speed required for a given operational amplifier OA. Generally, the unity gain frequency (f T ) and slew rate (SR) are used. In the simulation circuit of FIG. 14, the inventors have set the main operational amplifier OA1 and the sub operational amplifier OA2 at various unity gain frequencies (f T The simulation was performed by replacing the operational amplifier with one having a unity gain frequency (f) and a slew rate (SR), and the state of the measurement current Im when the heating current Ih was stopped was confirmed. TThe results of the simulation for each sample op-amp are summarized in Figure 16.

[0113] FIG. 16 is a graph showing the results of a study of the response speed required for a given operational amplifier OA. In the graph of FIG. 16, the horizontal axis is the unity gain frequency (f T ) [MHz], and the vertical axis shows the slew rate (SR) [V / μs] on a logarithmic scale. On this graph, each sample op-amp is plotted at its specified unity gain frequency (f T ) and slew rate (SR). Circles indicate plots of op-amps where the measurement current Im did not oscillate when the heating current Ih was stopped. Crosses indicate plots of op-amps where the measurement current Im oscillated when the heating current Ih was stopped. The area enclosed by the dashed dotted line is the unity gain frequency (f T ) is the range in which oscillation of the measurement current Im can be prevented when the heating current Ih is stopped, and the range surrounded by the two-dot chain line is the range in which oscillation of the measurement current Im can be prevented when the heating current Ih is stopped, when viewed from the slew rate (SR). It goes without saying that if oscillation of the measurement current Im can be prevented when the heating current Ih is stopped, the accuracy of measuring the transient thermal resistance can be improved in the very short period immediately after the heating current Ih is stopped.

[0114] Referring to Figure 16, the unity gain frequency (f T The higher the slew rate (SR), the better the results. To prevent the measurement current Im from oscillating when the heating current Ih is stopped, the operational amplifier must have a unity frequency (f) of approximately 4 MHz or higher. T) and a slew rate (SR) of 2 V / μs or more. The range that satisfies this requirement includes sample op-amps in which the measurement current Im oscillates when the heating current Ih is stopped, but this is unavoidable since each sample op-amp has specifications specialized for its respective application. To obtain an op-amp suitable for the specified op-amp OA of the present disclosure, it is sufficient to select, through simulation, experiment, calculation, etc., from commercially available op-amps that fall within this range, an op-amp that can prevent the measurement current Im from oscillating when the heating current Ih is stopped. Furthermore, op-amps that can prevent the measurement current Im from oscillating when the heating current Ih is stopped are those with a unity frequency (f) of 4 MHz or more. T ) and a dedicated operational amplifier with a slew rate (SR) of 2V / μs or greater.

[0115] Generally, the unity gain frequency (f T ) is said to be 50MHz or higher, so the unity gain frequency (f T The slew rate (SR) of the sample op-amp with a frequency of 100 kHz is 20 V / μs or more. The area surrounded by the dotted line is the unity gain frequency (f T ) is 50 MHz or higher and the slew rate (SR) is 20 V / μs or higher. All sample op-amps within this range are capable of preventing oscillation of the measurement current Im when the heating current Ih is stopped. It is estimated that commercially available high-speed op-amps fall within this range.

[0116] Plots numbered 1 to 5 are plots of representative sample operational amplifiers whose measurement current Im waveforms are shown in FIGS. 17A to 17F. FIG. 17A is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 1 in FIG. 16. FIG. 17B is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 2 in FIG. 16. FIG. 17C is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 3 in FIG. 16. FIG. 17D is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 4 in FIG. 16. FIG. 17E is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 5 in FIG. 16. In these figures, the horizontal axis represents time (μs) and the vertical axis represents current (A).

[0117] Referring to Figure 16, the sample op-amp number 1 is included in the area enclosed by the dotted line and has the second highest unity gain frequency (f T 17A, the measurement current Im exhibited a single pulse-like change with a peak value of about 0.036 A (36 mA) when the heating current Ih was stopped, and did not oscillate.

[0118] Referring to Figure 16, the sample operational amplifier number 2 is included in the area surrounded by the dotted line and has the highest unity gain frequency (f T 17B, ​​the measurement current Im exhibited a single pulse-like change with a peak value of about 0.038 A (38 mA) when the heating current Ih was stopped, and did not oscillate.

[0119] Referring to FIG. 16, the sample operational amplifier No. 3 is located at the lower limit of both the range enclosed by the dashed line and the range enclosed by the two-dot dashed line, and has a unity gain frequency (f T) and a slew rate (SR) of 2.5 V / μs. Referring to Fig. 17C, the measurement current Im exhibited a single pulse-like change with a peak value of about 0.049 A (49 mA) when the heating current Ih was stopped, and did not oscillate.

[0120] Referring to FIG. 16, the sample operational amplifier No. 4 is located at the lower limit of the range enclosed by the dashed line and at the bottom of the range enclosed by the dashed line, and has a unity gain frequency (f T 17D, the measurement current Im oscillated with an amplitude of approximately 0.030 A (30 mA) on one side when the heating current Ih was stopped.

[0121] Referring to FIG. 16, the sample operational amplifier No. 5 is located outside the range enclosed by the dashed line and the range enclosed by the two-dot chain line, and has the highest unity gain frequency (f T 17E, the measurement current Im oscillated with an amplitude of approximately 0.300 A (300 mA) on one side when the heating current Ih was stopped.

[0122] In this way, the state of the measurement current Im when the heating current Ih is stopped in the representative plots numbered 1 to 5 supports the conclusion regarding the consideration of the response speed required for the specified operational amplifier OA mentioned above. 30 Since the response speed of the constant current circuit 30 is substantially determined by the response speed of the operational amplifier, it goes without saying that the above considerations apply to the constant current circuit 30.

[0123] (Embodiment 2) FIG. 18 is a diagram showing a resistance element in a transient thermal resistance measurement apparatus according to the second embodiment of the present disclosure.

[0124] 18, in the transient thermal resistance measuring device according to the second embodiment, a resistance element 40 is arranged in the flow path of the measurement current Im instead of the constant current circuit 30 of the first embodiment. Also, the measurement power supply device 2 is configured as a constant current source. Other than this, the configuration of the transient thermal resistance measuring device according to the second embodiment is the same as that of the transient thermal resistance measuring device 100 according to the first embodiment.

[0125] The resistance element 40 has a resistance value of, for example, 2 Ω or more and 200 Ω or less. The resistance value of the resistance element 40 is determined according to the current value of the measurement current Im and the forward voltage VF of the DUT 8. The resistance value of the resistance element 40 is determined, for example, as follows. First, the measurement current Im is set to a current value at which internal heat generation of the DUT 8 is negligible. The measurement current Im is usually set so that internal heat generation of the DUT 8 is 1 W or less. For example, if the DUT 8 is a diode, and the measurement current Im is set to 0.1 A and the forward voltage VF of the diode (voltage Vm across the DUT 8) is, for example, 0.6 V, the internal heat generation of the DUT 8 will be 0.1 A × 0.6 V = 0.06 W (≦1 W).

[0126] The output voltage Vo (see FIG. 1) of the measurement power supply 2 depends on the specifications of the measurement power supply 2. For example, 2 When the output voltage Vo is 20 V and the measurement current Im is 0.1 A, the resistance value R of the resistive element 40 is given by the following formula:

[0127] R=(Vo-VF) / Im=(20V-0.6V) / 0.1A=194Ω According to the second embodiment, sudden changes in the measurement current Im are alleviated by the damping function of the resistance element 40. As a result, the measurement accuracy of the transient thermal resistance characteristics can be improved.

[0128] (Other embodiments) In the first or second embodiment, the heating power supply device 1 may supply a small heating current Ih.

[0129] In the first or second embodiment, the transient thermal resistance measuring apparatus 100 may measure the heat absorption characteristics (temperature rise characteristics) of the DUT 8.

[0130] In the first or second embodiment, the switch section 1b may be omitted from the heating power supply device 1, and the heating power supply device 1 may output the heating current Ih as a square-wave pulse current.

[0131] Many modifications and alternative embodiments will be apparent to those skilled in the art in light of the above description, and therefore the above description should be construed as illustrative only. [Industrial Applicability]

[0132] The transient thermal resistance measuring device of the present invention is useful as a transient thermal resistance measuring device that can improve the accuracy of measuring the transient thermal resistance in the very short period immediately after the heating current is stopped. [Explanation of symbols]

[0133] 1 Heating power supply 1a Heating power supply section 1b Switch section 2 Measurement power supply 3 Voltage measuring device 5. Control device 6,7 Reverse current blocking diode 8. Semiconductor device under test (DUT) 10. Predetermined voltage source 30 Constant current circuit 30A improved constant current circuit 40 Resistive element OA specified operational amplifier OA1 Main Op Amp OA2 Secondary operational amplifier

Claims

1. a heating power supply device for supplying a heating current to the semiconductor device under test for heating the device under test; a measurement power supply unit that supplies a minute DC measurement current to the semiconductor device under test; a voltage measuring device for measuring a voltage across both ends of the semiconductor device under test; a constant current circuit provided in a flow path of the measurement current, The constant current circuit is a transistor including a first main terminal, a second main terminal, and a control terminal, wherein the measurement current flows between the first main terminal and the second main terminal, and the current value of the measurement current changes in accordance with a potential difference of the control terminal relative to a potential of the second main terminal; a shunt resistor having one end electrically connected to the second terminal of the transistor; a predetermined operational amplifier having an inverting input terminal to which a voltage at one end of the shunt resistor is input, a non-inverting input terminal to which a voltage at a positive terminal of a predetermined voltage source that outputs a current value setting voltage is input, and an output of which is input to a control terminal of the transistor, the first main terminal of the transistor and the other end of the shunt resistor are electrically connected to the flow path of the measurement current so that the constant current circuit is interposed in the flow path of the measurement current, and a negative terminal of the predetermined voltage source is electrically connected to the other end of the shunt resistor, A transient thermal resistance measuring device, wherein the predetermined operational amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

2. 2. The transient thermal resistance measuring device according to claim 1, wherein the predetermined operational amplifier has a unity gain frequency of 50 MHz or more and a slew rate of 20 V / .mu.s or more.

3. the constant current circuit comprises a main operational amplifier which is the predetermined operational amplifier, and a differential amplifier circuit including a sub operational amplifier; the differential amplifier circuit amplifies a difference between a voltage at the one end of the shunt resistor and a voltage at the other end of the shunt resistor by the sub-op-amp; The output of the auxiliary operational amplifier is input to the inverting input terminal of the main operational amplifier as a voltage at one end of the shunt resistor relative to a voltage at the other end of the shunt resistor; and 2. The transient thermal resistance measuring device according to claim 1, wherein the auxiliary operational amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / .mu.s or more.

4. 4. The transient thermal resistance measuring device according to claim 3, wherein the main operational amplifier and the sub operational amplifier each have a unity gain frequency of 50 MHz or more and a slew rate of 20 V / .mu.s or more.

Images