Resonant converter and parameter calculation method thereof

Abstract

To provide a resonant converter and a parameter calculation method thereof.SOLUTION: Setting a preset operating condition, wherein the preset operating condition specifies that an operating frequency of the resonant converter is within a specific frequency range; calculating a plurality of sets of first resonance parameters in the specific frequency range, wherein the plurality of sets of first resonance parameters include a turn ratio of a transformer of the resonant converter; selecting a minimum first resonance parameter from the plurality of sets of first resonance parameters, wherein a current ripple flowing through a resonant inductor of the resonant converter has a minimum value; calculating a plurality of sets of second resonance parameters according to the turn ratio of the minimum first resonance parameter and the preset operating condition, and selecting one of the plurality of sets of second resonance parameters as a resonance parameter.SELECTED DRAWING: Figure 4A

Description

[Technical Field]

[0001] The present invention relates to a parameter calculation method for a resonant converter, and more particularly to a parameter calculation method for obtaining the resonance parameters of a resonant converter. [Background technology]

[0002] Dual active bridge (DAB) converters are generally used for high-power power conversion applications and are suitable for battery charging converters due to their wide voltage regulation range. Conventional dual active bridge converter configurations operate in a fixed frequency mode, and their circuit design is relatively simple. However, while dual active bridge converters can perform zero voltage switching (ZVS) operation during switch on operation using pulse width modulation control, they operate in hard switching during switch off operation, resulting in large current stress and large switching losses during switch off.

[0003] On the other hand, the circuit design of a dual active-bridge converter is usually performed by parameter design based on existing equations. However, this parameter design method makes it difficult to optimize the obtained parameters, resulting in excessive current ripple and a large resonant circuit size. Furthermore, non-optimized parameter design narrows the zero-voltage switching (ZVS) operating range of the dual active-bridge converter, making it difficult to achieve sufficient conversion efficiency. Summary of the Invention [Problem to be solved by the invention]

[0004] Therefore, in order to obtain more appropriate resonance parameters, it is an important task for the inventors to design a resonant converter and a parameter calculation method for the same that calculates parameters using an optimized process and highly accurate analysis. [Means for solving the problem]

[0005] In order to solve the above problems, the present invention provides a parameter calculation method for a resonant converter to overcome the problems of the prior art. Therefore, the parameter calculation method of the present invention is used to obtain resonance parameters of a resonant converter, and includes the steps of: setting a preset operating condition in which the operating frequency of the resonant converter is specified to be within a specific frequency range; calculating a plurality of sets of first resonance parameters including a turns ratio of a transformer of the resonant converter within the specific frequency range; selecting a minimum first resonance parameter from the plurality of sets of first resonance parameters that minimizes the current ripple flowing through a resonant inductor of the resonant converter; calculating a plurality of sets of second resonance parameters based on the turns ratio of the minimum first resonance parameter and the preset operating condition, and selecting one of the plurality of sets of second resonance parameters as the resonance parameter.

[0006] In order to solve the above problems, the present invention provides a resonant converter to overcome the drawbacks of the prior art, which includes a primary circuit connected to a DC power source, a resonant circuit including a plurality of transformers whose primary windings are connected to the primary circuit, and a secondary circuit including a plurality of sets of secondary switch circuits each having a first secondary bridge arm and a second secondary bridge arm connected in parallel, wherein the first secondary bridge arm and the second secondary bridge arm of the secondary switch circuit are respectively connected to both ends of the secondary windings of the plurality of transformers, and the secondary switch circuits are connected in parallel to a device. [Effects of the Invention]

[0007] The main object and effect of the present invention is to provide an optimized process for a resonant converter, which obtains more suitable resonant parameters through high-precision analysis, thereby reducing current ripple and improving the efficiency of the resonant converter.

[0008] In order to better understand the techniques, means, and advantages of the present invention which are contemplated to achieve the objects of the present invention, the objects and features of the present invention will be better understood by referring to the detailed description of the invention and the accompanying drawings, which are provided for reference and explanation only and are not intended to limit the invention. [Brief explanation of the drawings]

[0009] [Figure 1A] 1 is a circuit block diagram of a first embodiment of a resonant converter according to the present invention. [Figure 1B] FIG. 4 is a circuit block diagram of a second embodiment of a resonant converter according to the present invention. [Figure 1C] FIG. 10 is a circuit block diagram of a third embodiment of a resonant converter according to the present invention. [Figure 1D] FIG. 10 is a circuit block diagram of a fourth embodiment of the resonant converter of the present invention. [Figure 1E] FIG. 10 is a circuit block diagram of a fifth embodiment of the resonant converter of the present invention. [Figure 2A] 1 is a circuit block diagram of a first embodiment of a resonant transducer module according to the present invention. [Figure 2B] 1 is a simplified equivalent circuit diagram of a resonant conversion module according to a first embodiment of the present invention. [Figure 2C] 4 is a graph showing the power and voltage curves when the resonant transducer module of the present invention is used to charge and discharge a battery; [Figure 3A] FIG. 10 is a waveform diagram when the resonant conversion module of the present invention operates in a three-phase shift mode. [Figure 3B] FIG. 1 is a first waveform timing diagram when the resonant conversion module of the present invention operates in a three-phase shift mode. [Figure 3C] FIG. 2 is a second waveform timing diagram when the resonant conversion module of the present invention operates in three-phase shift mode. [Figure 3D] FIG. 3 is a third waveform timing diagram when the resonant conversion module of the present invention operates in three-phase shift mode. [Figure 3E]10 is a schematic waveform diagram when the resonant transducer module of the present invention operates in a single phase shift mode. FIG. [Figure 3F] 10 is a schematic waveform diagram when the resonant transducer module of the present invention operates in two-phase shift mode; [Figure 4A] 1 is a flowchart of a parameter calculation method for obtaining resonance parameters using a resonance conversion module of the present invention. [Figure 4B] FIG. 4 is a schematic diagram showing the numerical distribution of a second resonance parameter of the present invention. [Figure 5A] FIG. 10 is a first simulation waveform diagram when the resonant converter of the present invention uses the second resonant parameter of one embodiment. [Figure 5B] FIG. 10 is a second simulation waveform diagram when the resonant converter of the present invention uses the second resonant parameter of one embodiment. [Figure 5C] FIG. 10 is a third simulation waveform diagram when the resonant converter of the present invention uses the second resonant parameter of one embodiment. [Figure 6A] FIG. 10 is a first simulation waveform diagram when the resonant converter of the present invention uses the second resonant parameter of another embodiment. [Figure 6B] FIG. 10 is a second simulation waveform diagram when the resonant converter of the present invention uses the second resonant parameter of another embodiment. [Figure 6C] FIG. 10 is a third simulation waveform diagram when the resonant converter of the present invention uses the second resonant parameter of another embodiment. [Figure 7A] FIG. 2 is a system control block diagram of the resonant converter of the present invention. [Figure 7B] FIG. 2 is a block diagram of an operating frequency table reference control according to the present invention; [Figure 7C] FIG. 2 is a block diagram of a numerical value / time conversion according to the present invention. [Figure 7D] FIG. 2 is a block diagram of a dead time table reference control according to the present invention. [Figure 8] 3 is a flowchart of a closed-loop control of a resonant converter according to the present invention. [Figure 9A]FIG. 10 is a zero voltage switching region distribution diagram obtained when the resonant converter is substituted with the first implementation parameters using a simulation program. [Figure 9B] FIG. 10 is a zero voltage switching region distribution diagram obtained when the resonant converter is substituted with the second implementation parameters using a simulation program. [Figure 9C] FIG. 10 is a zero voltage switching region distribution diagram obtained when the resonant converter is substituted with the third implementation parameters using a simulation program. [Figure 10A] FIG. 10 is a circuit block diagram of a second embodiment of the resonant transducer module of the present invention. [Figure 10B] FIG. 10 is a simplified equivalent circuit diagram of a resonant conversion module according to a second embodiment of the present invention. [Figure 11A] FIG. 10 is a control block diagram of a closed-loop system for battery charging using a resonant conversion module according to a second embodiment of the present invention. [Figure 11B] FIG. 10 is a control block diagram of a closed-loop system for battery discharge using a resonant conversion module according to a second embodiment of the present invention. DETAILED DESCRIPTION OF THE INVENTION

[0010] The technical contents and detailed description of the present invention will be explained below with reference to the drawings.

[0011] The converter and its parameter calculation method described in this invention will be described using the circuit configuration of a dual active bridge (DAB) converter (hereinafter referred to as a DAB converter) or a series resonant dual active bridge (SR-DAB) converter (hereinafter referred to as an SR-DAB converter) as an example. However, while the technical content of this invention is suitable for DAB converters and SR-DAB converters, it is also applicable to resonant converters such as LLC and LC, and is not limited to these. Therefore, unless a specific converter is explicitly described, the term "resonant converter 100" in this specification includes resonant converters such as DAB converters, SR-DAB converters, LLC, and LC.

[0012] As shown in FIG. 1A, in the circuit block diagram of the first embodiment of the resonant converter of the present invention, the resonant converter 100 is composed of one or more resonant conversion modules (1A, 1B). The resonant conversion modules (1A, 1B) include a primary-side circuit A, a resonant circuit B, and a secondary-side circuit C. The primary-side circuit A is connected to a DC voltage Vdc, and the resonant circuit B includes multiple sets of resonant tanks (which may actually include a resonant inductor and a resonant capacitor, but FIGS. 1A and 1B show only a single resonant inductor as a representative example) and a transformer T. Each transformer T has a primary-side winding Wp and a secondary-side winding Ws, and the primary-side winding Wp is connected to the primary-side circuit A. The secondary-side circuit C includes multiple capacitors Co and multiple sets of secondary-side switch circuits 5. Each secondary-side switch circuit 5 includes a first secondary-side bridge arm 52 and a second secondary-side bridge arm 54 connected in parallel with the capacitor Co. The first secondary-side bridge arm 52 and the second secondary-side bridge arm 54 of the secondary-side switch circuit 5 are connected to both ends of the secondary-side winding Ws, and the secondary-side switch circuit 5 and the capacitor Co are connected in parallel to the device 200. Note that DAB converters and SR-DAB converters are capable of high-power conversion and the output voltage Vo can be adjusted over a wide range, making them particularly suitable for charging and discharging systems for electric vehicles. Therefore, the device 200 is preferably, but not limited to, a battery for an electric vehicle.

[0013] 1A, the primary-side circuit A includes a capacitor C1 and multiple switching circuits 4, each connected in parallel to the DC voltage Vdc. The capacitor C1 is primarily a voltage stabilizing or filtering element and can be replaced with any device (e.g., a filter) that has a stabilizing or filtering function. The switching circuits 4 include a first switching bridge arm 42 and a second switching bridge arm 44, which are connected in parallel. The first switching bridge arm 42 includes a first switching switch Q1 and a second switching switch Q2 connected in series, and the second switching bridge arm 44 includes a third switching switch Q3 and a fourth switching switch Q4 connected in series. A node Pp1 between the first changeover switch Q1 and the second changeover switch Q2 is connected to one end of the primary winding Wp, and a node Pp2 between the third changeover switch Q3 and the fourth changeover switch Q4 is connected to the other end of the primary winding Wp. A resonant tank (shown here as a single resonant inductor Lr1) is connected between one end of the primary winding Wp and the first switching bridge arm 42 or between the other end of the primary winding Wp and the second switching bridge arm 44.

[0014] Similarly, the first secondary-side bridge arm 52 includes a first secondary-side switch Qs1 and a second secondary-side switch Qs2, and the second secondary-side bridge arm 54 includes a third secondary-side switch Qs3 and a fourth secondary-side switch Qs4. A node Ps1 between the first secondary-side switch Qs1 and the second secondary-side switch Qs2 is connected to one end of the secondary-side winding Ws, and a node Ps2 between the third secondary-side switch Qs3 and the fourth secondary-side switch Qs4 is connected to the other end of the secondary-side winding Ws. Note that the configuration shown in FIG. 1A is an example of a DAB converter configuration, and the resonant tank is shown as only a resonant inductor Lr1. However, this is not limited to a single resonant inductor Lr1 and can be adjusted depending on the type of converter, and detailed description thereof will be omitted in this specification. In the embodiment, the turns ratio between the primary winding Wp and the secondary winding Ws is shown as n:1, but this is not limited to this and can be any ratio applicable to the resonant converter 100.

[0015] Furthermore, the controller 6 provides a pulse-width modulation signal PWM to control the transfer switches Q1-Q4 and the secondary-side switches Qs1-Qs4 to convert the DC voltage Vdc to the voltage Vo. Since the DAB converter and the SR-DAB converter have a bidirectional conversion function, the controller 6 can also provide a pulse-width modulation signal PWM to control the resonant converter 100 to convert the voltage Vo to the DC voltage Vdc. The resonant conversion modules (1A, 1B) shown in FIG. 1A are connected in parallel, so the resonant converter 100 can perform high-output power conversion. In the embodiment, the controller 6 can operate the resonant converter 100 in a single-phase shift (SPS) mode, a two-phase shift (DPS) mode, and a three-phase shift (TPS) mode. That is, the controller 6 can perform phase-shift control of the transfer switches Q1-Q4 or the secondary-side switches Qs1-Qs4, or even phase-shift control of both the transfer switches Q1-Q4 and the secondary-side switches Qs1-Qs4. Furthermore, the controller 6 also has a frequency variation function (that is, adjustment of the period T), which will be described later, and therefore will not be described in detail here.

[0016] Meanwhile, in the embodiment, the controller 6 may be a digital signal processor (DSP), but is not limited thereto. Any controller capable of controlling the resonant converter 100 is included in this embodiment. The resonant conversion modules (1A, 1B) shown in FIG. 1A can also be controlled using an interleaved control technique, which can reduce the current stress of the input filter caused by switching power supply (SPS) control. This is because the interleaved control has a ripple current canceling effect, which reduces the ripple current and thereby reduces the size of the filter element. Furthermore, in the topology of FIG. 1A, the resonant converter 100 mainly includes two resonant conversion modules (1A, 1B) configured in parallel (both the primary and secondary sides are connected in parallel), which enables conversion of high input voltages and even conversion of medium-voltage (MV, generally over 1000V) DC voltages Vdc and supply of high output power.

[0017] FIG. 1B is a circuit block diagram of a second embodiment of a resonant converter according to the present invention, and will be described with reference to both FIG. 1A and FIG. 1B. The difference between FIG. 1B and FIG. 1A is that the switching circuit 4 is connected in series to the DC voltage Vdc, and the primary side of the resonant converter 100 is configured in series and the secondary side is configured in parallel. In this embodiment, the secondary-side switch circuit 5 shown in FIG. 1B can be configured similarly to the primary side; that is, the secondary side can also be configured in series, parallel, or even as a single output (for example, when connected to a 400V / 800V battery). The same applies to the secondary-side switch circuit 5 in FIG. 1A, and a detailed description thereof will be omitted here.

[0018] Furthermore, because the primary side is configured in series, the DC voltage Vdc (which may be, but is not limited to, 800V) received by the resonant converter 100 is divided in half (Vdc / 2, i.e., 400V) by the two input-side capacitors C1. This allows the withstand voltage specifications of the changeover switches Q1 to Q4 to be reduced; for example, GaN elements with a withstand voltage of approximately 650V can be used instead of the changeover switches Q1 to Q4 with a withstand voltage of 800V or more. Please refer to FIG. 1A for other connection relationships and the effects obtained. A detailed description will be omitted here.

[0019] FIG. 1C is a circuit block diagram of a third embodiment of a resonant converter of the present invention, and will be described with reference to FIGS. 1A to 1C. FIG. 1C differs from FIG. 1A in that the primary circuit A includes a pair of switching circuits 4, which include a first switching bridge arm 42 and a second switching bridge arm 44 connected in parallel. The primary windings Wp of the transformer T are connected in series to form a primary winding series circuit. A node Pp1 between the first switch Q1 and the second switch Q2 is connected to one end of the primary winding series circuit, and a node Pp2 between the third switch Q3 and the fourth switch Q4 is connected to the other end of the primary winding series circuit.

[0020] Furthermore, the resonant converter 100 shown in FIG. 1C is configured as a single H-bridge resonant converter module 1, combining two resonant converter modules (1A, 1B). The two primary windings Wp are connected in series. Similarly, the secondary side can be configured in series, parallel, or with a single output. In FIG. 1C, the voltage across the transformer T is shared between the two primary windings Wp, halving the voltage across each primary winding Wp. This allows for a small turn ratio between the primary winding Wp and the secondary winding Ws (here, m:1, where m is an arbitrary value greater than 0). For this reason, the configuration of FIG. 1C is particularly suitable for high-voltage to low-voltage conversion applications (e.g., 400V to 12V conversion). For other connection relationships and the resulting effects, please also refer to FIG. 1A. Detailed descriptions are omitted here.

[0021] FIG. 1D is a circuit block diagram of a fourth embodiment of a resonant converter according to the present invention, and will be described with reference to FIGS. 1A to 1D. The difference between FIG. 1D and FIG. 1A is that the resonant inductor Lr1 of the resonant circuit B forms a common core structure M. Furthermore, the dotted ends (polarity terminals) of the resonant inductor Lr1 are located at the same position. By forming the two resonant inductors Lr1 into the common core structure M, positive and negative currents are connected in parallel, reducing current ripple. Furthermore, the reduced current ripple allows the core size to be reduced (compared to two independent cores). For other connection relationships and the resulting effects, please also refer to FIG. 1A. Detailed descriptions will be omitted here.

[0022] FIG. 1E is a circuit block diagram of a fifth embodiment of a resonant converter according to the present invention, and will be described with reference to FIGS. 1A to 1E. The difference between FIG. 1E and FIG. 1C is that the switching circuit 4 includes a third switching bridge arm 46 in parallel with the first switching bridge arm 42. The third switching bridge arm 46 includes a fifth changeover switch Q5 and a sixth changeover switch Q6 connected in series. The transformer T has a series-connected primary winding and a center tap terminal Pc between the two primary windings Wp. A node Pp3 between the fifth changeover switch Q5 and the sixth changeover switch Q6 is connected to the center tap terminal Pc. Because the primary winding Wp of the transformer T has a center tap structure, the maximum voltage of the two primary windings Wp can reach Vdc, making it suitable for applications with a small difference in turns ratio (e.g., 1:1). For other connections and the resulting effects, please also refer to FIG. 1C. Detailed descriptions are omitted here.

[0023] FIG. 2A is a circuit block diagram of a first embodiment of a resonant conversion module according to the present invention, and will be described with reference to FIGS. 1A to 2A. The resonant conversion module 1 shown in FIG. 2A primarily shows a series resonant dual active bridge circuit configuration, including a primary-side resonant tank and a secondary-side resonant tank. The primary-side resonant tank includes a resonant inductor Lr1 and a resonant capacitor Cr1, while the secondary-side resonant tank includes a resonant inductor Lr2 and a resonant capacitor Cr2. The detailed circuit configuration and connection relationships are similar to those of the resonant conversion module (1A, 1B) shown in FIG. 1A, and therefore will not be described here. The resonant circuit of the resonant conversion module 1 includes resonant capacitors Cr1 and Cr2, which resonate with the resonant inductors Lr1 and Lr2, respectively. As a result, the resonant capacitors Cr1 and Cr2 resonate with the resonant inductors Lr1 and Lr2, causing the inductor current Il to vary in a piecewise sinusoidal manner. In another example, if the resonant capacitors Cr1 and Cr2 are extremely large, they have no resonant effect and function only as DC blocking capacitors (blocking capacitors). In this case, the inductor current Il rises or falls in a diagonal linear fashion. Note that in one example, since the circuit configuration of FIG. 2A is composed of only a single resonant conversion module 1, this single resonant conversion module 1 can be considered as the resonant converter 100, and the resonant conversion module 1 described below can also be considered in the same way.

[0024] FIG. 2B is a simplified equivalent circuit diagram of a resonant conversion module according to a first embodiment of the present invention, and will be described with reference to FIGS. 1A to 2B. In FIG. 2B, the voltage between nodes Pp1 and Pp2 shown in FIG. 2A is equivalent to a voltage source (voltage) Vpp, and the voltage between nodes Ps1 and Ps2 is equivalent to a voltage source (voltage) Vss. Furthermore, the equivalent inductor of the resonant conversion module 1 is Le, and the equivalent capacitor is Ce. Under these conditions, in the above-mentioned three-phase shift mode, the voltages Vpp and Vss lead or lag due to the phase shift, resulting in improved dynamic performance of the circuit against load fluctuations and a shorter adjustment time.

[0025] FIG. 2C is a graph showing characteristic curves of power and voltage related to charging and discharging a battery using the resonant conversion module of the present invention. This will be explained with reference to FIGS. 1A to 2C. The controller 6 of the present invention controls the resonant conversion module 1 to supply a constant voltage Vo to the device 200 connected downstream. However, since the resonant conversion module 1 has the ability to adjust the voltage Vo, it is particularly suited to applications in which it is connected to a battery for charging and discharging. The following explanation of the present invention also mainly assumes that the device 200 is a battery. Specifically, FIG. 2C is a curve showing the relationship between the power Po and voltage Vo shown in FIG. 2A. When the voltage Vo reaches V1 (e.g., 200 V or higher), the controller 6 sets the resonant conversion module 1 to a constant current charging mode CC (constant current mode), and the power Po starts to increase from Pmin (e.g., 3.3 kW). Next, when the voltage Vo reaches V2 (e.g., 280 V), the controller 6 sets the charging and discharging mode to a constant power charging mode CP (constant power mode), and the power Po is maintained at Pmax (e.g., 6.6 kW). Furthermore, when the voltage Vo reaches V3 (for example, 460 V), the controller 6 can set the charging / discharging mode of the battery by the resonant conversion module 1 to a constant voltage charging mode (CV). As the battery is nearly fully charged, the output power begins to gradually decrease.

[0026] FIG. 3A is a waveform diagram of the resonant conversion module of the present invention when it operates in three-phase shift mode. This will be explained with reference to FIGS. 1A to 3A. In FIG. 3A, Vpp is the voltage waveform between nodes Pp1 and Pp2 when the changeover switches Q1 to Q4 operate in the switching circuit 4 (see FIG. 2A), and Vss is the voltage waveform between nodes Ps1 and Ps2 when the secondary-side switches Qs1 to Qs4 operate in the secondary-side switch circuit 5 (see FIG. 2A). Specifically, FIG. 3A shows waveforms when the resonant conversion module 1 shown in FIG. 2A operates in three-phase shift (TPS) mode. In this case, the controller 6 controls the switching circuit 4 to perform a phase shift, and the phase shift amount D1 can be set within a range of 0 to 0.5 (the phase shift of the changeover switches Q1 to Q4 changes the waveform of the voltage Vpp in a manner similar to an increase or decrease in the duty ratio). Similarly, the controller 6 controls the secondary-side switch circuit 5 to perform a phase shift, and the phase shift amount D2 can also be set within a range of 0 to 0.5. As a result, the voltages Vpp and Vss increase or decrease according to the phase shift amount. Furthermore, the controller 6 can also perform a phase shift by controlling both the switching circuit 4 and the secondary-side switch circuit 5, and the phase shift amount ψ (third phase shift amount) can be set within the range of -T / 2 to T / 2 (where T is the period, which corresponds to the operating frequency of the resonant conversion module 1). In this case, the voltages Vpp and Vss operate out of phase with each other. In addition to operation in three-phase shift mode, the present invention also allows for frequency variation, unlike conventional fixed frequency technology (where the period is fixed between 0 and T), and this will be discussed later.

[0027] Specifically, based on the operation of the three-phase shift (TPS) mode described above, the three waveform timing diagrams shown in Figures 3B to 3D can be summarized. In Region I of Figure 3B, the phase shift amount D1 is set to be larger than the phase shift amount D2 by a certain amount. This results in the waveforms of the voltages Vpp and Vss, as well as the waveforms of the inductor voltage Vl and the inductor current Il (see Figure 2A). This enables the transfer switches Q1 to Q4 and the secondary switches Qs1 to Qs4 to operate under zero-voltage switching (ZVS). Region III of Figure 3C is similar to Figure 3B, and similar waveforms are obtained. The transfer switches Q1 to Q4 and the secondary switches Qs1 to Qs4 also operate under ZVS. Furthermore, in Region VI of Figure 3D, the phase shift amount D2 is set to be larger than the phase shift amount D1 by a certain amount. This results in waveforms similar to those in Figure 3B. Similarly, the transfer switches Q1 to Q4 and the secondary switches Qs1 to Qs4 also operate under ZVS. Therefore, the three-phase shift (TPS) mode and variable frequency technology of the present application enable the resonant conversion module 1 to achieve excellent efficiency even at low power. Note that, as an example, the waveforms of the inductor current Il shown in Figures 3B to 3D are shown as straight line waveforms connecting points for simplification.

[0028] FIG. 3E shows an example of a waveform when the resonant conversion module of the present invention operates in single-phase shift mode, and FIG. 3F shows an example of a waveform when it operates in two-phase shift mode. These waveforms will be described with reference to FIGS. 1A to 3E. In FIG. 3E, the controller 6 operates the resonant conversion module 1 in single-phase shift (SPS) mode, phase-shifting only the control signal for either the transfer switches Q1 to Q4 or the secondary-side switches Qs1 to Qs4. Conversely, in FIG. 3F, the controller 6 operates the resonant conversion module 1 in two-phase shift (DPS) mode, phase-shifting both the control signals for the transfer switches Q1 to Q4 and the secondary-side switches Qs1 to Qs4. This causes the inductor current Il to vary in a piecewise sinusoidal manner via I0, I1, -I0, and -I1. Therefore, the waveform of the inductor current Il becomes more complex due to the effects of resonance, making it difficult to design parameters such as the resonant tank using a simple formula. It is also difficult to accurately grasp the waveform of the inductor current Il during operation by calculating its value.

[0029] FIG. 4A is a flowchart of a parameter calculation method for obtaining resonance parameters using a resonant conversion module of the present invention, and will be described with reference to FIGS. 1A to 4A. The parameter calculation method shown in FIG. 4A can be applied primarily to a resonant conversion module 1 with an SR-DAB configuration. A resonant conversion module 1 with a DAB configuration typically includes only a resonant inductor Lr1, and resonance parameters can be calculated using a relatively simple formula. However, by applying the parameter calculation method of FIG. 4A, it is possible to obtain more accurate resonance parameters even in a resonant conversion module 1 with a DAB configuration. Furthermore, the method can also be applied to other resonant converters, such as LLC and LC. However, to clarify the technical features of the present invention, the parameter calculation method of FIG. 4A will be described primarily when applied to a resonant conversion module 1 with an SR-DAB configuration.

[0030] Specifically, the parameter calculation method of the present invention begins by first setting preset operating conditions for the resonant converter. These preset operating conditions include the requirement that the operating frequency of the resonant converter be within a specific frequency range (S100). Prior to parameter calculation, it is necessary to determine the preset operating conditions for the resonant converter 100 (resonant conversion module 1). These preset operating conditions include, for example, the maximum power Po=Pmax (see FIG. 2C, e.g., 6.6 kW) at the device end of the resonant converter 100 (resonant conversion module 1) and the fluctuation range of the voltage Vo (see FIG. 2C, e.g., 280 V to 460 V). Other parameters that may be included include upper and lower limits for the values ​​of the resonant inductors Lr1 and Lr2, upper and lower limits for the values ​​of the resonant capacitors Cr1 and Cr2, and the DC voltage Vdc. Of these, the condition that must be included is the specific frequency range (e.g., 400 kHz to 600 kHz) of the operating frequency of the resonant converter 100 (resonant conversion module 1).

[0031] Note that the frequency fluctuation in the present invention corresponds to the so-called variable frequency function. Specifically, in the constant power charging mode CP, the output power of the resonant conversion module 1 is constant, while the output current varies in response to fluctuations in voltage. Therefore, when charging the device 200 (e.g., a battery), the current decreases as the voltage increases, and the output conditions of the resonant conversion module 1 change. As a result, the operating frequency calculated by the controller 6 also changes (for example, fluctuates within a range of 400 kHz to 600 kHz), making the frequency variable. Furthermore, the parameter calculation method of the present invention can be executed by a device equipped with a processor (e.g., a computer), and an operator can input preset operating conditions into pre-designed calculation software and further select and operate to obtain optimal parameters.

[0032] Next, multiple sets of first resonance parameters are calculated within a specific frequency range (S200). The first resonance parameters include the turns ratio of the transformer of the resonant converter. After setting the operating frequency (400 kHz to 600 kHz), multiple combinations of parameters for the resonant circuit B can be calculated based on the operating frequency. These combinations include, for example, multiple inductance values ​​of the resonant inductors Lr1 and Lr2, multiple capacitance values ​​of the resonant capacitors Cr1 and Cr2, and multiple turns ratios n of the transformer T. These resonance parameters (i.e., combinations of inductance values, capacitance values, and turns ratios n) can satisfy preset operating conditions when the controller 6 controls the resonant converter 100 (resonant conversion module 1).

[0033] Next, from among the multiple sets of first resonance parameters, a first resonance parameter that minimizes the current ripple flowing through the resonant inductor of the resonant converter is selected, and multiple sets of second resonance parameters are calculated based on the turns ratio of the selected first resonance parameter and preset operating conditions (S300). From the multiple combinations of first resonance parameters, a first resonance parameter that minimizes the current ripple (min_Irms) of the effective value Il (rms) of the inductor current Il flowing through the resonant inductor Lr1 of the resonant converter 100 (resonant conversion module 1) is selected. The current ripple Q can be calculated using, for example, the equation Q = min 1 / N ΣIl_rms(i)^2 × (1 + n^2), but is not limited to this and various other existing calculation methods can also be applied. Then, the turns ratio n of the selected first resonance parameter is set as the selected turns ratio N = n, and multiple sets of second resonance parameters are calculated based on this selected turns ratio N and the same preset operating conditions. These second resonant parameters include a fixed selected turns ratio N=n and multiple combinations of inductor and capacitance values.

[0034] FIG. 4B shows the value distribution of the second resonance parameter of the present invention, and will be described with reference to FIG. 4A. The above-described multiple sets of second resonance parameters are plotted in graphs based on the inductance value Lv and capacitance value Cv to obtain a distribution diagram as shown in FIG. 4B. Each point on this distribution diagram represents one second resonance parameter, and each second resonance parameter includes at least the inductance value Lv, capacitance value Cv, and a selected turns ratio N. Therefore, the turns ratios of all the second resonance parameters are the same. The band-shaped region BI shown in FIG. 4B is a preferred second resonance parameter, characterized by a relatively small inductance value Lv or capacitance value Cv, which is advantageous for reducing the size of inductors and capacitors.

[0035] Referring again to FIG. 4A, the parameter calculation method of the present invention includes a step (S400) of finally selecting one of the multiple sets of second resonance parameters and adopting it as the resonance parameter. The second resonance parameters in the band region B1 shown in FIG. 4B are all suitable second resonance parameters and can be used to set the resonance circuit B of the resonant converter 100 (resonant conversion module 1). Furthermore, among these second resonance parameters, the second resonance parameters located higher have a larger capacitance value Cv (lower voltage stress), a smaller inductance value Lv, and a higher total harmonic distortion (THD) of the inductor current Il. On the other hand, the second resonance parameters located lower have a smaller capacitance value Cv (higher voltage stress), a larger inductance value Lv, and a lower THD of the inductor current Il. Furthermore, the larger the numerical value, the larger the volume occupied by the component, and the difference in capacitance value Cv in particular becomes more pronounced.

[0036] Therefore, the designer can select any one of the multiple second resonance parameters within the band BI according to specific conditions and set the resonant circuit B of the resonant converter 100 (resonant conversion module 1). These specific conditions may include, for example, manufacturer or customer requirements, circuit size constraints, etc. In this way, the designer can select and apply the optimal parameters within the band BI according to the application and actual requirements of the resonant converter 100 (resonant conversion module 1). As described above, the present invention provides an optimization process for the resonant converter 100 (resonant conversion module 1), making it possible to obtain more appropriate resonance parameters (i.e., the inductor value Lv, the capacitance value Cv, and the turns ratio n) through accurate analysis.

[0037] FIGS. 5A to 5C are simulation waveform diagrams using one embodiment of the second resonant parameter in the resonant converter of the present invention, and FIGS. 6A to 6C are simulation waveform diagrams using another embodiment. These will be described with reference to FIGS. 1A to 4B. The second resonant parameters shown in FIGS. 5A to 5C are a combination of a low inductor value Lv and a high capacitance value Cv, and the operating frequency Fs varies in the range of 400 kHz to 600 kHz. As is clear from the power / voltage distribution diagram in FIG. 5A, the resonant converter 100 (resonant conversion module 1) can achieve zero-voltage switching (ZVS) at each point under control of the controller 6. Furthermore, FIGS. 5B and 5C are three-dimensional graphs. In FIG. 5B, the parameter of the operating frequency Fs is added to the power / voltage distribution, and in FIG. 5C, the parameter of the inductor current Il (rms) is also added. From these, it can be seen that zero-voltage switching (ZVS) can be achieved over a wide power / voltage range by using the second resonant parameters of a low inductor value Lv and a high capacitance value Cv shown in FIGS. 5A to 5C.

[0038] 6A to 6C are similar to FIGS. 5A to 5C, but differ in that the second resonance parameters used in FIGS. 6A to 6C have a high inductance value Lv and a low capacitance value Cv. Although the region in which zero voltage switching (ZVS) can be achieved is slightly different from that in FIGS. 5A to 5C, it can be seen that the overall range is almost the same. Therefore, by using the resonant converter parameter (optimization) calculation method of the present invention, it is possible to optimally select the resonant inductors Lr1 and Lr2, the resonant capacitors Cr1 and Cr2, and the transformer T in accordance with their respective costs. This allows for zero voltage switching (ZVS) to be achieved and the efficiency of the resonant converter 100 (resonant conversion module 1) to be improved while reducing the cost of the resonant circuit B.

[0039] After the calculation and design of the resonant parameters are completed, closed-loop design and operation of the resonant converter 100 (resonant conversion module 1) must be performed. This allows for confirmation of parameters such as the phase shift and dead time of the resonant converter 100 (resonant conversion module 1), and adjustment of the pulse-width modulation signal PWM output by the controller 6 to further improve the operational stability and efficiency of the resonant converter 100 (resonant conversion module 1). Specifically, FIG. 7A is a system control block diagram of the resonant converter of the present invention, which will be described in conjunction with FIGS. 1A to 6C and with repeated reference to FIGS. 2A and 7A. Because the resonant converter 100 (resonant conversion module 1) employs an SR-DAB configuration, the inductor current resonates, resulting in more complex current changes than in a DAB configuration. Therefore, it is difficult to calculate each parameter using a simple formula. Therefore, in addition to the parameter calculation method shown in FIG. 4A, the resonant converter 100 (resonant conversion module 1) needs to perform a program simulation to obtain the changes in the voltage Vo and power Po at the device end, and obtain the appropriate parameters through the closed-loop control block of FIG. 7A so that the controller 6 can properly adjust the pulse width modulation signal PWM.

[0040] In FIG. 7A, Ib is the battery current, and Ib* is the battery current command. Is(avg) is the measured output current. H(s) is the transfer function of the feedback term. C(s) calculates the conversion equation between Ib and Is(avg). f^-1 is the controller. e^-3Ts / 2 is the topology model used to calculate the output current. R / sRCin+1 is a virtual battery model. Specifically, when the resonant converter 100 (resonant conversion module 1) charges a battery, it is primarily required to control the current Ib at the device end. The sum of the current Ib and the current supplied to the capacitor Co becomes the average current Is(avg) after the secondary-side switch of the secondary-side switch circuit 5. Therefore, as shown in FIG. 7A, it is possible to configure a closed-loop system control block for battery charging using the resonant converter 100 (resonant conversion module 1). In FIG. 7A, the controller 6 sets the current command Ib* and performs closed-loop control based on the error between the current command Ib* and the actual current Ib. This adjustment is performed mainly by adjusting the operating frequency Fs, the phase shift amounts D1, D2, and ψ, so that the current Ib is appropriately controlled in accordance with the current command Ib*.

[0041] In addition, the processing speed and processing volume of the controller 6 are related to the cost of the controller 6. Furthermore, when the battery is charged in the constant power charging mode CP, the voltage Vo and current Ib fluctuate over time, so that if the controller 6 performs real-time calculations, it is required to have a faster processing capability or a larger amount of processing, which makes it difficult to reduce the cost of the controller 6. Therefore, the present invention improves the operational stability and efficiency of the resonant converter 100 (resonant conversion module 1) by acquiring appropriate parameters using a closed-loop system control block, and also reduces the cost of the controller 6 by using a table look-up method, allowing the use of a lower-cost digital signal processor (DSP).

[0042] Specifically, FIG. 7B is an operating frequency table reference control block diagram of the present invention, which will be described with repeated reference to FIGS. 1A to 7A, as well as FIGS. 2A and 7A to 7B. In the closed-loop block of FIG. 7A, the controller 6 reads the voltage Vo and power Po at the device terminal and inputs the voltage Vo and power Po into a preset first parameter table 7 shown in FIG. 7B to obtain the corresponding operating frequency Fs. Specifically, the controller 6 can obtain a frequency value Fv corresponding to the voltage Vo and power Po through the first parameter table 7. However, because the controller 6 reads the voltage Vo and power Po at high speed, the frequency value Fv may fluctuate due to temporary noise associated with the reading. Therefore, the frequency value Fv is filtered via a filter block PF to obtain a more stable frequency command Cf. This frequency command Cf corresponds to the operating frequency Fs used to switch the switches (change-over switches Q1 to Q4 and secondary switches Qs1 to Qs4). The filter block PF is preferably a low-pass filter capable of removing high-frequency noise, but is not limited to this. In addition to a method of directly using the value obtained by the filter block PF as the frequency command Cf, a method of using the difference between the frequency value Fv and the filtered value as the frequency command Cf is also possible. This difference method generally allows for more accurate adjustment. Therefore, the method of expressing the frequency command Cf is not limited to the above example, and various methods are applicable.

[0043] Referring to FIG. 7A, the average current Is(avg) is input to the f^-1 block, and phase shift amounts D1, D2, and ψ are obtained and used to adjust the current Ib via the closed-loop control block. The phase shift amounts D1, D2, and ψ are numerical values, and the controller 6 converts these numerical values ​​into delays, such as time differences, to achieve the phase shift effect. Referring to FIG. 7C, the phase shift amounts D1, D2, and ψ are converted into corresponding delay times via a ψ conversion block. A pulse-width modulation command Cpwm is then generated based on the delay times. The controller 6 then adjusts the pulse-width modulation signal PWM in accordance with the pulse-width modulation command Cpwm and controls the switching of the switches (the change-over switches Q1 to Q4 and the secondary switches Qs1 to Qs4) using the pulse-width modulation signal PWM. However, various numerical conversion methods by the controller 6 are conceivable, other than control using delay times, and the present invention is not limited to this embodiment.

[0044] FIG. 7D is a block diagram of the dead time table reference control of the present invention, which will be described with repeated reference to FIGS. 1A to 7A, 2A, 7A, and 7D. When switching the upper and lower arms of the transfer switches Q1 to Q4 and the secondary-side switches Qs1 to Qs4, a dead time is required, and zero-voltage switching (ZVS) control is also required. Therefore, the control method for the resonant converter 100 (resonant conversion module 1) of the present invention can obtain an appropriate dead time by referencing a table. In other words, based on the voltage Vo and power Po at the device end, the dead time when switching the upper and lower switches of the switching circuit 4 and secondary-side switch circuit 5 of the resonant converter 100 (resonant conversion module 1) can be obtained by referencing a table. In this way, the controller 6 controls the transfer switches Q1 to Q4 and the secondary-side switches Qs1 to Qs4 based on the dead time, thereby achieving zero-voltage switching (ZVS).

[0045] Specifically, in the closed-loop block of Fig. 7A, the controller 6 reads the voltage Vo and power Po at the device terminals and inputs the voltage Vo and power Po into preset second parameter tables 7A and 7B shown in Fig. 7D to obtain the dead times when the up and down switches of the switching circuit 4 and the secondary-side switch circuit 5 are switched. Specifically, the controller 6 uses the second parameter tables 7A and 7B to obtain a dead time value DTp corresponding to the primary-side dead time when the primary-side changeover switches Q1 to Q4 are switched and a dead time value DTs corresponding to the secondary-side dead time when the secondary-side switches Qs1 to Qs4 are switched. These dead time values ​​DTp and DTs are then converted into a dead time command Cdt by the dead time control block 9, and the dead time command Cdt is used as a control signal corresponding to the dead time when the up and down switches of the switching circuit 4 and the secondary-side switch circuit 5 are switched.

[0046] Finally, by integrating FIG. 2A and FIG. 7A to FIG. 7D, the closed-loop control flowchart of the resonant converter shown in FIG. 8 is obtained. Specifically, the closed-loop control flow of the resonant converter in FIG. 8 includes a first flow (S500 to S560) and a second flow (S600 to S680). The first flow (S500 to S560) is a fast-response flow, which is mainly intended to increase the response speed of the resonant converter 100 (resonant conversion module 1), improve stability, and provide better control effects. Therefore, the controller 6 usually executes this flow at a faster cycle (e.g., 50 μs) and always executes the steps of the first flow (S500 to S560) regardless of whether the resonant converter 100 is operating in a steady state. On the other hand, the second flow (S600 to S680) is a slow-response flow, which is mainly a process related to dead time and is intended to improve the operating efficiency of the resonant converter 100. In this flow, the response speed does not significantly affect the stability of the resonant converter 100 (resonant conversion module 1), so there is no need to frequently refer to the table or update the values. Therefore, the controller 6 normally executes this flow at a slower frequency (for example, 1 ms), and executes the second flow (S600 to S680) after the resonant converter 100 has transitioned to a steady state.

[0047] The first flow (S500-S560) first includes a step (S500) of confirming the device end current of the resonant converter. Subsequently, a step (S520) of calculating a phase shift of the resonant converter based on the current error is performed. Referring to FIG. 7A, the controller 6 sets a current command Ib*, calculates a secondary side average current Is(avg) based on a first error between the actual current Ib and the current command Ib*, and further calculates phase shifts D1, D2, and ψ based on the secondary side average current Is(avg). Next, a step (S540) of confirming an operating frequency corresponding to the device end voltage and power based on a first parameter table is performed (see FIG. 7B for details; a detailed description is omitted here). Finally, a step (S560) of adjusting a pulse width modulation signal for controlling the resonant converter based on the calculated phase shift and operating frequency is performed (see FIG. 7C for details; a detailed description is also omitted here).

[0048] The start of the second flow (S600-S680) can be determined during the execution of the first flow (S500-S560), and can be executed at any timing of steps (S500) to (S520) (here, an example will be described in which it is executed after step (S500)). Then, when the resonant converter 100 transitions to a steady state and a condition (S620) described below is satisfied, the main processing (S640 and thereafter) of the second flow (S600-S680) is executed. Note that the controller 6 can determine whether the resonant converter 100 is in a steady state by monitoring whether parameters such as the operating frequency Fs, current Ib, phase shift amount ψ, D1, and D2 of the resonant converter 100 are within predetermined ranges. The second flow (S600-S680) first includes a step (S600) of confirming the phase shift amount and operating frequency of the resonant converter. After the first flow (S500 to S560) is executed at least once, the phase shift amounts D1, D2, ψ and the operating frequency Fs of the resonant converter 100 (resonant conversion module 1) are acquired. Then, a step (S620) is performed to determine whether the phase shift amounts D1, D2, ψ and the operating frequency Fs are within a predetermined range. If both the phase shift amounts D1, D2, ψ and the operating frequency Fs are within the predetermined range, the efficiency of the resonant converter 100 (resonant conversion module 1) can be further improved by adjusting the dead time. Therefore, a step (S640) is performed to check the dead time corresponding to the voltage and power at the device end of the resonant converter by referring to the second parameter table (see FIG. 7D for details, and a detailed description will be omitted here).

[0049] Then, step S660 is performed to adjust the dead time and generate a pulse-width modulation signal for controlling the resonant converter. The controller 6 adjusts the pulse-width modulation signal PWM using a dead-time command Cdt corresponding to the dead time. The specific adjustment method is similar to the method shown in FIG. 7C, in which the value of the dead-time command Cdt is converted into a delay time and reflected in the pulse-width modulation signal PWM generation process (details on this point are also omitted). On the other hand, if step S620 determines that the dead time is not within the predetermined range, that is, if any of the phase shift amounts D1, D2, ψ, and the operating frequency Fs is not within the predetermined range, step S680 is performed to set the dead time to a preset value. In this case, since the resonant converter 100 (resonant conversion module 1) may not yet have reached an optimal state, the dead time is not adjusted and the preset value is used. Then, step S660 is performed to continue normal pulse-width modulation control.

[0050] The closed-loop control flow shown in FIG. 8 is preferably applied when the resonant converter 100 (resonant conversion module 1) operates in a constant power charging mode CP (i.e., when the battery charging / discharging mode is the constant power charging mode CP). In this case, it is preferable to enter step (S500) after confirming that the mode is the constant power charging mode CP. The main reason for this is that in the constant power charging mode CP, the voltage Vo and the current Ib are not constant values, and high-efficiency operation must be achieved by constantly adjusting parameters such as the phase shift amounts D1, D2, ψ, and the operating frequency Fs. On the other hand, the flow shown in FIG. 8 can also be applied to the constant current charging mode CC and the constant voltage charging mode CV. However, since the voltage Vo or the current Ib is fixed in these modes, some parameters such as the phase shift amounts D1, D2, ψ, and the operating frequency Fs are not constantly adjusted, and the overall efficiency improvement is not so significant.

[0051] FIG. 9A is a zero-voltage switching (ZVS) region distribution diagram obtained by executing a resonant converter using a simulation program and substituting the first design parameter. FIG. 9B is a ZVS region distribution diagram obtained by substituting the second design parameter. FIG. 9C is a ZVS region distribution diagram obtained by substituting the third design parameter. These diagrams should be referred to in conjunction with FIGS. 1A to 8. FIGS. 9A to 9C are ZVS region distribution diagrams created primarily using a simulation program such as MATLAB® / Simulink based on the circuit configuration of the resonant converter 100 (resonant conversion module 1) and the parameters described above in FIGS. 1A to 8. The differences between FIGS. 9A to 9C primarily lie in the differences in parameters. Specifically, in FIG. 9A, the inductance value Lv of the resonant inductor Lr1 is set to 4 μH, and the operating frequency Fs varies within a range of 400 kHz to 600 kHz. FIG. 9B is similar to FIG. 9A, but differs in that the inductance value Lv of the resonant inductor Lr1 is set to 5 μH. FIG. 9C is similar to FIG. 9B, except that the operating frequency Fs is fixed at 400 kHz.

[0052] The region in FIG. 9A shows that the resonant converter 100 (resonant conversion module 1) can operate in zero-voltage switching (ZVS) mode even in the low power Po and low voltage Vo regions. Comparing FIG. 9A and FIG. 9B, different inductor values ​​Lv have their own advantages. When Lv is 4 μH, the ZVS range is somewhat narrower under high power Po (or high voltage Vo) conditions, but the ZVS range is wider under low power Po (or low voltage Vo) conditions. Conversely, when Lv is 5 μH, the opposite trend is observed, allowing for the selection of an optimal configuration depending on the actual requirements of the circuit. Furthermore, because the operating frequency Fs is fixed at 400 kHz in FIG. 9C, comparing FIG. 9B and FIG. 9C shows that the resonant converter 100 (resonant conversion module 1) can still operate in a wider ZVS range in the low power Po and low voltage Vo region in FIG. 9B. Therefore, for the same inductor value Lv, adopting variable frequency operation makes it possible to achieve a wider ZVS range and improve the efficiency of the resonant converter 100 (resonant conversion module 1).

[0053] FIG. 10A is a circuit block diagram of a second embodiment of the resonant conversion module of the present invention. Please refer to FIGS. 1A to 9C together. The biggest difference between FIG. 10A and FIG. 2A is that the resonant conversion module 1 has a dual active bridge (DAB) circuit configuration, and the resonant circuit B includes a transformer T and a resonant inductor Lr1. Other circuit configurations and connections are similar to those in FIG. 2A, so detailed descriptions are omitted. Note that the resonant circuit B of the resonant conversion module 1 of FIG. 10A can also include capacitors Cr1 and Cr2 as shown in FIG. 2A, but their capacitance values ​​are set relatively small (e.g., several nF or pF). Therefore, Cr1 and Cr2 do not contribute to resonance but function as DC blocking capacitors. FIG. 10B is a simplified equivalent circuit diagram of the resonant conversion module of the second embodiment of the present invention. Le is the primary-side equivalent inductor, and Ce1 and Ce2 are the primary-side and secondary-side equivalent capacitors, respectively. Furthermore, Lb is the battery's equivalent inductor, Rb is the battery's equivalent impedance, and Eb is the battery voltage. Next, Figure 11A is a closed-loop system control block diagram used by the resonant conversion module in the second embodiment of the present invention when charging a battery. In the closed-loop control block, since the charging mode is mainly battery charging, the battery voltage rises or falls depending on the capacity and is not a constant value. Therefore, control of the current Ib is important in the charging mode.

[0054] Furthermore, under closed-loop control in the charging mode, the resonant converter 100 (resonant conversion module 1) can execute the closed-loop control flow shown in FIG. 8 in addition to the closed-loop control shown in FIGS. 7A and 11A. Specifically, the current command Ib* is compared with the current Ib actually fed back to the battery, and the current Is(avg) charging the capacitor Co is calculated from the difference. Meanwhile, the phase shift amounts D1, D2, and ψ derived from the controller are actually output to the topology (here represented by the equivalent transfer function e^-3Ts / 2), and then the output current Is(avg) can be measured. This current is further used to derive the current Ib charging the battery from the capacitor voltage Vc and the battery voltage Eb. In this closed-loop control operation, the first flow (S500 to S560) is first executed to improve the stability of the resonant converter 100. Then, by executing the second flow (S600 to S680) after the resonant converter 100 (resonant conversion module 1) has transitioned to a stable state, the operating efficiency of the resonant converter 100 (resonant conversion module 1) is improved. This is because when the resonant converter 100 (resonant conversion module 1) is in a transient state (immediately after start-up or when an abnormality occurs), the detected values ​​of the operating frequency Fs, current Ib, etc. tend to fluctuate significantly, and in such a case, it is necessary to first quickly stabilize the control of the resonant converter 100 using the first flow (S500 to S560). Then, after it is confirmed that the resonant converter 100 (resonant conversion module 1) is operating in a steady state, the process proceeds to the second flow (S600 to S680).

[0055] FIG. 11B is a block diagram of a closed-loop system control used by the resonant conversion module according to the second embodiment of the present invention when discharging the battery, and should be referred to in conjunction with FIGS. 1A to 11A. In the discharge mode, the battery discharges toward the power supply end of the resonant converter 100 (resonant conversion module 1), i.e., the DC voltage Vdc end, so the DC voltage Vdc at the power supply end is treated as a load (this simulates the equivalent resistance when an inverter is connected to the power supply end, and fluctuations in the load have a significant impact on the stabilization time). Therefore, the DC voltage Vdc at the power supply end is controlled to a constant value, but it is mainly the primary-side current Ip(avg) that fluctuates. Therefore, the focus of control in the discharge mode is to maintain the DC voltage Vdc.

[0056] Specifically, when the resonant converter 100 (resonant conversion module 1) performs closed-loop control in discharge mode, it compares the voltage command value Vdc* with the actual DC voltage Vdc, and calculates the primary-side current Ip(avg) and phase shift amounts D1, D2, and ψ using conversion equations based on the error. This results in a closed-loop control that stabilizes the DC voltage Vdc, as shown in Figure 11B. Because load fluctuations significantly affect the stabilization time, a feedforward loop is added in this invention to improve the stability of the closed-loop control. As shown in Figure 11B, 1 / R* is a compensation amount defined as R* = (2Vdc^2 / Vm*Im), where Vm and Im represent the excitation voltage and excitation current of the transformer T.

[0057] Specifically, the compensation amount is a value calculated in advance by a feedforward loop and is effective in counteracting the effects of load fluctuations. Adding the compensation amount in advance eliminates the need for large corrections when the controller 6 adjusts other parameters, improving the stability of closed-loop control in discharge mode. Therefore, the controller 6 first checks whether the resonant converter 100 (resonant conversion module 1) is operating in discharge mode. If it is, it applies the compensation amount 1 / R* based on the DC voltage. Then, it sets a voltage command value Vdc* and calculates the error between the actual DC voltage Vdc and the voltage command value Vdc*. The primary-side current Ip(avg) is calculated from this error. Finally, the primary-side current Ip(avg) is summed with the compensation amount 1 / R* calculated in advance to obtain a second error amount, and the DC voltage Vdc is adjusted based on this second error amount.

[0058] To summarize, the resonant converter 100 of the present invention can be configured with one or more resonant conversion modules 1, which allows the resonant converter 100 to process large power by connecting the resonant conversion modules 1 in parallel. In addition, the present invention mainly controls the resonant conversion modules 1 using an interleaved control technique to reduce the current stress on the input filter of the switching power supply (SPS) modulation. This is because the interleaved control has the effect of canceling out ripple current, and reducing the ripple current allows the size of the filter element to be reduced.

[0059] However, the above description is merely a detailed description of preferred embodiments of the present invention using drawings, and the features of the present invention are not limited thereto. That is, the scope of the present invention is defined by the claims below, and all embodiments that undergo similar modifications and improvements based on the spirit of the claims are also included in the scope of the present invention. Various modifications and changes that would be easily conceivable to a person familiar with the technical field to which the present invention pertains are also included in the scope of the claims of the present invention. [Explanation of symbols]

[0060] 100 Resonant Converter A Primary circuit 1, 1A, 1B Resonant Conversion Module C1, Co capacitor 4 Switching circuit 42 First switching bridge arm Q1 First changeover switch Q2 Second changeover switch 44 Second switching bridge arm Q3 Third switch Q4 4th changeover switch 46 Third switching bridge arm Q5 Third changeover switch Q6 4th changeover switch M Common Core Module Pgnd Primary side ground terminal B resonant circuit Lr1, Lr2 resonant inductors Cr1, Cr2 resonant capacitors T transformer Wp primary winding Wp1 1st primary winding Wp2 Second primary winding PC center tap terminal Ws Secondary winding C Secondary circuit 5 Secondary side switch circuit 52 First secondary bridge arm Qs1 1st secondary switch Qs2 Secondary side switch 54 Secondary bridge arm Qs3 Third secondary switch Qs4 4th secondary switch Sgnd Secondary ground terminal Pp1, Pp2, Pp3, Ps1, Ps2 nodes 6 Controller 200 equipment Vdc DC voltage Vo, V1, V2, V3, Vpp, Vss, Eb voltage PWM Pulse Width Modulation Signal Le, Lb Equivalent inductor Ce, Ce1, Ce2 equivalent capacitors Rb equivalent impedance Vl Inductor voltage Vc Capacitor voltage Vdc* voltage command Ip(avg), Is(avg), Ib current Ib* Current command Il, I0, I1, -I0, -I1 Inductor current Q current ripple CC constant current charging mode CP Constant Power Charging Mode CV constant voltage charging mode ψ, D1, D2 Phase shift amount 1 / R* Compensation amount n, N turns ratio Lv, Lv1, Lv2 inductor values Cv, Cv1, Cv2 capacitor values Fs Operating frequency Cf frequency command Fv frequency value T period DTp, DTs Dead time value Cpwm Pulse width modulation command Cdt Dead Time Command Po, Pmax, Pmin Power BI Band Area 7, 7A, 7B Parameter Tables PF filter block 8 ψ transformation block 9 Dead Time Control Block

Claims

1. A parameter calculation method for obtaining resonance parameters of a resonance converter, comprising: setting a preset operating condition in which the operating frequency of the resonant converter is defined to be within a specific frequency range; calculating a plurality of sets of first resonant parameters in the particular frequency range, the first resonant parameters including a turns ratio of a transformer of the resonant converter; selecting a minimum first resonance parameter from the plurality of sets of first resonance parameters, which minimizes a current ripple flowing through a resonance inductor of the resonance converter; calculating a plurality of sets of second resonance parameters based on the turns ratio of the minimum first resonance parameter and the preset operating conditions, and selecting one from the plurality of sets of second resonance parameters and adopting it as the resonance parameter.

2. The parameter calculation method according to claim 1 , wherein the second resonance parameter includes an inductance value of the resonance inductor and a capacitance value of the resonance capacitor.

3. (a) measuring a current at a device end of the resonant converter and calculating a phase shift of the resonant converter based on the current; (b) determining an operating frequency corresponding to the voltage and power at the device terminal according to a first parameter table; 2. The parameter calculation method of claim 1, further comprising the step (c) of adjusting a pulse width modulated signal for controlling the resonant converter according to the phase shift amount and the operating frequency.

4. setting a current command and calculating a secondary-side average current based on a first error amount between the current and the current command; The parameter calculation method according to claim 3 , further comprising the step of: calculating the amount of phase shift based on the secondary side average current.

5. determining a frequency value corresponding to the voltage and the power according to the first parameter table; The parameter calculation method according to claim 3 , further comprising the step of: filtering the frequency value to obtain the operating frequency.

6. 4. The parameter calculation method according to claim 3, wherein the resonant converter includes a switching circuit on a primary side and a secondary-side switch circuit on a secondary side, and the phase shift amounts include a first phase shift amount in the switching circuit, a second phase shift amount in the secondary-side switch circuit, and a third phase shift amount between the switching circuit and the secondary-side switch circuit.

7. 4. The parameter calculation method according to claim 3, further comprising converting the phase shift amount into a corresponding delay time by a value-to-time conversion, and adjusting the pulse width modulation signal according to the delay time.

8. (d) determining whether the phase shift amount and operating frequency of the resonant converter are within a predetermined range; (e) determining a dead time corresponding to a voltage and a power at a device end of the resonant converter according to a second parameter table when both the phase shift amount and the operating frequency are within the predetermined range; 4. The parameter calculation method of claim 3, further comprising the step (f) of adjusting a pulse width modulated signal for controlling the resonant converter according to the dead time.

9. The resonant converter includes a primary side switching circuit and a secondary side switch circuit; 9. The parameter calculation method according to claim 8, further comprising the step of calculating a primary side dead time in the switching circuit and a secondary side dead time in the secondary side switch circuit using the second parameter table when both the phase shift amount and the operating frequency are within the predetermined ranges.

10. 9. The parameter calculation method according to claim 8, further comprising the step of setting the dead time to a preset value when either the phase shift amount or the operating frequency is not within the predetermined range.

11. The device end of the resonant converter is connected to a battery to charge and discharge the battery; The parameter calculation method includes: determining whether the resonant converter is operating in a steady state; 9. The method of claim 8, further comprising the step of: if the resonant converter is operating in the steady state, proceeding to step (d).

12. The device end of the resonant converter is connected to a battery to charge and discharge the battery; The parameter calculation method includes: confirming that the charging mode for the battery is a constant power charging mode; The parameter calculation method of claim 3 , further comprising the step of: proceeding to step (a) if said confirmation is obtained.

13. determining whether the battery is discharging; providing a compensation amount based on a DC voltage at a power supply end of the resonant converter; setting a voltage command and calculating a primary side current based on the DC voltage and the voltage command; 13. The parameter calculation method according to claim 12, further comprising the steps of: summing the primary side current and the compensation amount to obtain a second error amount; and adjusting the DC voltage based on the second error amount.

14. A resonant converter to which the parameter calculation method according to claim 1 is applied, The resonant converter comprises: a primary side circuit connected to a DC power source; a resonant circuit comprising a plurality of transformers with primary windings connected to the primary circuit; a secondary circuit including a plurality of sets of secondary switch circuits each having a first secondary bridge arm and a second secondary bridge arm connected in parallel, a first secondary bridge arm and a second secondary bridge arm of the secondary switch circuit are respectively connected to both ends of secondary windings of the plurality of transformers, and the secondary switch circuit is connected in parallel to a device.

15. the primary side circuit includes a plurality of sets of switching circuits; each of the plurality of sets of switching circuits includes a first switching bridge arm and a second switching bridge arm connected in parallel; The resonant converter according to claim 14 , wherein a first switching bridge arm and a second switching bridge arm of the plurality of sets of switching circuits are respectively connected across primary windings of the plurality of transformers.

16. 16. The resonant converter of claim 15, wherein the plurality of sets of switching circuits are connected in parallel or in series to the DC power supply.

17. 16. The resonant converter of claim 15, wherein the switching circuit is connected in parallel with the DC power source, and the resonant inductors of the resonant circuit form a common core structure.

18. the primary side circuit includes a switching circuit having a first switching bridge arm and a second switching bridge arm connected in parallel; the primary windings of the plurality of transformers are connected in series to form a series circuit of primary windings; The resonant converter of claim 14 , wherein the first switched bridge arm and the second switched bridge arm are respectively connected across the series circuit of the primary winding.

19. the switching circuit further comprises a third switching bridge arm connected in parallel; 19. The resonant converter of claim 18, wherein a center tap terminal is formed between two primary windings of the transformer, and the third switched bridge arm is connected to the center tap terminal.

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