Power conversion device

Abstract

To obtain a power conversion device capable of reducing the possibility of malfunction due to noise.SOLUTION: A power conversion device has a first power terminal, a power conversion part including a switching circuit, a second power terminal, and a control circuit. The control circuit generates a command value of a duty ratio based on current flowing through a terminal on the first or second power terminal side of the power conversion part, makes the switching circuit perform switching to a first transition direction based on a start timing of a switching cycle period, makes, in the switching cycle period, the switching circuit perform switching to a second transition direction based on a first timing when the first timing at which current becomes a peak or a bottom is within a period of a first period, and makes the switching circuit perform switching to the second transition direction based on a second timing according to the command value of the duty ratio when the first timing is within a period except the first period.SELECTED DRAWING: Figure 8

Description

Technical Field

[0001] The present invention relates to a power conversion device that converts power.

Background Art

[0002] There are power conversion devices that convert DC power into DC power. For example, Patent Document 1 discloses a power conversion device that performs peak current control.

Prior Art Documents

Patent Documents

[0003]

Patent Document 1

Summary of the Invention

Problems to be Solved by the Invention

[0004] A power conversion device that performs peak current control has the characteristic that the response of current control is fast, but there is a possibility of malfunction due to noise included in the detection signal of the current sensor. Therefore, it is desirable that malfunction is less likely to occur due to noise.

[0005] It is desirable to provide a power conversion device capable of reducing the possibility of malfunction due to noise.

Means for Solving the Problems

[0006] The power conversion device of the present invention includes a first power terminal, a power conversion unit, a second power terminal, and a control circuit. The power conversion unit is connected to the first power terminal. It is possible to alternately perform switching in the first transition direction and switching in the second transition directionIt includes a switching circuit. The second power terminal is led to a power conversion unit. The control circuit can control the operation of the switching circuit. The control circuit can generate a command value for the duty ratio of the switching circuit based on the current flowing through either one of the terminals on the side of the first power terminal of the power conversion unit and the terminals on the side of the second power terminal of the power conversion unit. The control circuit can cause the switching circuit to perform switching in the first transition direction based on the start timing of the switching period of the switching circuit. During the switching period, when the first timing at which the current becomes either a peak or a bottom is within the period of a first period preset in the switching period, the control circuit causes the switching circuit to perform switching in the second transition direction based on the first timing Performing peak current control thereby and when the first timing is outside the period of the first period, the control circuit causes the switching circuit to perform switching in the second transition direction based on a second timing corresponding to the command value of the duty ratio Performing average current control thereby and it is possible to do so.

Advantages of the Invention

[0007] According to the power conversion device of the present invention, the possibility of malfunction due to noise can be reduced.

Brief Description of the Drawings

[0008]

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[0009] Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[0010] [Configuration Example] FIG. 1 shows a configuration example of a power conversion device (power conversion device 1) according to an embodiment of the present invention. The power conversion device 1 is a non-insulated bidirectional DC / DC conversion device and is configured using a buck-boost chopper circuit. The power conversion device 1 includes power terminals T11 and T12, and power terminals T21 and T22. A DC power supply PDC is connected to the power terminals T11 and T12. The DC power supply PDC may be, for example, a DC power supply device. The power terminals T21 and T22 are connected to the battery BT. The power conversion device 1 can charge or discharge the battery BT by performing bidirectional operation.

[0011] The power conversion device 1 includes a voltage sensor 11, a capacitor 12, a power conversion unit 100, a current sensor 15, a filter circuit 16, a voltage sensor 17, and a control circuit 20.

[0012] The voltage sensor 11 is configured to detect the DC bus voltage Vdc. One end of the voltage sensor 11 is connected to the voltage line L11 led to the power terminal T11, and the other end is connected to the reference voltage line L12 led to the power terminal T12. The voltage sensor 11 detects the voltage on the voltage line L11 with reference to the voltage on the reference voltage line L12 as the DC bus voltage Vdc. Then, the voltage sensor 11 supplies the detection result of the DC bus voltage Vdc to the control circuit 20.

[0013] One end of the capacitor 12 is connected to the voltage line L11, and the other end is connected to the reference voltage line L12. The capacitor 12 has a capacitance Cdc.

[0014] The power conversion unit 100 has a switching circuit 14. The switching circuit 14 is configured to perform a switching operation based on the drive signal Sctl.

[0015] FIG. 2 shows a configuration example of the switching circuit 14. In FIG. 2, for convenience of explanation, the peripheral circuit of the switching circuit 14 is also shown. In this example, the switching circuit 14 has transistors SW1 and SW2. The transistors SW1 and SW2 are switching elements that perform switching operations based on the gate signals S1 and S2 included in the drive signal Sctl. In this example, the transistors SW1 and SW2 are configured using insulated gate bipolar transistors (IGBTs: Insulated Gate Bipolar Transistors). Although insulated gate bipolar transistors are used in this example, any switching element may be used.

[0016] Transistor SW1 is provided in the path connecting voltage line L11 and node N1, and is configured to connect node N1 to voltage line L11 when it is turned on. The collector of transistor SW1 is connected to voltage line L11, gate signal S1 is supplied to the gate, and the emitter is connected to node N1. Transistor SW2 is provided in the path connecting node N1 and reference voltage line L12, and is configured to connect node N1 to reference voltage line L12 when it is turned on. The collector of transistor SW2 is connected to node N1, gate signal S2 is supplied to the gate, and the emitter is connected to reference voltage line L12. Node N1 is the connection point of the emitter of transistor SW1 and the collector of transistor SW2. Node N1 is connected to voltage line L21 led to power terminal T21.

[0017] Note that switching circuit 14 is not limited to the circuit shown in FIG. 2, and various circuits can be applied. The waveform of voltage V1 on the power terminal T21, T22 side of switching circuit 14 is a rectangular waveform corresponding to the switching operation by PWM (Pulse Width Modulation).

[0018] Current sensor 15 is provided on voltage line L21 and is configured to detect current Ibat flowing in the path connecting filter circuit 16 and power conversion unit 100. Current sensor 15 includes, for example, a current transformer, an operational amplifier, etc. One end of current sensor 15 is connected to power conversion unit 100, and the other end is connected to filter circuit 16. Current sensor 15 detects current Ibat with the polarity that becomes positive when flowing from filter circuit 16 toward power conversion unit 100. Current sensor 15 supplies the detection result of current Ibat to control circuit 20.

[0019] The filter circuit 16 is a low-pass filter configured to remove high-frequency components included in the voltage V1 generated by the switching operation in the switching circuit 14. The filter circuit 16 has an inductor 16L and a capacitor 16C. The inductor 16L is provided on the voltage line L21, one end thereof is connected to the other end of the current sensor 15, and the other end is connected to one end of the capacitor 16C, one end of the voltage sensor 17, and the power terminal T21. The inductor 16L has an inductance Lbat and an internal resistance Rbat. One end of the capacitor 16C is connected to the voltage line L21, and the other end is connected to the reference voltage line L22 led to the power terminal T22. The capacitor 16C has a capacitance Cbat.

[0020] The voltage sensor 17 is configured to detect the battery voltage Vbat. One end of the voltage sensor 17 is connected to the voltage line L21, and the other end is connected to the reference voltage line L22. The voltage sensor 17 detects the voltage on the voltage line L21 with respect to the voltage on the reference voltage line L22 as the battery voltage Vbat. Then, the voltage sensor 17 supplies the detection result of the battery voltage Vbat to the control circuit 20.

[0021] The control circuit 20 is configured to control the operation of the power conversion device 1. The control circuit 20 is configured using, for example, one or more microcontrollers. The control circuit 20 generates a drive signal Sctl by performing arithmetic processing based on the DC bus voltage Vdc, the current Ibat, and the battery voltage Vbat. Thereby, the control circuit 20 controls the operation of charging the battery BT and the operation of discharging the battery BT. In the operation of charging the battery BT, the power conversion device 1 supplies power from the power terminals T11, T12 toward the power terminals T21, T22, and in the operation of discharging the battery BT, the power conversion device 1 supplies power from the power terminals T21, T22 toward the power terminals T11, T12. The control circuit 20 is configured to control the operation of the power conversion device 1 using peak current control and average current control.

[0022] FIG. 3 shows a configuration example of the control circuit 20. The control circuit 20 includes a command value generation unit 21, a DC bus voltage control unit 22, a battery voltage control unit 23, a division unit 24, a switching unit 25, a current ripple calculation unit 26, a current control unit 27, a slope compensation unit 28, a comparison unit 29, a window processing unit 31, a PWM processing unit 32, a drive signal generation unit 33, a carrier signal generation unit 34, and a setting unit 39.

[0023] The control circuit 20 has an AD conversion circuit that samples the DC bus voltage Vdc, the current Ibat, and the battery voltage Vbat at a sampling period corresponding to the sampling frequency fs. In this example, the sampling period Tadc (= 1 / fs) is half of the switching period Tpwm of the switching circuit 14. Then, the control circuit 20 performs arithmetic processing at the control period Tctl based on the average values of a plurality of digital values related to the DC bus voltage Vdc, the average values of a plurality of digital values related to the current Ibat, and the average values of a plurality of digital values related to the battery voltage Vbat. Hereinafter, the DC bus voltage Vdc, the current Ibat, and the battery voltage Vbat are appropriately used to represent the digital values generated by the AD conversion.

[0024] The control circuit 20 is configured to selectively perform one of constant control of the charge / discharge current (Constant Current control), constant control of the charge / discharge power (Constant Power control), constant control of the battery voltage Vbat during charging (Constant Voltage control), and constant control of the DC bus voltage Vdc during discharging (Constant Voltage control).

[0025] When the control circuit 20 performs constant control of the charge and discharge current, the command value generation unit 21 generates a current command value Icc* which is the command value of the current. When the control circuit 20 performs constant control of the charge and discharge power, it generates a power command value Pcp* which is the command value of the power. When the control circuit 20 performs constant control of the battery voltage Vbat during charging, it generates a battery voltage command value Vbat* which is the command value of the battery voltage. When the control circuit 20 performs constant control of the DC bus voltage Vdc during discharging, it is configured to generate a DC bus voltage command value Vdc* which is the command value of the DC bus voltage Vdc.

[0026] When the control circuit 20 performs constant control of the DC bus voltage Vdc during discharging, the DC bus voltage control unit 22 is configured to generate a current command value Ibat* which is the command value of the current Ibat by performing control so that the DC bus voltage Vdc becomes the same as the DC bus voltage command value Vdc* based on the DC bus voltage command value Vdc* and the DC bus voltage Vdc.

[0027] When the control circuit 20 performs constant control of the battery voltage Vbat during charging, the battery voltage control unit 23 is configured to generate a current command value Ibat* which is the command value of the current Ibat by controlling so that the battery voltage Vbat becomes the same as the battery voltage command value Vbat* based on the battery voltage command value Vbat* and the battery voltage Vbat.

[0028] When the control circuit 20 performs constant control of the charge and discharge power, the division unit 24 is configured to generate a current command value Ibat* which is the command value of the current Ibat by dividing the power command value Pcp* by the battery voltage Vbat.

[0029] When the control circuit 20 performs constant control of the charge and discharge current, the switching unit 25 supplies the current command value Icc* as the current command value Ibat* to the current control unit 27. When the control circuit 20 performs constant control of the charge and discharge power, the switching unit 25 supplies the current command value Ibat* supplied from the division unit 24 to the current control unit 27. When the control circuit 20 performs constant control of the battery voltage Vbat during charging, the switching unit 25 supplies the current command value Ibat* supplied from the battery voltage control unit 23 to the current control unit 27. When the control circuit 20 performs constant control of the DC bus voltage Vdc during discharging, the switching unit 25 supplies the current command value Ibat* supplied from the DC bus voltage control unit 22 to the current control unit 27. The current command value Ibat* is a negative value during charging and a positive value during discharging.

[0030] The current ripple calculation unit 26 is configured to generate a ripple value ΔIbat of the current Ibat based on the DC bus voltage Vdc and the battery voltage Vbat. The ripple value ΔIbat is calculated using, for example, the following equation EQ1.

Equation

[0031] Based on the current command value Ibat* supplied from the switching unit 25, the current Ibat detected by the current sensor 15, the ripple value ΔIbat of the current Ibat, the DC bus voltage Vdc, and the battery voltage Vbat, the current control unit 27 controls the current Ibat to be the same as the current command value Ibat*, thereby calculating a duty ratio command value Dbat*, which is a command value for the duty ratio of the switching circuit 14. This duty ratio command value Dbat* is used when performing average current control. The duty ratio command value Dbat* is calculated using, for example, the following equation EQ2.

Equation

Number

Number

[0032] The slope compensation unit 28 is configured to generate a signal SLP that gradually decreases from the peak current command value Ip* at a slope indicated by the slope value Ks based on the peak current command value Ip*, the slope value Ks, the signal ZRO, and the signal TP1.

[0033] The comparison unit 29 is configured to compare the current Ibat detected by the current sensor 15 with the signal SLP and generate a signal TP1 according to the comparison result.

[0034] The window processing unit 31 is configured to mask the signal TP1 or output the signal TP1 as the signal TP2 according to the timing of the signal TP1. The window processing unit 31 uses a time window set by the times Ton, Toff1, and Toff2 to mask the signal TP1 or output the signal TP1 as the signal TP2.

[0035] The PWM processing unit 32 is configured to control the operation of the drive signal generation unit 33 based on the duty ratio command value Dbat* and the signal TP2. When performing peak current control, the PWM processing unit 32 controls the operation of the drive signal generation unit 33 based on the signal TP2, and when performing average current control, it controls the operation of the drive signal generation unit 33 based on the duty ratio command value Dbat*. The PWM processing unit 32 operates based on the carrier signal CR. The PWM processing unit 32 causes the drive signal generation unit 33 to generate the drive signal Sctl so that the switching circuit 14 operates with the dead time Td.

[0036] The drive signal generation unit 33 is configured to generate the drive signal Sctl based on an instruction from the PWM processing unit 32. The control circuit 20 supplies this drive signal Sctl to the switching circuit 14.

[0037] The carrier signal generation unit 34 is configured to generate the carrier signal CR. The period of the carrier signal CR is the switching period Tpwm.

[0038] The setting unit 39 is configured to set parameters necessary for the operation of the control circuit 20. The setting unit 39 is configured to set, for example, the times Ton, Toff1, Toff2, and the dead time Td.

[0039] Figure 4 shows an example of average current control during discharge. (A) shows the waveform of the carrier signal CR, (B) shows the duty ratio command value Dbat*, (C) shows the waveform of the current Ibat, and (D) shows the waveform of the voltage V1. Since it is during discharge, the current Ibat is positive.

[0040] When performing average current control, the PWM processing unit 32 controls the operation of the drive signal generation unit 33 by comparing the duty ratio command value Dbat* and the carrier signal CR, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl. For example, during a period when the duty ratio command value Dbat* is higher than the value of the carrier signal CR, the voltage V1 becomes a low level and the current Ibat increases (Figs. 4(C) and (D)). This low level is the voltage of the reference voltage line L12. In the case of the circuit configuration shown in Fig. 2, when the transistor SW1 is turned off and the transistor SW2 is turned on, the voltage V1 becomes a low level. Also, for example, during a period when the duty ratio command value Dbat* is lower than the value of the carrier signal CR, the voltage V1 becomes a high level and the current Ibat decreases. This high level is the voltage of the voltage line L11. In the case of the circuit configuration shown in Fig. 2, when the transistor SW1 is turned on and the transistor SW2 is turned off, the voltage V1 becomes a high level.

[0041] Fig. 5 shows an example of peak current control during discharge. (A) shows the waveform of the current Ibat, (B) shows the waveform of the signal SLP, (C) shows the waveform of the signal TP1, (D) shows the waveform of the carrier signal CR, (E) shows the waveform of the signal ZRO, and (F) shows the waveform of the voltage V1. Since it is during discharge, the current Ibat, the signal SLP, and the peak current command value Ip* are positive.

[0042] When performing peak current control, at the start timing of the period related to the switching period Tpwm, the PWM processing unit 32 controls the operation of the drive signal generation unit 33, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a high level to a low level (Fig. 5(F)). In the case of the circuit configuration shown in Fig. 2, as the transistor SW1 changes from the on state to the off state and the transistor SW2 changes from the off state to the on state, the voltage V1 changes from a high level to a low level. As a result, the current Ibat increases (Fig. 5(A)). Also, at this start timing, the PWM processing unit 32 generates a pulse of the signal ZRO (Fig. 5(E)). The slope compensation unit 28 gradually decreases the signal SLP from the peak current command value Ip* at the slope indicated by the slope value Ks based on this signal ZRO (Fig. 5(B)). When the signal SLP and the current Ibat match, the comparison unit 29 generates a pulse of the signal TP1 (Fig. 5(C)). The slope compensation unit 28 returns the signal SLP to the peak current command value Ip* based on this signal TP1. In this example, the window processing unit 31 outputs the pulse of this signal TP1 as it is as the pulse of the signal TP2. The PWM processing unit 32 controls the operation of the drive signal generation unit 33 based on this signal TP2, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a low level to a high level (Fig. 5(F)). In the case of the circuit configuration shown in Fig. 2, as the transistor SW1 changes from the off state to the on state and the transistor SW2 changes from the on state to the off state, the voltage V1 changes from a low level to a high level. As a result, the current Ibat decreases (Fig. 5(A)).

[0043] During discharge, since the current Ibat is positive, when the current Ibat matches the signal SLP, the current Ibat is at its peak. The control circuit 20 performs peak current control by switching the switching circuit 14 at the timing when the current Ibat reaches its peak in this way.

[0044] Figure 6 shows an example of average current control during charging. (A) shows the waveform of the carrier signal CR, (B) shows the duty ratio command value Dbat*, (C) shows the waveform of the current Ibat, and (D) shows the waveform of the voltage V1. Since it is during charging, the current Ibat is negative.

[0045] When performing average current control, the PWM processing unit 32 controls the operation of the drive signal generation unit 33 by comparing the duty ratio command value Dbat* and the carrier signal CR, and the drive signal generation unit 33 generates the drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl. For example, during a period when the duty ratio command value Dbat* is higher than the value of the carrier signal CR, the voltage V1 becomes high level and the absolute value of the current Ibat increases (Figs. 6(C) and (D)). In the case of the circuit configuration shown in Fig. 2, when the transistor SW1 is turned on and the transistor SW2 is turned off, the voltage V1 becomes high level. Also, for example, during a period when the duty ratio command value Dbat* is lower than the value of the carrier signal CR, the voltage V1 becomes low level and the absolute value of the current Ibat decreases. In the case of the circuit configuration shown in Fig. 2, when the transistor SW1 is turned off and the transistor SW2 is turned on, the voltage V1 becomes low level.

[0046] Figure 7 shows an example of peak current control during charging. (A) shows the waveform of the current Ibat, (B) shows the waveform of the signal SLP, (C) shows the waveform of the signal TP1, (D) shows the waveform of the carrier signal CR, (E) shows the waveform of the signal ZRO, and (F) shows the waveform of the voltage V1. Since it is during charging, the current Ibat, the signal SLP, and the peak current command value Ip* are negative.

[0047] When performing peak current control, at the start timing of the period related to the switching period Tpwm, the PWM processing unit 32 controls the operation of the drive signal generation unit 33, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a high level to a low level (Fig. 7(F)). In the case of the circuit configuration shown in Fig. 2, as the transistor SW1 changes from the on state to the off state and the transistor SW2 changes from the off state to the on state, the voltage V1 changes from a high level to a low level. As a result, the absolute value of the current Ibat decreases (Fig. 7(A)). Also, at this start timing, the PWM processing unit 32 generates a pulse of the signal ZRO (Fig. 7(E)). The slope compensation unit 28 gradually decreases the signal SLP from the peak current command value Ip* at the slope indicated by the slope value Ks based on this signal ZRO (Fig. 7(B)). In other words, since the signal SLP and the peak current command value Ip* are negative, the absolute value of the signal SLP gradually increases. When the signal SLP and the current Ibat match, the comparison unit 29 generates a pulse of the signal TP1 (Fig. 7(C)). The slope compensation unit 28 returns the signal SLP to the peak current command value Ip* based on this signal TP1. In this example, the window processing unit 31 outputs the pulse of this signal TP1 as it is as the pulse of the signal TP2. The PWM processing unit 32 controls the operation of the drive signal generation unit 33 based on this signal TP2, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a low level to a high level (Fig. 7(F)). In the case of the circuit configuration shown in Fig. 2, as the transistor SW1 changes from the off state to the on state and the transistor SW2 changes from the on state to the off state, the voltage V1 changes from a low level to a high level. As a result, the absolute value of the current Ibat increases (Fig. 7(A)).

[0048] During charging, since the current Ibat is negative, when the current Ibat matches the signal SLP, the current Ibat is at the bottom. The control circuit 20 performs peak current control by switching the switching circuit 14 at the timing when the current Ibat reaches the bottom. That is, the peak current control according to the present technology is different from the so-called general peak current control that operates based on the peak.

[0049] In this way, when performing peak current control, the control circuit 20 generates the drive signal Sctl based on the signal TP2, and when performing average current control, the control circuit 20 generates the drive signal Sctl based on the duty ratio command value Dbat*. The control circuit 20 is configured to perform either peak current control or average current control in each period related to the switching period Tpwm.

[0050] Here, the power terminals T11, T12 correspond to a specific example of the "first power terminal" in the present disclosure. The power conversion unit 100 corresponds to a specific example of the "power conversion unit" in the present disclosure. The switching circuit 14 corresponds to a specific example of the "switching circuit" in the present disclosure. The power terminals T21, T22 correspond to a specific example of the "second power terminal" in the present disclosure. The control circuit 20 corresponds to a specific example of the "control circuit" in the present disclosure. The duty ratio command value Dbat* corresponds to a specific example of the "command value" in the present disclosure. The period related to the time Ton corresponds to a specific example of the "first period" in the present disclosure. The period related to the time Toff1 corresponds to a specific example of the "second period" in the present disclosure. The period related to the time Toff2 corresponds to a specific example of the "third period" in the present disclosure. The DC bus voltage Vdc corresponds to a specific example of the "first voltage" in the present disclosure. The battery voltage Vbat corresponds to a specific example of the "second voltage" in the present disclosure.

[0051] [Operation and Function] Subsequently, the operation and function of the power conversion device 1 of the present embodiment will be described.

[0052] (Overall Operation Outline) First, referring to FIG. 1, the overall operation outline of the power conversion device 1 will be described. The voltage sensor 11 detects the DC bus voltage Vdc. The switching circuit 14 performs a switching operation based on the drive signal Sctl. The current sensor 15 detects the current Ibat flowing in the path connecting the filter circuit 16 and the power conversion unit 100 in the voltage line L21. The filter circuit 16 removes high-frequency components included in the signal generated by the switching operation in the switching circuit 14. The voltage sensor 17 detects the battery voltage Vbat. The control circuit 20 generates the drive signal Sctl by performing arithmetic processing based on the DC bus voltage Vdc, the current Ibat, and the battery voltage Vbat. The control circuit 20 performs either peak current control or average current control in each period related to the switching period Tpwm.

[0053] (Detailed operation) FIG. 8 shows an operation example of the power conversion device 1 during discharge. (A) shows the waveform of the current Ibat detected by the current sensor 15, (B) shows the waveform of the signal SLP, (C) shows the waveform of the actual current Ibatreal flowing from the filter circuit 16 to the power conversion unit 100 in the voltage line L21, (D) shows the waveform of the signal TP1, (E) shows the time window in the window processing unit 31, (F) shows the waveform of the signal TP2, (G) shows the waveform of the carrier signal CR, (H) shows the duty ratio command value Dbat*, (I) shows the waveform of the signal ZRO, and (J) shows the waveform of the voltage V1.

[0054] The window processing unit 31 uses the time window set by the times Ton, Toff1, and Toff2 shown in FIG. 8(E) to mask the signal TP1 or output the signal TP1 as the signal TP2. In each period related to the switching period Tpwm, the period related to the time Toff1, the period related to the time Ton, and the period related to the time Toff2 are arranged in this order. When the timing of the pulse of the signal TP1 is within the period related to the time Ton, the window processing unit 31 outputs the pulse of this signal TP1 as the pulse of the signal TP2. Further, when the timing of the pulse of the signal TP1 is within the periods related to the times Toff1 and Toff2, the window processing unit 31 masks the pulse of this signal TP1. The times Toff1, Ton, and Toff2 are set, for example, as in the following equations EQ5 to EQ7. [Number] [Number] [Number] Here, a is a constant.

[0055] At the timing t11, when the period related to the switching period Tpwm starts, the PWM processing unit 32 controls the operation of the drive signal generation unit 33, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a high level to a low level (FIG. 8(J)). As a result, the currents Ibat and Ibatreal increase (FIGS. 8(A) and (C)).

[0056] At this timing t11, the PWM processing unit 32 generates a pulse of the signal ZRO (FIG. 8(I)). The slope compensation unit 28 gradually decreases the signal SLP from the peak current command value Ip* at the slope indicated by the slope value Ks based on this signal ZRO (FIG. 8(B)).

[0057] At timing t12, when the signal SLP and the current Ibat coincide with each other, the comparison unit 29 generates a pulse of the signal TP1 (Fig. 8(D)). The slope compensation unit 28 returns the signal SLP to the peak current command value Ip* based on this signal TP1. Since the pulse timing of this signal TP1 is within the period related to the time Ton (Fig. 8(D), (E)), the window processing unit 31 outputs the pulse of this signal TP1 as it is as the pulse of the signal TP2 (Fig. 8(F)). The PWM processing unit 32 controls the operation of the drive signal generation unit 33 based on this signal TP2, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a low level to a high level (Fig. 8(J)). As a result, the currents Ibat, Ibatreal decrease (Fig. 8(A), (C)).

[0058] In this way, during the period of timings t11 to t13, the control circuit 20 controls the switching circuit 14 to perform a switching operation based on the timing t12 indicated by "〇" in Fig. 8(F) where the pulse of the signal TP2 occurs. In this way, during the period of timings t11 to t13, the control circuit 20 performs peak current control.

[0059] Next, at timing t13, when the next period corresponding to the switching period Tpwm starts, the PWM processing unit 32 controls the operation of the drive signal generation unit 33, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a high level to a low level (Fig. 8(J)). As a result, the currents Ibat, Ibatreal increase (Fig. 8(A), (C)). In this example, the actual current Ibatreal increases as it did after timing t11, but the current Ibat detected by the current sensor 15 suddenly increases due to switching noise. That is, the current sensor 15 transiently misdetects, for example, based on switching noise, and the circuit such as an operational amplifier responds, and the detected value suddenly increases.

[0060] At this timing t13, the PWM processing unit 32 generates a pulse of the signal ZRO (FIG. 8(I)). Based on this signal ZRO, the slope compensation unit 28 gradually decreases the signal SLP from the peak current command value Ip* at a slope indicated by the slope value Ks (FIG. 8(B)).

[0061] At timing t14, when the signal SLP and the current Ibat match each other, the comparison unit 29 generates a pulse of the signal TP1 (FIG. 8(D)). Based on this signal TP1, the slope compensation unit 28 returns the signal SLP to the peak current command value Ip*. Since the pulse timing of this signal TP1 is within the period related to the time Toff1 (FIG. 8(D), (E)), the window processing unit 31 masks the pulse of this signal TP1, and the signal TP2 maintains a low level (FIG. 8(F)). Thus, in the periods related to the times Toff1 and Ton, since no pulse occurs in the signal TP2, the PWM processing unit 32 controls the operation of the drive signal generation unit 33 at the timing t15 when the duty ratio command value Dbat* and the carrier signal CR match, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a low level to a high level (FIG. 8(J)).

[0062] Thus, in the period from timing t13 to t16, the control circuit 20 controls the switching circuit 14 to perform a switching operation based on the timing t15 indicated by "〇" in FIGS. 8(G) and (H), where the carrier signal CR and the duty ratio command value Dbat* match. Thus, in the period from timing t13 to t16, the control circuit 20 performs average current control.

[0063] Similarly, in the period from timing t16 to t18, the control circuit 20 controls the switching circuit 14 to perform a switching operation based on the timing t17 indicated by "〇" in FIG. 8(F), where a pulse of the signal TP2 occurs. Thus, in the period from timing t16 to t18, the control circuit 20 performs peak current control.

[0064] During the period from timing t18 to t20, the control circuit 20 controls the switching circuit 14 to perform a switching operation based on the timing t19 indicated by "〇" in FIG. 8(F) where a pulse of the signal TP2 occurs. Thus, during the period from timing t18 to t20, the control circuit 20 performs peak current control.

[0065] In this way, in the power conversion device 1, average current control and peak current control are used in combination. As a result, in the power conversion device 1, for example, even when switching noise occurs, it can operate stably.

[0066] That is, for example, as shown in FIG. 9, when only peak current control is performed, the control circuit controls the switching circuit 14 to perform a switching operation based on the signal TP1 (FIGS. 9(D) and (G)). For example, at timing t23, the current Ibat detected by the current sensor 15 rapidly increases due to switching noise (FIG. 9(A)), and at timing t24, when the signal SLP and the current Ibat match each other (FIGS. 9(A) and (B)), the comparison unit 29 generates a pulse of the signal TP1 (FIG. 9(D)). The control circuit controls the switching circuit 14 to perform a switching operation based on this pulse of the signal TP1. Thus, when only peak current control is performed, there is a possibility of performing a switching operation at an incorrect timing due to switching noise.

[0067] On the other hand, in the power conversion device 1, as shown in FIG. 8, the pulse of the signal TP1 at timing t14 is masked, and average current control is performed during the period from timing t13 to t16, so that malfunction due to switching noise can be suppressed.

[0068] Also, in the power conversion device 1, average current control is performed based on the duty ratio command value Dbat* calculated using Equation EQ2. This duty ratio command value Dbat* includes an element of the ripple value ΔIbat of the current Ibat. This duty ratio command value Dbat* corresponds to the duty ratio of the switching circuit 14 when peak current control is being performed. Therefore, for example, in FIG. 8, the duty ratio of the switching circuit 14 during the periods of timing t11 to t13 when peak current control is being performed and the duty ratio of the switching circuit 14 during the periods of timing t13 to t16 when average current control is being performed can be made almost the same. As a result, the power conversion device 1 can achieve stable operation.

[0069] Also, in the power conversion device 1, since average current control and peak current control are used in combination, stable operation can be achieved even when the control period is longer than the switching period Tpwm.

[0070] Specifically, for example, as shown in FIGS. 10 and 11, when average current control is used, the control period Tctl, which is the period for generating various command values, may be the same as the switching period Tpwm or different from the switching period Tpwm. As shown in FIG. 11, when the control period Tctl is longer than the switching period Tpwm and the control operation and the switching operation are not synchronized, the update timing of the duty ratio command value Dbat* is delayed, so there is a possibility that the feedback control loop will oscillate. Although average current control has the merit of being easy to implement digital control, in such a case, the operation becomes unstable. For example, as shown in FIG. 12, when peak current control is used, even when the control period Tctl is longer than the switching period Tpwm and the control operation and the switching operation are not synchronized, peak current control is performed according to the peak current command value Ip*, so stable operation can be achieved.

[0071] On the other hand, in the power conversion device 1, since average current control and peak current control are used in combination, even when the control period Tctl is longer than the switching period Tpwm and the control operation and the switching operation are not synchronized, the advantages of peak current control can be utilized, and stable operation can be achieved. Thus, in the power conversion device 1, the degree of freedom in setting the control period Tctl and the switching period Tpwm can be increased. For example, since stable operation can be achieved even when the control period Tctl is long, the control circuit 20 can be configured using a microcontroller with low processing power, and costs can be reduced. Also, for example, the switching period Tpwm can be shortened and the frequency of the switching operation can be increased.

[0072] Also, in the power conversion device 1, as shown in FIG. 13, the sampling period Tadc is set to half of the switching period Tpwm, and for example, arithmetic processing is performed based on the average value of two digital values in the switching period Tpwm. Thereby, the influence of switching noise on the circuit operation can be reduced. In this example, the sampling frequency is set to twice the switching frequency, but it is not limited thereto. The sampling frequency may be further increased, and arithmetic processing may be performed based on the average value of a plurality of digital values in the switching period Tpwm. Thereby, for example, the average value of the current Ibat in each period related to the switching period Tpwm can be accurately obtained, and for example, the stability of the feedback operation in average current control can be enhanced. It is not limited thereto, and the sampling frequency can be, for example, equal to or higher than the switching frequency.

[0073] In the above, the operation of the power conversion device 1 during discharging has been described by way of example. The same applies during charging. That is, during discharging, since the current Ibat is positive, as shown in FIGS. 5 and 8, when the current Ibat coincides with the signal SLP, the current Ibat is at a peak. When the timing at which the current Ibat thus peaks is within the period related to the time Ton, the control circuit 20 switches the switching circuit 14 at this timing. On the other hand, during charging, since the current Ibat is negative, as shown in FIG. 7, when the current Ibat coincides with the signal SLP, the current Ibat is at a bottom. Similar to FIG. 8, when the timing at which the current Ibat thus bottoms is within the period related to the time Ton, the control circuit 20 switches the switching circuit 14 at this timing.

[0074] Hereinafter, the operation and action of the power conversion device 1 will be described using some experimental results during discharging. This experiment was conducted using the circuit configuration shown in FIG. 2.

[0075] FIG. 14 shows an experimental example of a power conversion device that performs only peak current control. From the top, the first shows the waveform of the signal TP1, the second shows the waveform of the current Ibat, the third shows the waveform of the current Ibatreal, and the fourth shows the waveform of the gate signal S2 supplied to the gate of the transistor SW2. In FIG. 14, the operation at timings t33 to t36 is the same as the operation at timings t23 to t26 in FIG. 9.

[0076] FIG. 15 shows an experimental example of the power conversion device 1 according to the present embodiment. In FIG. 15, the operation at timings t43 to t46 is the same as the operation at timings t13 to t16 in FIG. 8. FIG. 16 shows the set values of various parameters in the experiments shown in FIGS. 14 and 15.

[0077] As shown in FIG. 14, when only peak current control is performed, for example, at timing t33, the current Ibat detected by the current sensor 15 rapidly increases due to switching noise, and at timing t34, when the signal SLP and the current Ibat match each other, the comparison unit 29 generates a pulse of the signal TP1. The control circuit controls the switching circuit 14 to perform a switching operation based on the pulse of this signal TP1. Thus, when only peak current control is performed, there is a possibility of performing a switching operation at an incorrect timing due to switching noise.

[0078] On the other hand, in the power conversion device 1, as shown in FIG. 15, for example, at timings t43 to t46, no pulse occurs in the signal TP2, and average current control is performed. Thereby, malfunction due to switching noise can be suppressed.

[0079] FIG. 17 shows another experimental example of the power conversion device 1 according to the present embodiment. FIG. 18 shows the set values of various parameters in this experiment. In this experiment, the battery voltage Vbat is set to 180V. As a result, it is expected that the pulse width of the PWM will be 2 μs or less, and it is expected that the pulse of the signal TP1 will occur within 2 μs from the start timing of the switching period Tpwm. In this case, since the timing of the pulse of this signal TP1 is within the period related to the time Toff1, no pulse occurs in the signal TP2. Therefore, in this case, the power conversion device 1 always performs average current control.

[0080] Thus, in the power conversion device 1, the control circuit 20 generates a duty ratio command value Dbat* which is a command value of the duty ratio of the switching circuit 14 based on the current Ibat flowing through the terminals on the power terminals T21 and T22 sides of the power conversion unit 100. The control circuit 20 causes the switching circuit 14 to perform switching in the first transition direction based on the start timing of the switching period of the switching circuit 14. In the switching period, when the first timing at which the current becomes either the peak or the bottom is within the period of the first period (the period related to the time Ton), the control circuit 20 causes the switching circuit 14 to perform switching in the second transition direction based on the first timing. Further, in the switching period, when the first timing is outside the period of the first period (the period related to the time Ton), the control circuit 20 causes the switching circuit 14 to perform switching in the second transition direction based on the second timing corresponding to the duty ratio command value Dbat*. Thereby, in the power conversion device 1, when the first timing is within the period of the first period (the period related to the time Ton) in the switching period, peak current control is performed, and when the first timing is outside the period of the first period (the period related to the time Ton) in the switching period, average current control with a fast response of current control can be performed. Thus, in the power conversion device 1, since peak current control and average current control can be used in combination, it is possible to reduce the possibility of malfunction due to noise while increasing the response of current control.

[0081] [Effect] As described above, in this embodiment, a duty ratio command value, which is a command value for the duty ratio of the switching circuit, is generated based on the current flowing through the terminals on the power terminal T21, T22 sides of the power conversion unit. Further, based on the start timing of the switching period of the switching circuit, the switching circuit is caused to perform switching in the first transition direction. Further, in the switching period, when the first timing at which the current becomes either the peak or the bottom is within the period of the first period, the switching circuit is caused to perform switching in the second transition direction based on the first timing. Further, when the first timing is outside the period of the first period in the switching period, the control circuit causes the switching circuit to perform switching in the second transition direction based on the second timing corresponding to the duty ratio command value. Thereby, while accelerating the response of the current control, it is possible to reduce the possibility of malfunction due to noise.

[0082] [Modification Example 1] In the above embodiment, the DC power supply PDC is connected to the power terminals T11 and T12, and the battery BT is connected to the power terminals T21 and T22, but the present invention is not limited thereto. Instead, for example, a DC power supply may be connected to the power terminals T11 and T12, and another DC power supply may be connected to the power terminals T21 and T22. Further, for example, a battery may be connected to the power terminals T11 and T12, and another battery may be connected to the power terminals T21 and T22.

[0083] [Modification Example 2] In the above embodiment, the present technology is applied to a non-insulated bidirectional DC / DC converter, but the present invention is not limited thereto. For example, the present technology may be applied to an insulated bidirectional DC / DC converter, a non-insulated unidirectional DC / DC converter, or an insulated unidirectional DC / DC converter. Hereinafter, an example in which the present technology is applied to a non-insulated unidirectional DC / DC converter will be described.

[0084] FIG. 19 shows a configuration example of the power conversion device 1A according to this modified example. This power conversion device 1A is a non-insulated unidirectional DC / DC conversion device and is configured using a boost chopper circuit. The power conversion device 1A is configured to discharge the battery BT by supplying power from the power terminals T21 and T22 to the power terminals T11 and T12. In FIG. 19, the illustration of the control circuit 20 is omitted. The power conversion unit 100 of the power conversion device 1A has a switching circuit 14A. The switching circuit 14A has a transistor SW2 and a diode D1. The anode of the diode D1 is connected to the node N1, and the cathode is connected to the voltage line L11. The power conversion device 1A discharges the battery BT by performing a unidirectional power conversion operation. In this power conversion device 1A, the control circuit 20 always performs constant control of the DC bus voltage Vdc during discharge. Therefore, the switching unit 25 of the control circuit 20 supplies the current command value Ibat* supplied from the DC bus voltage control unit 22 to the current control unit 27.

[0085] [Modified Example 3] In the above embodiment, as shown in FIG. 7, during charging, the signal SLP was gradually decreased from the peak current command value Ip* at the slope indicated by the slope value Ks, but it is not limited to this. Instead, for example, as shown in FIG. 20, the absolute value of the peak current command value Ip* may be set to a value larger than the absolute value of the current Ibat, and the signal SLP may be gradually increased from the peak current command value Ip* at the slope indicated by the slope value Ks. The control circuit according to this modified example has a slope compensation unit and a comparison unit, similar to the control circuit 20 (FIG. 3) according to the above embodiment. The slope compensation unit gradually increases the signal SLP from the peak current command value Ip* at the slope indicated by the slope value Ks based on the signal ZRO (FIG. 20(B)). When the signal SLP and the current Ibat match, the comparison unit generates a pulse of the signal TP1 (FIG. 20(C)). The slope compensation unit returns the signal SLP to the peak current command value Ip* based on this signal TP1.

[0086] During charging, since the current Ibat is negative, as shown in FIG. 20, when the current Ibat coincides with the signal SLP, the current Ibat is at its peak. Similar to FIG. 8, the control circuit 20 switches the switching circuit 14 at this timing when the timing at which the current Ibat peaks is within the period related to the time Ton.

[0087] [Modification Example 4] In the above embodiment, the control is performed based on the current Ibat between the power conversion unit 100 and the filter circuit 16. However, the present invention is not limited to this. Instead, for example, the control may be performed based on the current flowing through the terminal on the power terminal T11 side of the power conversion unit 100. Hereinafter, the power conversion device 1B according to this modification example will be described in detail.

[0088] FIG. 21 shows a configuration example of the power conversion device 1B according to this modification example. The power conversion device 1B includes a current sensor 13B and a control circuit 20B.

[0089] The current sensor 13B is provided on the voltage line L11 and is configured to detect the current Idc flowing through the terminal on the power terminal T11 side of the power conversion unit 100. The current sensor 13B includes, for example, a resistance element and detects the current flowing through this resistance element based on the voltage between both ends of the resistance element. One end of the current sensor 13B is connected to one end of the voltage sensor 11, one end of the capacitor 12, and the power terminal T11, and the other end is connected to the power conversion unit 100. The current sensor 13B detects the current Idc with a polarity that becomes positive when flowing from the capacitor 12 and the power terminal T11 toward the power conversion unit 100. The current sensor 13B supplies the detection result of the current Idc to the control circuit 20B.

[0090] FIG. 22 shows a configuration example of the control circuit 20B. The control circuit 20B includes a DC bus voltage control unit 22B, a battery voltage control unit 23B, a division unit 24B, a current ripple calculation unit 26B, a current control unit 27B, a slope compensation unit 28B, and a comparison unit 29B.

[0091] When the control circuit 20B performs constant control of the DC bus voltage Vdc during discharge, the DC bus voltage control unit 22B is configured to generate a current command value Idc*, which is a command value of the current Idc, by performing control such that the DC bus voltage Vdc becomes the same as the DC bus voltage command value Vdc* based on the DC bus voltage command value Vdc* and the DC bus voltage Vdc.

[0092] When the control circuit 20B performs constant control of the battery voltage Vbat during charging, the battery voltage control unit 23B is configured to generate a current command value Idc*, which is a command value of the current Idc, by performing control such that the battery voltage Vbat becomes the same as the battery voltage command value Vbat* based on the battery voltage command value Vbat* and the battery voltage Vbat.

[0093] When the control circuit 20B performs constant control of the charge / discharge power, the division unit 24B is configured to generate a current command value Idc*, which is a command value of the current Idc, by dividing the power command value Pcp* by the DC bus voltage Vdc.

[0094] When the control circuit 20B performs constant control of the charge / discharge current, the switching unit 25 supplies the current command value Icc* as the current command value Idc* to the current control unit 27B. When the control circuit 20B performs constant control of the charge / discharge power, the switching unit 25 supplies the current command value Idc* supplied from the division unit 24B to the current control unit 27B. When the control circuit 20B performs constant control of the battery voltage Vbat during charging, the switching unit 25 supplies the current command value Idc* supplied from the battery voltage control unit 23B to the current control unit 27B. When the control circuit 20B performs constant control of the DC bus voltage Vdc during discharge, the switching unit 25 supplies the current command value Idc* supplied from the DC bus voltage control unit 22B to the current control unit 27B. The current command value Idc* is a negative value during charging and a positive value during discharge.

[0095] The current ripple calculation unit 26B is configured to generate a ripple value ΔIdc of the current Idc based on the DC bus voltage Vdc and the battery voltage Vbat.

[0096] The current control unit 27B is configured to calculate a duty ratio command value Dbat* which is a command value of the duty ratio of the switching circuit 14 by controlling the current Idc to be the same as the current command value Idc* based on the current command value Idc* supplied from the switching unit 25, the current Idc detected by the current sensor 13B, the ripple value ΔIdc of the current Idc, the DC bus voltage Vdc, and the battery voltage Vbat. Further, the current control unit 27B is also configured to calculate a peak current command value Ip* which is a command value of the peak current value of the current Idc and a slope value Ks used for peak current control.

[0097] The slope compensation unit 28B is configured to generate a signal SLP that gradually decreases from the peak current command value Ip* at a slope indicated by the slope value Ks based on the peak current command value Ip*, the slope value Ks, the signal ZRO, and the signal TP1.

[0098] The comparison unit 29B is configured to compare the current Idc detected by the current sensor 13B with the signal SLP and generate a signal TP1 according to the comparison result.

[0099] Similar to the power conversion device 1 according to the above embodiment, the power conversion device 1B can perform average current control and peak current control.

[0100] FIG. 23 shows an example of peak current control during charging. (A) shows the waveform of the current Idc, (B) shows the waveform of the signal SLP, (C) shows the waveform of the signal TP1, (D) shows the waveform of the carrier signal CR, (E) shows the waveform of the signal ZRO, and (F) shows the waveform of the voltage V1.

[0101] When performing peak current control, at the start timing of the period related to the switching period Tpwm, the PWM processing unit 32 controls the operation of the drive signal generation unit 33, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a low level to a high level (Fig. 23(F)). As a result, the current Idc starts to flow and the current value increases (Fig. 23(A)). Also, at this start timing, the PWM processing unit 32 generates a pulse of the signal ZRO (Fig. 23(E)). The slope compensation unit 28B gradually decreases the signal SLP from the peak current command value Ip* at the slope indicated by the slope value Ks based on this signal ZRO (Fig. 23(B)). When the signal SLP and the current Idc match, the comparison unit 29B generates a pulse of the signal TP1 (Fig. 23(C)). The slope compensation unit 28B returns the signal SLP to the peak current command value Ip* based on this signal TP1. In this example, the window processing unit 31 outputs the pulse of this signal TP1 as it is as the pulse of the signal TP2. The PWM processing unit 32 controls the operation of the drive signal generation unit 33 based on this signal TP2, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14 performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a high level to a low level (Fig. 23(F)). As a result, the current Idc becomes "0" (Fig. 23(A)).

[0102] Note that in this example, the peak current control during charging has been described, but the same applies to discharging.

[0103] For example, when this modified example is applied to a unidirectional DC / DC converter that discharges the battery BT, this control circuit 20B always performs constant control of the DC bus voltage Vdc during discharging. Therefore, the switching unit 25 of the control circuit 20B supplies the current command value Idc* supplied from the DC bus voltage control unit 22B to the current control unit 27B.

[0104] Also, for example, when this modification is applied to a unidirectional DC / DC converter that charges the battery BT, this control circuit 20B always performs constant control of the battery voltage Vbat during charging. Therefore, the switching unit 25 of the control circuit 20B supplies the current command value Idc* supplied from the battery voltage control unit 23B to the current control unit 27B.

[0105] In this way, in the power conversion device 1B, the control circuit 20B is configured to perform control based on the current Idc flowing through the terminals on the side of the power terminals T11 and T12 of the power conversion unit 100. The current sensor 13B that detects this current Idc can use a cheaper one compared to the current sensor 15 according to the above embodiment. Thereby, in the power conversion device 1B, the cost can be reduced.

[0106] [Modification 5] In the above embodiment, the filter circuit 16 is provided, but the present invention is not limited to this, and instead, for example, the filter circuit 16 may not be provided. Hereinafter, the power conversion device 1C according to this modification will be described in detail.

[0107] FIG. 24 shows a configuration example of the power conversion device 1C. The power conversion device 1C is an isolated unidirectional DC / DC converter and is configured using a flyback circuit. The power conversion device 1C is configured to discharge the battery BT by supplying power from the power terminals T21 and T22 toward the power terminals T11 and T12. The power conversion device 1C includes a diode 43C, a power conversion unit 100C, and a capacitor 16C. The anode of the diode 43C is connected to the power conversion unit 100C, and the cathode is connected to one end of the voltage sensor 11, one end of the capacitor 12, and the power terminal T11.

[0108] FIG. 25 shows a configuration example of the power conversion unit 100C. The power conversion unit 100C includes a transformer 44C and a switching circuit 14C. The transformer 44C has a first winding and a second winding. One end of the first winding is connected to the anode of the diode 43C, and the other end is connected to the reference voltage line L12. One end of the second winding is connected to the other end of the current sensor 15, and the other end is connected to the switching circuit 14C. The switching circuit 14C includes a transistor SW2. The collector of the transistor SW2 is connected to the other end of the second winding of the transformer 44C, and the emitter is connected to the reference voltage line L22. In this example, the voltage between the collector and emitter of the transistor SW2 is the voltage V1.

[0109] Similar to the power conversion device 1 according to the above embodiment, the power conversion device 1C can perform average current control and peak current control.

[0110] FIG. 26 shows an example of peak current control during discharge. (A) shows the waveform of the current Ibat, (B) shows the waveform of the signal SLP, (C) shows the waveform of the signal TP1, (D) shows the waveform of the carrier signal CR, (E) shows the waveform of the signal ZRO, and (F) shows the waveform of the voltage V1.

[0111] When performing peak current control, at the start timing of the period related to the switching period Tpwm, the PWM processing unit 32 controls the operation of the drive signal generation unit 33, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14C performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a high level to a low level (Fig. 26(F)). In the case of the circuit configuration shown in Fig. 25, when the transistor SW2 changes from the off state to the on state, the voltage V1 changes from a high level to a low level. As a result, the current Ibat increases (Fig. 26(A)). Also, at this start timing, the PWM processing unit 32 generates a pulse of the signal ZRO (Fig. 26(E)). The slope compensation unit 28 gradually decreases the signal SLP from the peak current command value Ip* at the slope indicated by the slope value Ks based on this signal ZRO (Fig. 26(B)). When the signal SLP and the current Ibat match, the comparison unit 29 generates a pulse of the signal TP1 (Fig. 26(C)). The slope compensation unit 28 returns the signal SLP to the peak current command value Ip* based on this signal TP1. In this example, the window processing unit 31 outputs the pulse of this signal TP1 as it is as the pulse of the signal TP2. The PWM processing unit 32 controls the operation of the drive signal generation unit 33 based on this signal TP2, and the drive signal generation unit 33 generates a drive signal Sctl. The switching circuit 14C performs a switching operation based on this drive signal Sctl, and the voltage V1 changes from a low level to a high level (Fig. 26(F)). In the case of the circuit configuration shown in Fig. 25, when the transistor SW2 changes from the on state to the off state, the voltage V1 changes from a low level to a high level. As a result, the current Ibat becomes "0" (Fig. 26(A)).

[0112] Thus, the power conversion device 1C can operate based on the current Ibat having a trapezoidal waveform.

[0113] [Modification Example 6] In the above embodiment, the control circuit 20 performs calculations using all of the equations EQ1 to EQ7 so that it can operate with various DC bus voltages Vdc and various battery voltages Vbat. However, for example, when the voltage ranges of the DC bus voltage Vdc and the battery voltage Vbat are narrow, calculations may be performed using only some of the equations EQ1 to EQ7. Specifically, for example, the control circuit 20 may perform calculations using a fixed ripple value ΔIbat instead of using Equation EQ1. Also, the control circuit 20 may perform calculations using a fixed slope value Ks instead of using Equations EQ3 and EQ4.

[0114] The present invention has been described above with reference to the embodiments and modification examples, but the present invention is not limited to these embodiments and the like, and various modifications are possible.

[0115] For example, each value such as the DC bus voltage Vdc, the battery voltage Vbat, the switching period Tpwm, and the sampling frequency fs is an example and may be changed as appropriate.

Description of Reference Numerals

[0116] 1, 1A, 1B, 1C... power conversion device, 11... voltage sensor, 12... capacitor, 13B... current sensor, 14, 14A, 14C... switching circuit, 15, 15B... current sensor, 16, 16B... filter circuit, 16L... inductor, 16C... capacitor, 17... voltage sensor, 20, 20B... control circuit, 21... command value generation unit, 22, 22B... DC bus voltage control unit, 23, 23B... battery voltage control unit, 24, 24B... division unit, 25... switching unit, 26, 26B... current ripple calculation unit, 27, 27B... current control unit, 28, 28B... slope compensation unit, 29, 29B... comparison unit, 31... window processing unit, 32... PWM processing unit, 33... drive signal generation unit, 34... carrier signal generation unit, 39... setting unit, 43C... diode, 44C... transformer, 100, 100C... power conversion unit, Ibat... current, BT... battery, CR... carrier signal, Dbat*... duty ratio command value, D1... diode, fs... sampling frequency, Ibat... current, Ibat*... current command value, Icc*... current command value, Idc... current, Idc*... current command value, Ip*... peak current command value, Ks... slope value, L11, L21... voltage line, L12, L22... reference voltage line, N1... node, Pcp*... power command value, PDC... DC power supply, Sctl... drive signal, SLP... signal, SW1, SW2... transistor, S1, S2... gate signal, Tadc... sampling period, Tctl... control period, Td... dead time, Ton, Toff1, Toff2... time, Tpwm... switching period, TP1, TP2... signal, T11, T12, T21, T22... power terminal, Vbat... battery voltage, Vbat*... battery voltage command value, Vdc... DC bus voltage, Vdc*... DC bus voltage command value, V1... voltage, ZRO... signal, ΔIbat, ΔIdc... ripple value.

Claims

1. A first power terminal, a power conversion unit connected to the first power terminal and including a switching circuit capable of alternately performing switching in a first transition direction and switching in a second transition direction, a second power terminal led to the power conversion unit, and a control circuit capable of controlling the operation of the switching circuit are provided, wherein the control circuit can generate a command value of a duty ratio of the switching circuit based on a current flowing through either one of a terminal on the first power terminal side of the power conversion unit and a terminal on the second power terminal side of the power conversion unit, can cause the switching circuit to perform switching in the first transition direction based on a start timing during a switching period of the switching circuit, and in the switching period, when a first timing at which the current becomes either a peak or a bottom is a timing within a preset first period during the switching period, peak current control can be performed by causing the switching circuit to perform switching in the second transition direction based on the first timing, and when the first timing is a timing outside the first period, average current control can be performed by causing the switching circuit to perform switching in the second transition direction based on a second timing corresponding to the command value of the duty ratio a power conversion device.

2. The switching period has a second period starting from the start timing, which is different from the first period, and the first period is a period following the second period The power conversion device according to claim 1.

3. The switching period has a third period ending at an end timing of the switching period, which is different from the first period and the second period, and the first period is a period between the second period and the third period The power conversion device according to claim 2.

4. The control circuit can calculate a ripple amount of the current based on a first voltage at the first power terminal and a second voltage at the second power terminal, and can generate the command value of the duty ratio based on the current and the ripple amount The power conversion device according to any one of claims 1 to 3.

5. The control circuit can control the operation of the switching circuit such that the power conversion device performs a first operation of supplying power from the first power terminal to the second power terminal and a second operation of supplying power from the second power terminal to the first power terminal. The power conversion device according to any one of claims 1 to 4. **Claim 6** The power conversion unit is connected to the switching circuit and further includes a transformer having a first winding and a second winding. The power conversion device according to any one of claims 1 to 5.

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