Rectification circuit

Abstract

To provide a circuit obtained by integrating an automatic matching circuit with a rectification circuit in a simpler configuration.SOLUTION: A series circuit of switching elements S1 and S2 is connected between DC output terminals in a rectification circuit 11, and each of diodes D1 and D2 is connected between a drain and a source of each of the switching elements S1 and S2. A series resonance filter 12 whose resonance frequency is set equal to an AC power supply frequency is connected between one end of an AC input terminal and a common connection point of the switching elements S1 and S2, and a control part 13 makes the switching elements S1 and S2 perform mutually switching at the same frequency as AC power supply frequency thereof, at each duty ratio of 50%, and at a phase difference of 180 degrees. Further, the control part 13 changes a phase of a gate signal Vg1 for switching the switching element S1 or a phase of an inter-terminal voltage Vds2 of the switching element S2 according to a generation reactance ΔX generated subsequently on a transmission route of power.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present invention relates to a circuit for rectifying AC power into DC power. [Background technology]

[0002] Wireless Power Transfer (WPT) systems, such as those using electric field coupling or magnetic field coupling, are equipped with matching circuits that cancel reactance components in the power transmission path or match impedance, and resonant circuits that resonate with the LC value of the coupler and the AC power supply frequency, in order to improve power transmission efficiency and suppress power reflection.

[0003] The resonant frequency of the coupler and the reactance component on the power transmission path may fluctuate due to dimensional changes over time of the coupler, movement of the power receiving side, etc. If the matching circuit or resonant circuit is composed of LC elements with fixed constants, it will not be able to handle the generated reactance component, which may cause power reflection and degrade power transmission performance.

[0004] To address this issue, methods exist that add a mechanical configuration with a variable LC value or an electronic automatic matching circuit consisting of a switching element and an LC element to track fluctuations in the reactance component of the power transmission path and cancel its effects. However, these methods pose a concern as they increase the circuit size. In particular, the receiving circuit is often subject to size and weight restrictions due to the assumption that it will be installed on mobile or aircraft vehicles, making it necessary to reduce its size. Therefore, a circuit that integrates an automatic matching circuit and a rectifier circuit has been proposed to achieve both power transmission performance and a compact receiving circuit.

[0005] Patent Document 1 discloses a configuration that combines bridgeless PFC (Power Factor Correction) and a magnetic energy recovery switch (MERS). This method improves the power factor by switching SW1 and SW2 at a frequency higher than that of the AC power supply, i.e., rectifies while suppressing the influence of the reactance component, and at the same time, achieves soft switching by SW1 to SW3 using capacitor CM and FET SW, resulting in highly efficient operation.

[0006] Non-Patent Document 1 discloses a rectifier circuit that includes +jX or -jX, a rectifier unit that includes two switches and a capacitor, and a DC-DC converter. The converter controls V1 and V2, and controls the contribution of +jX and -jX to control impedance Zr, thereby canceling the reactance on the power transmission path. [Prior art documents] [Patent documents]

[0007] [Patent Document 1] Japanese Patent Application Laid-Open No. 2014-36573 [Non-patent literature]

[0008] [Non-Patent Document 1] IEEE JOURNAL OF EMERGING AND SELECTED TOPICS IN POWER ELECTRONICS, VOL.8, NO.3, SEPTEMBER 2020 Summary of the Invention [Problem to be solved by the invention]

[0009] The PFC used in Patent Document 1 generally requires SW1 to SW3 to be controlled by switching them at a frequency several hundred times higher than the frequency of the AC power supply, making it difficult to apply to contactless power supply where the power supply frequency is 85 kHz to 13.56 MHz.

[0010] Furthermore, in the configuration of Non-Patent Document 1, the switching frequency of the rectifier is the same as the frequency of the AC power supply, but two sets of combinations of ±jX parts, rectifiers, and converters are required, which raises concerns about an increase in the circuit size. The present invention has been made in view of the above circumstances, and an object of the present invention is to provide a circuit that integrates an automatic matching circuit and a rectifier circuit with a simpler configuration. [Means for solving the problem]

[0011] According to the rectifier circuit of claim 1, a series circuit of first and second switching elements is connected between DC output terminals, and a diode is connected between the low-voltage side terminal and the high-voltage side terminal of each switching element. A series resonant filter having a resonant frequency set equal to the AC power supply frequency is connected between one end of the AC input terminal and a common connection point of the first and second switching elements, and a control unit switches the first and second switching elements at the same frequency as the AC power supply frequency, with a duty ratio of 50% each and a phase difference of 180 degrees.

[0012] Furthermore, the control unit controls the phase of the drive signal for switching the first switching element or the phase of the voltage across the second switching element to have a predetermined phase difference from the phase of the input AC voltage, depending on the generated reactance that occurs later on in the power transmission path. Specifically, if there is no generated reactance, the control unit turns off the first and second switching elements, or makes the phase of the drive signal for switching the first switching element or the phase of the voltage across the second switching element the same as the phase of the AC voltage. If the generated reactance is inductive, the control unit delays the phase of the drive signal for the first switching element or the phase of the voltage across the second switching element by a predetermined value relative to the phase of the AC voltage. If the generated reactance is capacitive, the control unit advances the phase of the drive signal for the first switching element or the phase of the voltage across the second switching element by a predetermined value relative to the phase of the AC voltage.

[0013] That is, the control unit controls the phase of the drive signal that switches the first switching element or the phase of the voltage between the terminals of the second switching element so as to cancel out the effect of reactance that occurs subsequently on the power transmission path. This allows AC power to be converted to DC power while automatically matching impedance. Furthermore, since the switching frequency is the same as the AC power supply frequency, it is easy to apply to a contactless power transfer system and switching loss can be reduced. Furthermore, the circuit size can be made smaller than the configuration disclosed in Non-Patent Document 1.

[0014] According to the rectifier circuit of claim 2, the predetermined value by which the phase of the drive signal for the first switching element or the phase of the terminal voltage of the second switching element is delayed or advanced relative to the phase of the AC voltage is set to 90°. With this configuration, if the AC voltage and AC current are Vin and Iin, and the terminal voltage of the smoothing capacitor is Vo, the voltage Vo and the operation of the first and second switching elements function as an auxiliary power supply that is 90° out of phase with the voltage Vin. When the generated reactance is capacitive, the voltage Vo advances the phase of the current Iin, and when the generated reactance is inductive, the voltage Vo delays the phase of the current Iin. When the charging and discharging of the smoothing capacitor are balanced, the phases of the voltage Vin and the current Iin are aligned. This allows efficient impedance matching. [Brief explanation of the drawings]

[0015] [Figure 1] FIG. 1 is a diagram illustrating a configuration of a rectifier circuit according to a first embodiment. [Figure 2] A diagram showing variations of the configuration shown in Figure 1 [Figure 3] A diagram showing the circuit configuration of the current-voltage phase difference detection unit [Figure 4] Diagram showing the configuration of the control unit [Figure 5] Functional block diagram showing the configuration of a wireless power supply system [Figure 6] A signal waveform diagram showing a state in which the switching elements S1 and S2 are turned off when no generated reactance exists. [Figure 7] A signal waveform diagram showing the state in which the pulse phase of the gate signal Vg1 is made in phase with the AC voltage phase when no generated reactance exists. [Figure 8] Signal waveform diagram showing an unsteady state in which the pulse phase of the gate signal Vg1 is advanced and delayed relative to the AC voltage phase when the generated reactance is inductive. [Figure 9] Signal waveform diagram showing the steady state [Figure 10] This is a signal waveform diagram showing an unsteady state in which the pulse phase of the gate signal Vg1 is advanced from the AC voltage phase when the generated reactance is capacitive. [Figure 11] Signal waveform diagram showing the steady state [Figure 12] Voltage-current vector diagram when the generated reactance is inductive [Figure 13] A diagram showing an unsteady state that explains why a phase difference of 90° is appropriate when the generated reactance is inductive. [Figure 14] The figure shows the steady state [Figure 15] Voltage-current vector diagram when the generated reactance is capacitive [Figure 16] This diagram shows an unsteady state that explains why a phase difference of 90° is appropriate when the generated reactance is capacitive. [Figure 17] The figure shows the steady state [Figure 18] FIG. 10 is a diagram illustrating how the phase difference of the drain-source voltage Vds of the switching element S2 with respect to the phase of the AC voltage Vin is set to 90° when adding dead time to the gate signals Vg1 and Vg2. [Figure 19] FIG. 10 is a signal waveform diagram illustrating a second embodiment in which the generated reactance is inductive and the dead time provided between the gate signals Vgs1 and Vgs2 is shorter than the fall time of the voltage Vds1. [Figure 20] A signal waveform diagram showing a case where the dead time is equal to or longer than the fall time of the voltage Vds1. [Figure 21]A signal waveform diagram showing the case where the dead time applied is shorter than the fall time of the current Is1 when the generated reactance is capacitive. [Figure 22] A signal waveform diagram showing a case where the dead time is equal to or longer than the fall time of the current Is1. [Figure 23] FIG. 10 is a diagram showing the circuit configuration of a current-voltage phase difference detection unit according to the third embodiment (part 1); [Figure 24] A diagram showing the change in output voltage VDC relative to the current-voltage phase difference [Figure 25] Circuit configuration of current-voltage phase difference detection unit (part 2) [Figure 26] A diagram showing the change in output voltage VDC relative to the current-voltage phase difference [Figure 27] FIG. 10 is a diagram illustrating a configuration of a rectifier circuit according to a fourth embodiment, in which the output source of the reference signal is different. [Figure 28] FIG. 10 is a diagram illustrating a configuration of a rectifier circuit according to a fifth embodiment. [Figure 29] FIG. 10 is a diagram illustrating a configuration of a rectifier circuit according to a sixth embodiment. [Figure 30] A signal waveform diagram showing the state in which the phase difference of the gate signal Vg1 relative to the phase of the AC voltage Vin is swept according to the power Po output to the load. [Figure 31] Flowchart showing control details DETAILED DESCRIPTION OF THE INVENTION

[0016] (First embodiment) A first embodiment will be described below. As shown in Fig. 5, a typical contactless power transfer system has an AC power source 1, an automatic matching circuit 2, and a matching circuit / resonance circuit 3 on the power transmitting side, and a matching circuit / resonance circuit 4, an automatic matching circuit 5, a rectifier circuit 6, and a load 7 on the power receiving side. A coupler 8 is disposed between the matching circuit / resonance circuits 3 and 4, and contactless power is transferred from the power transmitting side to the power receiving side via this coupler 8 by electric field coupling or magnetic field coupling. The configuration of this embodiment corresponds to the automatic matching circuit 5 and rectifier circuit 6 on the power receiving side.

[0017] As shown in FIG. 1, the rectifier circuit 11 of this embodiment includes a smoothing capacitor Cm connected across a load 7, and a series circuit of switching elements S1 and S2. The switching elements S1 and S2 are, for example, N-channel MOSFETs, with diodes D1 and D2 connected between their drains and sources, respectively. These diodes D1 and D2 may also be body diodes of FETs. The switching elements S1 and S2 are first and second switching elements, respectively. The drains and sources are high-voltage side terminals and low-voltage side terminals, respectively.

[0018] A resonant circuit 12 consisting of a series circuit of a capacitor Cf and a coil Lf is connected between one end of the AC input terminal on the power receiving side and the common connection point of the switching elements S1 and S2. The switching operations of the switching elements S1 and S2 are controlled by a control unit 13. The control unit 13 outputs gate signals Vg1 and Vg2, which are drive signals, to the gates of the switching elements S1 and S2, respectively.

[0019] A current-voltage phase difference detector 14 is connected to both ends of the AC power supply 1. The current-voltage phase difference detector 14 detects the phase difference between the current and voltage of the AC power supply 1 and transmits the detection result to the controller 13, for example, as data. The data includes a reference signal that is output in synchronization with the zero-crossing point of the voltage. As shown in FIG. 2, the current-voltage phase difference detector 14 may be connected in parallel with the switching element S2 on the power receiving side, between the resonant circuit 12 and the switching element S2. Also, "ΔX" shown in the figure indicates reactance that subsequently occurs in the power transmission path due to a change over time, and will hereinafter be referred to as the generated reactance ΔX.

[0020] 3, in current-voltage phase difference detection unit 14, a series circuit of capacitor 15 and a primary coil of transformer 16 is connected between terminals 1a and 1b of AC power supply 1. Furthermore, a primary coil of transformer 17 is connected to terminal 1a. Both ends of the secondary coil of transformer 16 are Vo_H and Vo_L, respectively, and both ends of the secondary coil of transformer 17 are Io_H and Io_L, respectively.

[0021] Between terminals Vo_H and Vo_L, a series circuit of resistor element 18V and resistor elements 19V and 20V is connected. Between the power supply and ground, a series circuit of resistor elements 21V and 22V is connected, and their common connection point is connected to the common connection point of resistor elements 19V and 20V. Terminals Vo_H and Vo_L are connected to the non-inverting input terminal and inverting input terminal of amplifier 23V, respectively, and the output terminal of amplifier 23V is connected to clock terminal CLK of D flip-flop 24. The configuration connected between terminals Io_H and Io_L is symmetrical to the above, and corresponding components are indicated by adding "I" instead of "V". The output terminal of amplifier 23I is connected to data terminal D of D flip-flop 24.

[0022] According to the current-voltage phase difference detection unit 14 configured as described above, when the current of the AC power supply 1 is in a leading phase with respect to the voltage, the output terminal Q of the D flip-flop 24 becomes "1", and when it is in a lagging phase with respect to the voltage, the output terminal Q bar becomes "1".

[0023] As shown in Fig. 4, the control unit 13 includes two selection circuits 25(1) and 25(2) and delay circuits 26 to 28. The delay phase amounts provided by these delay circuits 26 to 28 are 90°, 270°, and 180° of the AC power supply cycle, respectively. The movable contacts of the selection circuits 25(1) and 25(2) are connected to the gates of the switching elements S1 and S2, respectively. A reference signal synchronized with the voltage phase of the AC power supply 1 is applied to the fixed contact (1) of the selection circuit 25(1).

[0024] The output terminal of a delay circuit 28 is connected to the fixed contact (1) of the selection circuit 25(2). The output terminal of a delay circuit 26 is connected to the fixed contact (2) of the selection circuit 25(1) and the fixed contact (3) of the selection circuit 25(2). The output terminal of a delay circuit 27 is connected to the fixed contact (3) of the selection circuit 25(1) and the fixed contact (2) of the selection circuit 25(2). The switching of the movable contact of the selection circuit 25 is performed according to the determination result of the generated reactance ΔX. If there is no phase difference between the current and the voltage, ΔX = 0, and the fixed contact (1) is selected. If the current is in a lagging phase with respect to the voltage, ΔX > 0, and the fixed contact (2) is selected. If the current is in a leading phase with respect to the voltage, ΔX < 0, and the fixed contact (3) is selected.

[0025] Next, the operation of this embodiment will be described. <When ΔX=0> When no reactance is generated and there is no phase difference between the current and voltage, as shown in Figure 6, both switching elements S1 and S2 are turned off and diodes D1 and D2 are turned on to perform rectification. Alternatively, as shown in Figure 7, the phase of the pulse of the gate signal Vg1 of the switching element S1 is made the same as the AC power supply voltage. In other words, the rising edge of the pulse of the gate signal Vg1 is made to coincide with the zero-crossing point when the polarity of the AC voltage transitions from negative to positive. The pulse of the gate signal Vg2 of the switching element S2 has the opposite phase. Note that in the figure, Vin is the voltage of the AC power supply 1, and Vo is the output voltage of the rectifier circuit 11.

[0026] <When ΔX> 0> If the generated reactance is inductive, the phase of the pulse of the gate signal Vg1 is delayed by a predetermined amount Φ=90° from the phase of the AC power supply voltage, as shown in Figure 8. Figure 9 shows the steady state after the switching operation shown in Figure 8 has been performed.

[0027] <When ΔX<0> If the generated reactance is capacitive, the phase of the gate signal Vg1 pulse is set to lead the phase of the AC power supply voltage by a predetermined amount Φ=90°, as shown in Fig. 10. Fig. 11 shows the steady state after the switching operation shown in Fig. 10 has been performed.

[0028] Here, we will explain the appropriateness of setting the phase adjustment amount to 90° when ΔX > 0 and ΔX < 0. A vector diagram for the case when ΔX > 0 is shown in FIG. 12. In the diagram, Iin' is the AC current flowing in a non-steady state. As shown in FIG. 13, when switching element S1 is turned on with a 90° delay phase relative to the AC voltage phase and a duty ratio of 50%, smoothing capacitor Cm is charged and discharged by AC current Iin during half a cycle of the on period. As a result, charge is accumulated in smoothing capacitor Cm by the phase difference of current Iin, and output voltage Vo is obtained.

[0029] The output voltage Vo and the switching operation of the switching elements S1 and S2 function as an auxiliary power supply that is 90° phase-delayed from the AC voltage Vin, causing the output voltage Vo to lead the phase of the AC current Iin. The voltage Vo then rises until the phases of the current Iin and voltage Vin match. Eventually, when the charging and discharging of the smoothing capacitor Cm are balanced and the voltage Vo stabilizes, the phases of the current Iin and voltage Vin match, as shown in Figure 14.

[0030] In other words, when the switching element S1 is operated with a 90° delay phase relative to the phase of the AC voltage Vin and with a duty ratio of 50%, the current Iin and voltage Vin will be in phase when the charging and discharging of the smoothing capacitor Cm by the current Iin are balanced.

[0031] 15 to 17 are diagrams equivalent to FIGS. 12 to 14 for the case where ΔX<0. In this case, the output voltage Vo and the switching operation of switching elements S1 and S2 function as an auxiliary power supply that is 90° ahead of the AC voltage Vin, causing the output voltage Vo to lag the phase of the AC current Iin. The voltage Vo then rises until the phases of the current Iin and voltage Vin match. Eventually, when the charging and discharging of smoothing capacitor Cm are balanced and voltage Vo stabilizes, the phases of the current Iin and voltage Vin will match, as shown in FIG. 17.

[0032] The above description deals with the case where no dead time is applied to the gate signals Vg1 and Vg2. In contrast, if the dead time α is applied by shifting the phase of the gate signal Vg1, as shown in FIG. 18, the phase difference between the voltage Vin and the gate signal Vg1 becomes (90° + α). Even in this case, the phase difference between the drain-source voltage Vds2 of the switching element S2 and the voltage Vin remains at 90°. Therefore, when applying a dead time as described above, it is sufficient to control the phase difference between the voltage Vin and the voltage Vds2 to be 90°.

[0033] As described above, according to this embodiment, in the rectifier circuit 11, a series circuit of switching elements S1 and S2 is connected between the DC output terminals, and diodes D1 and D2 are connected between the drain and source of each switching element S1 and S2, respectively. A series resonant filter 12, whose resonant frequency is set equal to the AC power supply frequency, is connected between one end of the AC input terminal and the common connection point of the switching elements S1 and S2, and a control unit 13 switches the switching elements S1 and S2 at the same frequency as the AC power supply frequency, with a duty ratio of 50% each and a phase difference of 180 degrees.

[0034] Furthermore, the control unit 13 controls the phase of the gate signal Vg1 that switches the switching element S1 or the phase of the voltage Vds2 between the terminals of the switching element S2 so that it has a predetermined phase difference from the phase of the input AC voltage Vin, depending on the generated reactance ΔX that occurs later on in the power transmission path. Specifically, if the generated reactance ΔX does not exist, the control unit 13 turns off the switching elements S1 and S2, or makes the phase of the gate signal Vg1 or the voltage Vds2 the same as the phase of the AC voltage Vin. If the generated reactance ΔX is inductive, the control unit 13 delays the phase of the gate signal Vg1 or the voltage Vds2 by a predetermined value relative to the phase of the AC voltage Vin. If the generated reactance ΔX is capacitive, the control unit 13 advances the phase of the gate signal Vg1 or the voltage Vds2 by a predetermined value relative to the phase of the AC voltage Vin.

[0035] That is, the control unit 13 controls the phase of the gate signal Vg1 or the phase of the voltage Vds2 so as to cancel the effect of the generated reactance ΔX. This allows AC power to be converted to DC power while automatically matching impedance. Furthermore, since the switching frequency is the same as the AC power supply frequency, application to a contactless power supply system is easy and switching loss can be reduced. Furthermore, compared to the configuration disclosed in Non-Patent Document 1, the circuit size can be made smaller. Furthermore, by setting the predetermined value for delaying or advancing the phase to 90°, the output voltage Vo of the rectifier circuit 11 and the operation of the switching elements S1 and S2 function as an auxiliary power supply that is 90° out of phase with the voltage Vin, thereby enabling efficient impedance matching.

[0036] (Second embodiment) Hereinafter, the same parts as in the first embodiment will be assigned the same reference numerals and their explanation will be omitted, and only the differences will be explained. In the second embodiment, the dead time added to reduce switching loss is shown depending on whether the generated reactance ΔX is inductive or capacitive. The drain current flowing through the switching element S1 is Is1, the drain-source voltage is Vds1, and the current flowing through the diode D1 is Id1.

[0037] As shown in FIG. 19, in the case of an inductive reactance where the generated reactance ΔX>0, if the dead time provided between the gate signals Vgs1 and Vgs2 is shorter than the fall time of the voltage Vds1, when the gate signal Vgs1 rises and the switching element S1 turns on, the current Is1 starts to flow at a timing when the voltage Vds1 does not reach 0 V, resulting in switching loss.

[0038] In contrast, as shown in Figure 20, if a dead time equal to or longer than the fall time of voltage Vds1 is added, current Id1 flows through diode D1 during that time, voltage Vds1 becomes approximately 0 V, and current Is1 becomes 0 A. When gate signal Vgs1 rises from that point, current Is1 begins to rise from 0 A. As a result, current Id1 begins to flow while voltage Vds1 is 0 V, so that turning on of switching element S1 becomes so-called zero voltage switching, and switching loss is reduced.

[0039] Furthermore, as shown in FIG. 21, if the dead time provided in the capacitive case where the generated reactance ΔX<0 is shorter than the fall time of the drain current Is1, the current Is1 will start to flow at a timing when the voltage Vds1 does not reach 0 V, resulting in switching loss.

[0040] 22, if a dead time equal to or longer than the fall time of current Is1 is provided, current Id1 flows through diode D1 during that time, causing voltage Vds1 to become approximately 0 V, and current Is1 to become 0 A. From this point, voltage Vds1 rises, resulting in so-called zero-current switching at turn-off, and switching loss is reduced.

[0041] (Third embodiment) The third embodiment shows a variation of the configuration of a current-voltage phase difference detection circuit. In a current-voltage phase difference detection circuit 14A shown in Fig. 23, a series circuit of capacitors 31 and 32 is connected between AC terminals. A circuit combining a transformer 33, a capacitor 34, and diodes 35a and 35b is connected in parallel to the capacitor 31. The primary winding of the transformer 33 is connected to the AC terminals, and the capacitor 34 is connected in parallel to the secondary winding.

[0042] The cathode of diode 35a and the anode of diode 35b are connected to the common connection point of capacitors 31 and 32. The anode of diode 35a and the cathode of diode 35b are connected to one end and the other end of capacitor 34, respectively. A voltage VDC is output from the common connection point of capacitors 31 and 32 via a resistor 36. A parallel circuit of a capacitor 37 and a resistor 38 is connected between the output terminal of voltage VDC and ground.

[0043] If the voltage at the common connection point of capacitors 31 and 32 is Vvd and the terminal voltage of capacitor 34 is Vid, the output voltage VDC of the current-voltage phase difference detection circuit 14A is a value corresponding to the phase difference of voltage Vid relative to voltage Vvd, that is, the current-voltage phase difference, as shown in Fig. 24. Depending on the lag or lead of the phase difference, voltage VDC changes from a negative value to a positive value.

[0044] 25, a primary winding of a transformer 39 is connected instead of the capacitor 32. The secondary winding of the transformer 39 is connected to terminals a and b of a diode bridge circuit 40. The midpoint of the secondary winding is connected to ground. On the other hand, the midpoint of the secondary winding of the transformer 33 is not connected to ground but serves as an output terminal for the voltage VDC. The secondary winding is also connected to terminals c and d of the diode bridge circuit 40.

[0045] If the terminal voltage of the secondary winding of transformer 39 is Vvd and the terminal voltage of the secondary winding of transformer 33 is Vid, the output voltage VDC of current-voltage phase difference detection circuit 14B will have a value that corresponds to the phase difference between voltage Vid and voltage Vvd, as shown in Fig. 26. Depending on whether the phase difference is a lag or lead, voltage VDC changes by a positive value, and when the phase difference is zero, voltage VDC is at its maximum value.

[0046] (Fourth embodiment) The fourth embodiment shows a variation in obtaining the reference signal to be input to the control unit 13. In the rectifier circuit 11A shown in Fig. 27, when the AC power supply 1A is configured as a circuit, if a clock signal is used to generate the frequency of the AC power supply therein, the clock signal is used as the reference signal.

[0047] (Fifth embodiment) 28, a rectifier circuit 11B of the fifth embodiment uses a current-voltage detection circuit 41 instead of the current-voltage phase difference detection circuit 14. The current-voltage detection circuit 41 detects the AC voltage Vin and the AC current Iin and outputs them to a control unit 13A. The control unit 13A estimates the generated reactance ΔX based on the input AC voltage Vin and AC current Iin, and controls the switching elements S1 and S2 in accordance with the estimation result, as in the first embodiment.

[0048] (Sixth embodiment) A rectifier circuit 11C of the sixth embodiment shown in FIG. 29 includes a current-voltage detection circuit 42 connected in parallel to the load 7, instead of the current-voltage detection circuit 41 of the fifth embodiment. The current-voltage detection circuit 42, which corresponds to a current-voltage detection unit, detects the terminal voltage of the load 7 and the current flowing through the load 7, i.e., the output voltage and output current, and calculates the power Po consumed by the load 7. Then, as shown in FIG. 30, a control unit 13B sweeps the pulse phase difference of the gate signal Vg1 so that the power Po is equal to or greater than a predetermined value. The current-voltage detection circuit 42 corresponds to a power detection unit.

[0049] Next, the operation of the sixth embodiment will be described. As shown in Fig. 31, when the control unit 13B first detects that the current power Po is less than a predetermined value P (S1), it sets the phase of the pulse of the gate signal Vg1 of the switching element S1 to the same phase as the phase of the AC voltage Vin (S2). Then, it detects the output voltage and output current via the current-voltage detection circuit 42 and calculates the power Po (S3).

[0050] Next, it is determined whether the power Po is equal to or greater than a predetermined value P (S4). If it is equal to or greater than the predetermined value P (YES), the phase of the pulse of the gate signal Vg1 is fixed to the same phase as the phase of the AC voltage Vin (S11). If the power Po is less than the predetermined value P (NO), the phase of the pulse of the gate signal Vg1 is set to have a phase difference of -Φ or +Φ with respect to the phase of the AC voltage Vin (S5). Then, the power Po is calculated (S6) in the same manner as in step S3.

[0051] Next, the same determination as in step S4 is made (S7). If the determination is "YES," the phase of the pulse of gate signal Vg1 is fixed to a state in which it has a phase difference of -Φ or +Φ with respect to the phase of AC voltage Vin (S12). On the other hand, if the determination is "NO," the phase of the pulse of gate signal Vg1 is set to have a phase difference of +Φ or -Φ with respect to the phase of AC voltage Vin (S8). That is, here, the phase difference is set with a polarity opposite to the polarity selected in step S5.

[0052] Thereafter, the same processing and judgment as in steps S3 and S4 are performed (S9, S10), and if the judgment is "YES", the phase of the pulse of the gate signal Vg1 is fixed to a state where the phase difference with respect to the phase of the AC voltage Vin is +Φ or -Φ (S13). On the other hand, if the judgment is "NO", it is estimated that the decrease in power Po is due to a factor other than the generated reactance ΔX, and it is determined that there is a possibility of an abnormality occurring (S14).

[0053] As described above, according to the sixth embodiment, the current / voltage detection circuit 42 that detects the power Po consumed by the load 7 between the DC output terminals of the rectifier circuit 11C is provided, and the control unit 13B changes the phase of the pulse of the gate signal Vg1 relative to the phase of the AC voltage Vin in accordance with the power Po, and fixes the phase difference of the gate signal Vg1 relative to the phase of the AC voltage Vin when the power Po becomes equal to or greater than the target value P. Even with this configuration, the same effects as those of the first embodiment etc. can be obtained.

[0054] (Other embodiments) The second embodiment may be applied to the third to sixth embodiments. Moreover, the fourth embodiment may be applied to the fifth and sixth embodiments. The switching element is not limited to an N-channel MOSFET, but a P-channel MOSFET may be used as the first switching element, or an IGBT or the like may be used. The present invention is not limited to applications in contactless power supply systems, but can be widely applied to applications for converting AC power to DC power.

[0055] Although the present disclosure has been described with reference to the embodiments, it is understood that the present disclosure is not limited to the embodiments or structures. The present disclosure also encompasses various modifications and equivalent modifications. In addition, various combinations and forms, including only one element, more than one element, or less than one element, are also within the scope and spirit of the present disclosure. [Explanation of symbols]

[0056] In the drawing, 1 is an AC power supply, 7 is a load, 11 is a rectifier circuit, 12 is a resonance circuit, 13 is a control unit, 14 is a current-voltage phase difference detection unit, Cm is a smoothing capacitor, S1 and S2 are switching elements, and D1 and D2 are diodes.

Claims

1. a series circuit of first and second switching elements (S1, S2) connected between the DC output terminals; In each of the first and second switching elements, a diode (D1, D2) is connected at its anode to a low-voltage side terminal and at its cathode to a high-voltage side terminal; a series resonant filter (12) connected between one end of the AC input terminal and a common connection point of the first and second switching elements, the resonant frequency of which is set equal to the AC power supply frequency; a control unit (13, 13A, 13B) that switches the first and second switching elements at the same frequency as the AC power supply frequency, with a duty ratio of 50%, and with a phase difference of 180 degrees from each other; the control unit controls a phase of a drive signal for switching the first switching element or a phase of a voltage between terminals of the second switching element in accordance with a reactance that occurs subsequently on a power transmission path so that a predetermined phase difference is achieved with respect to a phase of an AC voltage to be input; If the generated reactance does not exist, the first and second switching elements are turned off, or the phase of a drive signal for switching the first switching element or the phase of a voltage between terminals of the second switching element is made the same as the phase of the AC voltage; If the generated reactance is inductive, the phase of the drive signal for the first switching element or the phase of the voltage between the terminals of the second switching element is delayed by a predetermined value with respect to the phase of the AC voltage, If the generated reactance is capacitive, a rectifier circuit advances the phase of the drive signal of the first switching element or the phase of the voltage between the terminals of the second switching element by a predetermined value relative to the phase of the AC voltage.

2. a smoothing capacitor (Cm) connected in parallel to the series circuit; 2. The rectifier circuit according to claim 1, wherein the predetermined value is 90 degrees.

3. 3. The rectifier circuit according to claim 1, wherein, if the generated reactance is inductive, the control unit sets a dead time, in the drive signal that switches the first and second switching elements, that is equal to or longer than a transition time of the voltage between the output terminals of the switching elements.

4. 3. The rectifier circuit according to claim 1, wherein, if the generated reactance is capacitive, the control unit sets a dead time in the drive signal that switches the first and second switching elements that is equal to or longer than a transition time of the current flowing through the switching elements.

5. a phase difference detection unit (14) for detecting a current-voltage phase difference of input AC power; 3. The rectifier circuit according to claim 1, wherein the control unit determines the state of the generated reactance in accordance with the current-voltage phase difference.

6. 6. The rectifier circuit according to claim 5, wherein the phase difference detector is disposed on the output side of the series resonant filter.

7. a current / voltage detection unit (41) for detecting the current and voltage of AC power input to the output side of the series resonant filter; 3. The rectifier circuit according to claim 1, wherein the control unit (13A) determines the state of the generated reactance according to the current and the voltage.

8. a power detection unit (42) for detecting power input between the DC output terminals, 3. The rectifier circuit according to claim 1, wherein the control unit (13B) changes the phase of the drive signal of the first switching element relative to the phase of the AC voltage in accordance with the power, and fixes the phase difference of the drive signal relative to the phase of the AC voltage when the power becomes equal to or greater than a target value.

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