Differential voltage and current detector

Abstract

1. An apparatus for detecting and / or measuring a pulsed voltage source, the apparatus comprising: a transformer having a primary winding electromagnetically coupled with a secondary winding, the primary winding having a first inductance; and a differentiator including at least one resistive element coupled with the primary winding, thereby forming a closed circuit with the pulsed voltage source, the differentiator configured to electromagnetically induce an induced pulsed voltage on the secondary winding, the induced pulsed voltage being indicative of the pulsed voltage source.

Description

[Technical Field]

[0001] The disclosed technology relates generally to voltage and current sensing, and more particularly to an apparatus for detecting and measuring differential voltage and current in power electronics systems. [Background technology]

[0002] In general, various methods and devices for sensing voltage and current are known in the art. There are various measurement and detection technologies, some of which include those based on Faraday's law of electromagnetic induction and the Rogowski coil method, such as sensing (shunt) resistors and current transformers (CTs), those based on magnetic field sensors (e.g., Hall-effect sensors and fluxgate sensors), those based on the magnetoresistive effect using magnetoresistive current sensors, those based on Faraday's magneto-optical effect in optical fibers positioned around current-carrying conductors, dedicated integrated circuits (ICs), conductor trace resistance sensing in printed circuit boards (PCBs), and direct methods using conventional ammeters and voltmeters. Each of these methods has advantages and disadvantages. For example, the shunt resistor technique is relatively simple but exhibits increasing power loss with increasing current flow through the shunt resistor. CT techniques can be used to measure high currents, but generally may exhibit hysteresis and typically exhibit an undesirable direct current (DC) component that can cause saturation of the CT's magnetic core material (e.g., ferrite). Rogowski coil technology exhibits low inductance and does not saturate due to the lack of a magnetic core, but this method exhibits relatively low sensitivity and typically requires the use of amplifiers and integrating circuits, which require power. Technologies based on magnetic field sensors, such as Hall-effect sensors, can achieve high-precision measurements but typically produce low-level outputs that require signal-conditioning electronics, including low-noise amplifiers and temperature compensation circuits. Technologies based on the magnetoresistive effect generally enable highly sensitive measurements but may exhibit nonlinear behavior and be susceptible to damage from external magnetic fields. Trace resistance sensing technologies are low-cost but typically suffer from thermal drift of the conductor traces, which effectively alters current measurements and requires the use of amplifiers to obtain useful measurements.

[0003] Referring now to FIG. 1, FIG. 1 is a schematic diagram of a series connection detection circuit (shown generally at 10) for measuring current flow through a transistor implementation switch, which is prior art. Circuit 10 includes a transistor 12, a sense resistor 14, and a voltage sensing sub-circuit 16 (implemented via a low-pass filter / integrator circuit). The voltage sensing sub-circuit 16 includes a resistor 18 and a capacitor 20 that form a series RC circuit. The transistor 12 functions as a power switch and is typically implemented by a metal oxide semiconductor field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), or the like. The transistor 12 typically includes terminals denoted as gate (G), drain (D), source (S), and body (B), with the drain terminal connected to a high voltage power supply 22, the gate terminal connected to an input (gate drive) signal source (not shown), the body terminal and the source terminal interconnected with each other and further connected to the sense resistor 14 and the low-pass filter / integrator 16. The high voltage is connected between the drain terminal of the transistor 12 and ground.

[0004] FIG. 1 shows an input signal 24 (as a function of time) that drives the gate terminal of the transistor 12. The input signal 24 is used to control the switching state of the transistor 12, and the transistor 12 functions as a switch between a conducting ('on') state and a non-conducting ('off') state. For simplicity, the transient state is not described in this prior art example. The input signal 24 at t < t1 is 0, and the transistor 12 is switched off (i.e., not made conductive between the high voltage source 22 and the sense resistor 14). The input signal 24 between t1 ≦ t ≦ t2 drives the transistor 12 to switch to the on state, resulting in current flowing through the transistor 12 (i.e., between the drain terminal and the source terminal), as well as through the sense resistor 14. The current flowing through the sense resistor 14 (typically a low ohm resistor) causes a voltage drop V Rs across it. The voltage sensing sub-circuit 16, which is a series RC low-pass filter / integrator circuit, senses the voltage drop V Rsand generates an output signal 26. By measuring the output signal 26 and knowing the resistance of the sense resistor 14, the current flowing through the switch (i.e., transistor 12) and the sense resistor 14 is determined.

[0005] Some drawbacks of current sensing techniques using series-connected sense resistors such as that shown in FIG. 1 include time-accumulated energy loss (e.g., as heat) resulting from current flow through sense resistor 14 (i.e., depending on the current and resistor value), the creation of undesirable inductance between transistor 12 and sense resistor 14 which can cause inductive spikes, a typically low voltage output signal (on the order of millivolts), minimal or no protection against high voltages, and a voltage drop V Rs V of transistor 12 caused by gs This may include conduction losses indicated by a reduction in the gate-source voltage.

[0006] Another known prior art technique for sensing current uses a current sensing circuit or dedicated integrated circuit (IC) connected in parallel with a power switch (e.g., a switching transistor) through which the intended current flows. Reference is now made to FIGS. 2A and 2B. FIG. 2A is a schematic diagram of a prior art parallel-connected sensing circuit (generally designated 30) for measuring current flow through a transistor-implemented switch. FIG. 2B is a timing diagram (generally designated 50) corresponding to the operation of the integrated circuit (IC) in the parallel-connected sensing circuit of FIG. 2A. Circuit 30 (FIG. 2A) includes a transistor 32, an integrated circuit (IC) 34, and a gate resistor 36. While transistor 32 is shown as a MOSFET in FIG. 2A, the same principles apply to an IGBT. IC 34 is an IR25750 current sensing IC manufactured by International Rectifier Corporation (acquired by Infineon Technologies AG). IC 34 includes a CS pin ("pin," "connector," and "terminal" are synonymous herein), a GATE pin, a VS pin, and a COM pin. The GATE terminal of IC 34 is connected to the gate terminal of transistor 32 and to one end of gate resistor 36. The other end of gate resistor 36 is connected to gate drive input signal terminal 40. The COM terminal is connected to the source terminal of transistor 32, which is grounded. The VS terminal is connected to a high voltage switching node / terminal 42. The CS terminal constitutes an output signal terminal 44.

[0007] Transistor 32 functions as a power switch used to switch external devices and voltages (not shown) "on" and "off." The configuration described in connection with FIG. 2A shows IC 34 connected in parallel with the drain and source terminals of transistor 32. IC 34 can therefore operate as a parallel-connected sense circuit used to measure the voltage between the drain and source terminals of transistor 32, i.e., VDS(on) (if transistor 32 is a MOSFET), or the voltage between the collector and emitter terminals, i.e., VCE(on) (if transistor 32 is an IGBT). IC 34's internal circuitry (not shown but available online) includes a MOSFET (HVFET), an RC delay circuit, and a p-type metal-oxide semiconductor (PMOS) hold-down transistor. Details and principles of operation of the IR25750, including an internal circuit diagram (not shown), are described in Application Note AN-1199, published by International Rectifier Corporation, available at www.ifr.com. IC 34 utilizes a gate drive input signal 40 to power and turn its internal circuitry on and off.

[0008] 2B, during the "off" time of transistor 32, gate drive input signal 40 is "low" (i.e., at COM) and similarly at the CS terminal. When gate drive input signal 40 goes "high" (a square waveform), transistor 32 turns on and the drain voltage increases in proportion to the current flowing through transistor 32 and its characteristic R DS V depends on the product of (on) and (temperature dependent) DSThe voltage at the drain of transistor 32 decreases from a high voltage toward COM (on). Following a short time delay caused by the RC delay circuit of IC34, the HVFET turns on, the PMOS turns off, and the drain voltage of transistor 32 is conducted through the HVFET to the CS output terminal of IC34 as output signal 44. Gate resistor 36 enables IC34 to turn on with a short time delay after the turn-on time of transistor 32. Output signal 44 at the CS output terminal is a target current sense signal that can be supplied to auxiliary circuitry (not shown). Conversely, when gate drive input signal 40 transitions to "low," transistor 32 turns off, the HVFET turns off, and the CS terminal is at COM.

[0009] Although the parallel connected sense circuit 30 (FIG. 2A) is considered an improvement over the conventional series connected resistor and sense circuit 10 (FIG. 1), both circuits can exhibit noise spikes as a result of switching. DS (on) is the current flowing through transistor 32 and R DS (on) product, a "high" current (e.g., 2 amps) or a relatively "high" R DS At least one of the two (e.g., 0.25 Ω) is approximately

number

[0010] Another prior art technique for sensing current uses a current transformer connected to a sensing circuit that includes a sense resistor through which the current intended to be sensed flows. Reference is now made to Figures 3A, 3B, 3C, and 3D. Figure 3A is a schematic diagram of a prior art current sensing circuit (generally designated 60) that uses a current transformer to measure current flow. Figure 3B is a schematic diagram of a simple equivalent circuit (generally designated 80) of a portion of the current sensing circuit of Figure 3A. Figure 3C is a schematic diagram of an intermediate output voltage waveform (generally designated 90) corresponding to the operation of the current sensing circuit of Figure 3A. Figure 3D is a schematic diagram of an output voltage waveform (generally designated 96) corresponding to the operation of the current sensing circuit of Figure 3A. The current sensing circuit 60 (Figure 3A) includes a current transformer 62, a diode 72, a reset resistor 70, and a sense resistor 74 (R S ) Current transformer 62 includes a primary winding 64 (e.g., having one winding) electromagnetically coupled to a secondary winding 66 (e.g., typically having multiple windings). The inductance of the secondary winding is represented by inductor 68 (while its parasitic capacitance (i.e., between its windings) is not shown in FIG. 3A). The anode of diode 72 is connected to one terminal of secondary winding 66, and the cathode of diode 72 is connected to sense resistor 74, which forms a closed circuit with a second terminal of secondary winding 66.

[0011] Time-varying input current 76(I in ) (e.g., a pulse) flows through the primary winding 64, thereby generating a voltage that electromagnetically induces an induced voltage, i.e., voltage V2, in the secondary winding 66, as shown in FIG. 3C and represented by intermediate output voltage waveform 90. Reset resistor 70 is used to allow time for the intermediate output voltage V2 (after each input pulse) to reset so that the current transformer 62 does not saturate. FIG. 3B shows an equivalent circuit 80 of a portion of the current sense circuit 60 of FIG. 3A. The equivalent circuit 80 shows the secondary winding 66 (and primary winding 64) coupled to resistor R T and the equivalent capacitor C T 3B further illustrates a particular non-ideal aspect of the non-ideal current transformer 62, showing the parasitic capacitance (i.e., between its windings) represented by inductor LTs The inductance 68 of the secondary winding 66 is represented by the inductance of the capacitor C D The capacitance of the diode 72 is expressed as D The resistance of the diode 72 (as well as the reset resistor Rr and the sense resistor R S ) indicates the input current I in The "on" time between t1 and t2 (FIG. 3C), which corresponds to the on time of (FIG. 3A), is the current flow I through the reset resistor 70. m , and the positive component 92 of voltage V2 of intermediate output voltage waveform 90 (FIG. 3C). in The "off" time of the component L in the equivalent circuit 80 Ts , R T , C T and R r I in the subcircuit formed by m The output voltage V is characterized by a negative component 94 of the voltage V2 of the intermediate output voltage waveform 90. The diode 72 rectifies the negative component 94 of the intermediate output voltage waveform 90. Out (FIGS. 3A and 3D) show the relationship between voltage V2 and the forward voltage drop (V) across diode 72, as represented by output voltage waveform 96. D (i.e., V Out =V2+V D The resistance value of the reset resistor 70 is typically R T is chosen to be not much larger than the value of R r >>R T In the case of m Most of R will effectively flow through the secondary winding 66. Furthermore, the value of the sense resistor 74 is typically selected to be not much larger than the value of the reset resistor 70, otherwise R S >>R r In this case, most of the current flow will occur through reset resistor 70. Therefore, the low resistance of sense resistor 70 multiplied by the low current flowing through sense resistor 70 will result in a low voltage drop across sense resistor 70, which in turn will typically result in a low output voltage V Out amplification is required.

[0012] Summary of the Invention [Problem to be solved by the invention]

[0013] An object of the disclosed technique is to provide a novel apparatus for detecting and / or measuring a pulsed voltage source. The apparatus includes a transformer and a differentiator. The transformer has a primary winding electromagnetically coupled to a secondary winding. The primary winding has a first inductance. The differentiator includes at least one resistive element (e.g., a resistor) coupled to the primary winding, thereby forming a closed circuit with the pulsed voltage source. The differentiator is configured to electromagnetically induce an induced pulsed voltage on the secondary winding such that the induced pulsed voltage is indicative of the pulsed voltage source. [Means for solving the problem]

[0014] Therefore, according to another aspect of the disclosed technique, there is provided an apparatus for detecting and / or measuring a pulsed voltage source. The apparatus includes a differentiator and a transformer. The differentiator includes at least one resistive element (e.g., a resistor) coupled to at least one capacitive element (e.g., a capacitor), thereby forming a closed circuit with the pulsed voltage source. The transformer has a primary winding electromagnetically coupled to a secondary winding. The primary winding is connected in parallel to the at least one resistive element. The primary winding is configured to electromagnetically induce an induced pulsed voltage on the secondary winding such that the induced pulsed voltage is indicative of the pulsed voltage source.

[0015] Therefore, according to a further aspect of the disclosed technique, there is provided an apparatus for detecting and / or measuring a pulsed voltage source. The apparatus includes a transformer having a primary winding electromagnetically coupled to a secondary winding. The primary winding has a first inductance. A resonant circuit includes at least one capacitive element (e.g., a capacitor) that forms an LC circuit with the primary winding. The resonant circuit forms a closed circuit with the pulsed voltage source. The primary winding is configured to electromagnetically induce an induced pulsed voltage on the secondary winding such that the induced pulsed voltage is indicative of the pulsed voltage source.

[0016] Therefore, according to another aspect of the disclosed technique, there is provided a method for detecting and / or measuring a pulsed voltage source. The method includes differentiating a pulse signal of the pulsed voltage source via a differentiator, upconverting the differentiated signal, and detecting and / or measuring the upconverted differentiated signal. The differentiator has an inductive component that is a winding of a transformer such that differentiation generates a differentiated signal. Upconversion of the differentiated signal via the transformer generates an upconverted differentiated signal. The upconverted differentiated signal is indicative of the pulsed voltage source.

[0017] The techniques of the present disclosure will be more fully understood and appreciated from the following detailed description taken in conjunction with the drawings, in which: [Brief explanation of the drawings]

[0018] [Figure 1] FIG. 1 is a schematic diagram of a series-connected sensing circuit for measuring current flow through a prior art transistor-implemented switch. [Figure 2A] FIG. 1 is a schematic diagram of a parallel-connected sensing circuit for measuring current flow through a prior art transistor-implemented switch. [Figure 2B] 2B is a timing diagram corresponding to the operation of the integrated circuits (ICs) in the parallel-connected sensing circuit of FIG. 2A. [Figure 3A] FIG. 1 is a schematic diagram of a prior art current sensing circuit that uses a current transformer to measure current flow. [Figure 3B] FIG. 3B is a schematic diagram of a simple equivalent circuit of a portion of the current sensing circuit of FIG. 3A. [Figure 3C] 3B is a schematic diagram of an intermediate output voltage waveform corresponding to the operation of the current sensing circuit of FIG. 3A. [Figure 3D] 3B is a schematic diagram of an output voltage waveform corresponding to the operation of the current sensing circuit of FIG. 3A. [Figure 4A] FIG. 1 is a schematic diagram of an apparatus for differential voltage current sensing of a pulsed voltage source, constructed and operative in accordance with one embodiment of the disclosed technique; [Figure 4B] FIG. 4B is a schematic diagram of a basic equivalent circuit of the device of FIG. 4A in accordance with the techniques of the present disclosure. [Figure 4C] FIG. 1 is a schematic diagram of a characteristic plot of output voltage as a function of time plotted together with a plot of an input time-dependent pulse signal of a voltage source in accordance with the techniques of the present disclosure. [Figure 5] 1 is a schematic diagram of one implementation of an apparatus 100 for differential voltage current sensing in a transistor-implemented switch used to switch voltages and currents, in accordance with the techniques of the present disclosure. [Figure 6A] FIG. 1 is a schematic diagram of an apparatus for differential voltage current sensing of a pulsed voltage source, constructed and operative in accordance with another embodiment of the disclosed technique; [Figure 6B] FIG. 6B is a schematic diagram of a simulation circuit for the apparatus shown in FIG. 6A, constructed and operative in accordance with an embodiment of the disclosed technique. [Figure 6C] 6C is a plot of the output voltage produced by the simulation circuit of FIG. 6B. [Figure 6D] 6C is a plot of the output voltage of the simulation circuit of FIG. 6B as a function of the rise time of the input voltage of the pulsed voltage source. [Figure 6E] 6C is a plot of the output voltage of the simulation circuit of FIG. 6B as a function of the capacitor value of the device of FIG. 6A. [Figure 7] FIG. 10 is a schematic diagram of an apparatus for differential voltage current sensing of a pulsed voltage source, constructed and operative in accordance with a further embodiment of the disclosed technique; [Figure 8A]FIG. 1 is a schematic diagram of an apparatus utilizing a resonant circuit for differential voltage current sensing of a pulsed voltage source, constructed and operative in accordance with another embodiment of the disclosed technique; [Figure 8B] FIG. 8B is a schematic diagram of a basic model of an equivalent circuit of the resonant circuit of the device of FIG. 8A. [Figure 8C] 1 is a schematic diagram of a characteristic plot of an input time-dependent pulse signal of a voltage source; [Figure 8D] 8B is a characteristic plot of the time-dependent voltage drop in the primary winding of the transformer according to a first model configuration of the resonant circuit of the device of FIG. 8A. [Figure 8E] 8B is a characteristic plot of the time-dependent voltage drop in the primary winding of the transformer according to a second model configuration of the resonant circuit of the device of FIG. 8A. [Figure 8F] 8B is a characteristic plot of output voltage as a function of time for the device of FIG. 8A in accordance with an embodiment of the disclosed technique. [Figure 9A] FIG. 10 is a schematic diagram of one implementation of an auxiliary output detection measurement comparison circuit configured for use with the apparatus of the disclosed technology. [Figure 9B] FIG. 10 is a schematic diagram of another implementation of an auxiliary output peak / envelope detection circuit configured for use with the apparatus of the disclosed technology. [Figure 9C] FIG. 10 is a schematic diagram of a further implementation of an auxiliary output comparator peak detector configured for use with the apparatus of the disclosed technology. [Figure 10] FIG. 1 is a schematic diagram of a method for detecting and / or measuring a pulsed voltage source in accordance with the principles of the disclosed technique. DETAILED DESCRIPTION OF THE INVENTION

[0019] The disclosed technology overcomes the shortcomings of the prior art by providing an apparatus for detecting and / or measuring (1) the voltage of a voltage source and / or (2) the current flowing through a voltage source through at least one differential method through a "novel" use of a transformer. A transformer is generally a device configured and operable to transfer electrical energy between at least two circuits via electromagnetic induction. Some ubiquitous conventional uses of transformers include stepping down and stepping up voltage and current, electrical isolation between circuits, impedance matching, signal filtering, alternating current (AC) phase angle adjustment, etc. The disclosed technology simultaneously uses the transformer in a conventional manner (i.e., to transfer electrical energy between circuits) and, more importantly, in a novel manner. Considering a transformer including a primary winding and a secondary winding electromagnetically coupled to each other, the transformer is configured to transfer electrical energy between the primary winding and the secondary winding (i.e., depending on the characteristics of an applied energy source to at least one of the windings). A novel use of a transformer according to one embodiment of the disclosed technology involves utilizing the primary winding as an inductive element in a differentiation circuit. Thus, the primary winding is a key component shared by the dual uses of both the transformer and the differentiation circuit. According to another embodiment of the disclosed technology, the primary winding is an inductive element in a resonant circuit.

[0020] Specifically, according to one embodiment of the disclosed technology, an apparatus for detecting and / or measuring a pulsed voltage source (i.e., synonymously referred to herein as a "pulsed voltage source") that is characterized by not having a constant voltage over time is provided. The apparatus includes a transformer and a differentiator. The transformer includes a primary winding (having a first inductance) electromagnetically coupled to a secondary winding. The differentiator includes at least one resistive element (e.g., a resistor, a plurality of resistors having effective electrical resistances) coupled to the primary winding, thereby forming a closed circuit with the pulsed voltage source. Therefore, the main inductive component of the differentiator is common to a part of the transformer, i.e., its primary winding. The differentiator is configured to electromagnetically induce an induced pulsed voltage (i.e., a pulsed voltage electromagnetically induced according to the principle of electromagnetic induction) on the secondary winding, where the induced pulsed voltage represents a pulsed voltage source.

[0021] The techniques of this disclosure generally address the need for voltage and current sensing in power electronics systems for purposes of measurement (e.g., data derivation), control (e.g., power regulation, feedback, etc.), monitoring (e.g., self-monitoring of power circuits), and protection and safety (e.g., to prevent overload conditions, short circuits, etc.). For example, there is a need to obtain information regarding current flow through a particular high-voltage switching device (e.g., a high-voltage transistor or a series of interconnected high-voltage switching transistors).

[0022] To further disclose this embodiment of the disclosed technology in more detail, reference is now made to FIGS. 4A, 4B, and 4C. FIG. 4A is a schematic diagram of an apparatus (generally designated 100) for differential voltage-current sensing of a pulsed voltage source, constructed and operative in accordance with one embodiment of the disclosed technology. FIG. 4B is a schematic diagram of a basic equivalent circuit (generally designated 130) of the apparatus of FIG. 4A in accordance with the disclosed technology. FIG. 4C is a diagram illustrating a characteristic plot of the output voltage as a function of time (generally designated 140) plotted together with a plot of the input time-dependent pulse signal of the voltage source in accordance with the disclosed technology. Apparatus 100 includes a transformer 104 and a differentiator 106. Transformer 104 includes a primary winding 108 and a secondary winding 110, the primary winding and secondary windings being electromagnetically coupled to one another. Primary winding 108 is a N P The secondary winding 110 includes N windings and has a first inductance L1 (which is non-zero). S It includes two windings and has a second inductance L2 (which may be 0). (N P and N S is a number.) Differentiator 106 includes at least one (electrical) resistive element represented by resistor 112 coupled to primary winding 108. Therefore, the main inductive component in differentiator 106, i.e., primary winding 108, is also part of transformer 104. Apparatus 100 further includes input terminals 1141, 1142 and output terminals 1161, 1162, as shown in FIG. 4A. Apparatus 100 receives a pulsed external time-varying voltage source 120 ("pulsed voltage source") (V S (t), and in this specification, R S4A ). The differentiator 106, specifically the series coupled pair of resistor 112 and primary winding 108, is configured to form a closed circuit with pulsed voltage source 120, as shown in FIG. 4A . The differentiator 106 is configured and operable to electromagnetically induce an induced pulsed voltage in secondary winding 110 such that the induced pulsed voltage is indicative of pulsed voltage source 120, as will be described in more detail below. Additionally, device 100, specifically secondary inductor 110, couples an external load 122 (R L ), i.e., a "sensing load" (e.g., a shunt resistor ("sensing resistor" or "resistive load")) through which current may flow and the current may be detected and measured. Apparatus 100, including peripheral elements such as voltage source 120 and external load 122, is shown collectively at 102 in FIG. 4A. It should be noted that at least one resistive element, effectively represented by resistor 112, may embody a single resistor, multiple resistors (series and / or parallel), an impedance (i.e., ohmic resistance and reactance components), or an equivalent device exhibiting electrical resistance consistent with the principles of the disclosed techniques.

[0023] FIG. 4B shows a basic equivalent circuit 130 of the device 100 (FIG. 4A), where a voltage source 120 applies a pulse signal 142 (FIG. 4C) to the series RL circuit of the differentiator 106 (FIGS. 4A, 4B), generating a current I PThe pulse signal 142 is configured to form a closed circuit (FIG. 4B) through which V flows. The pulse signal 142 is a time-varying signal having a waveform that includes rising and falling edges. While the pulse signal 142 is shown to be rectangular in FIG. 4C, other pulse shapes (e.g., having rising and falling edges) are also feasible. Furthermore, for purposes of illustrating the main principles of the disclosed techniques, it should be noted that the basic equivalent circuit 130 represents a simple model that does not include equivalent subcircuits of a "real" (i.e., non-ideal) transformer that exhibit leakage reactance, core losses (or magnetizing current losses such as hysteresis), flux leakage, parasitic capacitance, etc. The resistance of the resistor 112 is represented by R, and the non-zero inductance of the inductor 108 is represented by L1. The (time-dependent) voltage drop across the resistor 112 is represented by V R (t) (simply put, V R ) and the voltage drop across the inductor 108 is V L1 (t) (simply put, V L1 ) and the following equation holds:

number

number

number

[0024] The apparatus 100 is implemented for differential voltage / current sensing of a pulsed voltage source that is a switching device (or equivalent). For example, the switching device is configured and operable to rapidly turn on and off a connected voltage source such that the switching device itself effectively becomes a voltage source of pulsed voltage, referred to herein as a “pulsed voltage source.” The switching device may be embodied in the form of an electronic switch such as a switching circuit, a transistor (e.g., bipolar, metal-oxide semiconductor field-effect transistor (MOSFET), insulated-gate bipolar transistor (IGBT), etc.), a silicon-controlled rectifier (SCR), a triode alternating current (TRIAC) switch, a diode alternating current (DIACETYL) switch, an electromagnetic switch such as a relay, a mechanical switch such as a toggle switch, a pushbutton switch, and a physical property-dependent switch (i.e., switches of various physical properties such as pressure, temperature, magnetic field, light intensity, etc., in a pressure-dependent switch, a temperature-dependent switch, a magnetic field-dependent switch, a light intensity-dependent switch, etc.). Alternatively, the pulsed voltage of the pulsed voltage source is generated by a current flowing through a resistive element (e.g., a resistor, a resistive load, a sense resistor, etc.), resulting in a voltage drop across the resistive element. This resistive element is the internal resistance (R) of the "on" state (e.g., of a switching device such as a silicon (Si) MOSFET, a gallium nitride (GaN) high electron mobility transistor (HEMT), etc.). S (on)). According to a particular implementation, current flow through the sense resistor occurs by the sense resistor being connected in series with a switching device (i.e., a pulsed voltage source). For further details of this exemplary implementation, reference is now made to FIG. 5, which is a schematic diagram of one implementation of an apparatus 100 (generally designated 150) for differential voltage current sensing in a transistor-implemented switch used to switch voltages and currents in accordance with the techniques of this disclosure.

[0025] 5 illustrates an implementation 150 of device 100 (FIG. 4A) in which the input of device 100 is coupled to a transistor-implemented switch, which in this example (without loss of generality) is embodied as a MOSFET 152. MOSFET 152 includes a gate terminal (G), a body terminal (B), a drain terminal (D), and a source terminal (S). Specifically, input terminals 1141, 1142 (FIG. 4A) of device 100 are connected to the drain and source terminals, respectively, of MOSFET 152. The drain terminal of MOSFET 152 is connected to voltage source 154, and the source terminal is connected to ground. MOSFET 152 is coupled to an externally controlled gate-source voltage (V GS MOSFET 152 is configured and operable as a (voltage-controlled) switch between an “off” state (i.e., no conduction between the drain and source terminals) and an “on” state (i.e., conduction between the drain and source terminals) according to a MOSFET 152 signal (not shown). During the off state of MOSFET 152, there is no conduction between the drain and source terminals; in fact, the switch is in an open state, and no current flows between voltage source 154 and ground, and to device 100. In the transition of MOSFET 152 from the off state to the on state, current begins to flow between the drain and source terminals, and device 100 receives as input, via input terminals 1141, 1142 (FIG. 4A), a voltage pulse 156 (e.g., having a rising edge from a “high” voltage (absolute) value to a “low” voltage (absolute) value). When MOSFET 152 is in the on state, the on state is controlled by an internal resistance (herein R ) that depends on the type of transistor and other factors, such as temperature. DS A typical example of a value is R DS (on)≒1mΩ. The internal resistance R DS (on) is the voltage drop V of MOSFET 152 DS (on), and then the device 100 generates a voltage drop V DS(on). Similarly, in the transition of MOSFET 152 from an on state to an off state, current ceases to flow between the drain and source terminals, and voltage pulse 156 changes from "high" to "low." Apparatus 100 detects the sudden voltage drop across MOSFET 152 by the falling edge of voltage pulse 156. As discussed above in connection with FIGS. 4A, 4B, and 4C, voltage pulse 156 of the pulsed voltage source (FIG. 5) generates a corresponding voltage V at the terminals of inductor 108. d =V L1 , which in turn induces a corresponding induced pulse voltage in the secondary inductor 110, representing a voltage pulse 156. In accordance with the techniques of this disclosure, equation (3), where V S ≡V DS ) to R DS (on) and the value of R of resistor 112, we can further determine the output 158 ​​(V O ), the current I flowing through MOSFET 152 is determined according to the following equation:

number

[0026] In an alternative configuration (not shown), MOSFET 152 is replaced with a small value sense resistor (not shown) such that voltage source 154 is connected to the sense resistor and configured and operable as a pulsed voltage source. The sense resistor has an R value corresponding to the value of MOSFET 152. DS (on), or other values. The same principles of the techniques of this disclosure detailed above apply to this alternative configuration.

[0027] Thus, in accordance with another embodiment of the disclosed technique, there is provided a direct current (DC) filter and a high voltage (HV) protector configured to couple a resistor of a differentiator and a pulsed voltage source. The DC filter and the HV protector may be implemented, for example, by a capacitor. For further details of this embodiment, reference is now made to FIGS. 6A, 6B, and 6C. FIG. 6A is a schematic diagram of an apparatus for differential voltage-current sensing of a pulsed voltage source, constructed and operative in accordance with another embodiment of the disclosed technique. FIG. 6B is a schematic diagram of a simulation circuit of the apparatus shown in FIG. 6A, constructed and operative in accordance with an embodiment of the disclosed technique. FIG. 6C is a plot of the output voltage (generally designated 230) generated by the simulation circuit of FIG. 6B. Apparatus 200 (FIG. 6A) is essentially the same as apparatus 100 (FIG. 4A), with all components and their respective reference numerals remaining the same, except for the inclusion of capacitor 204 within apparatus 200. Apparatus 200 is implemented for differential voltage-current sensing of a pulsed voltage source (or equivalent, such as a switching device). Apparatus 200, including peripheral elements such as voltage source 120 and external load 122, is collectively designated 202 in FIG. 6A . One terminal of capacitor 204 is connected to resistor 112, and the other terminal is connected to pulsed voltage source 120. Capacitor 204 is a DC decoupler configured and operable to filter out (i.e., block) the DC component (i.e., at least one time-invariant DC component) of the pulsed voltage source signal (thus avoiding potential transformer saturation) and provide HV protection (to some extent) for apparatus 200. Because differentiator 106 effectively generates an output signal from the time-varying signal components of the pulsed voltage source signal, the time-invariant DC component of the pulsed voltage source signal can be filtered out by capacitor 204. If the pulsed voltage source signal includes a high-voltage DC component, the capacitor is configured and operable to provide HV protection for the remainder of apparatus 200 by filtering out such high voltages. The capacitance value of capacitor 204 is selected according to the characteristics of the pulsed voltage source signal. Furthermore, if the pulsed voltage source 120 generates repetitive periodic pulses, then X CThe capacitive reactance of the capacitor 204 is expressed as X C =-1 / (2πfC) and its reactance decreases (in absolute value) with increasing frequency. Thus, for a given capacitance C of capacitor 204, the higher the value of f, the greater the C The value of becomes lower.

[0028] 6B shows a simulation circuit (generally designated 220) for apparatus 200 (FIG. 6A), in which pulsed voltage source 120 is simulated by l1 and resistor R5, capacitor 204 (FIG. 6A) is simulated by capacitor C1 (FIG. 6B), transformer 104 (FIG. 6A) is simulated by transformer equivalent simulation subcircuit TRF1 (FIG. 6B), and load resistor 122 (FIG. 6A) is simulated by resistor R3. The transformer equivalent simulation subcircuit includes inductor L1 (simulating primary winding 108), inductor L2 (simulating secondary inductor 110), resistor R1 (simulating resistor 112), resistor R2, capacitor C2, and resistor R4, which simulates the resistance and parasitic capacitance of transformer 104. Furthermore, it should be noted that real (i.e., non-ideal) transformers exhibit a coupling coefficient (typically k<1). Typical values ​​used in simulation circuit 220 are shown in Figure 6B. Plot 230 in Figure 6C shows the output voltage ("V") generated by the simulation circuit of Figure 6B. OUT The peak-to-peak output voltage is approximately 1.6 volts, thus eliminating the need for an amplifier at the output.

[0029] Output voltage V OUT (Fig. 6B) generally indicates the input voltage V of the pulse voltage source l1. IN To further illustrate this dependency, reference is now made to FIG. 6D, which shows the input voltage V of the pulsed voltage source. IN The output voltage V of the simulation circuit of Figure 6B as a function of the rise time OUT1 is a plot (generally designated 250) of the pulse voltage l1. Plot 250 shows output voltage (in millivolts) on the vertical axis as a function of time (in nanoseconds) on the horizontal axis. The data points in plot 250 represent the input voltage V IN Output voltage V as a function of rise time OUT The plots demonstrate a fairly flat (i.e., insignificant) sensitivity or dependence on t. Generally, the output voltage decreases with increasing rise time of the (input) pulsed voltage source. In the example plot shown in Figure 6D, the value of capacitor C1 was 30 pF. The rise time range shown is from 0 (exclusive) to 300 ns (i.e., t r = (0, 300]), whereas the device 100 (FIG. 4A) and device 200 (FIG. 6A) of the disclosed technology typically support a very narrow range of rise times, e.g., t r = (0, 100 ns], in which case the output voltage has only a small dependence on the rise time.

[0030] Output voltage V OUT (FIG. 6B) also generally depends on the value of capacitor 204 (FIG. 6A) or C1 (FIG. 6B). To further illustrate this dependence, reference is made to FIG. 6E, which shows the output voltage V of the simulated circuit of FIG. 6B as a function of the capacitor value of the device of FIG. 6A. OUT 6E is a plot (generally designated 260) of . Plot 260 shows the output voltage (in volts) on the vertical axis as a function of the capacitance (in picofarads) of capacitor C1 (FIG. 6C) on the horizontal axis. The data points on plot 260 demonstrate that the output voltage generally decreases as the capacitance value of capacitor C1 increases. In the example plot shown in FIG. 6E, the rise time of pulsed voltage source l1 is t r =100ns.

[0031] In accordance with a further embodiment of the disclosed technique, there is provided a differentiator coupled to a primary winding of a transformer. The differentiator is separate from the transformer (i.e., the differentiator and the transformer have no shared components, i.e., are mutually exclusive in terms of components). To describe this embodiment in further detail, reference is now made to FIG. 7, which is a schematic diagram of an apparatus (generally designated 270) for differential voltage / current sensing of a pulsed voltage source, constructed and operative in accordance with a further embodiment of the disclosed technique.

[0032] Apparatus 270 is similar to apparatus 100 (FIG. 4A) in that identical components and their respective reference numerals are the same, except for differentiator 282, which is separate from transformer 104 (i.e., differentiator 282 and transformer 104 have no shared components). Specifically, apparatus 280 includes differentiator 282 and transformer 104, where differentiator 282 is of an RC type including at least one capacitive element represented (synonymously) by capacitor 284 (C3) and at least one resistive element represented (synonymously) by resistor 286 (R6). One terminal of capacitor 284 is connected to resistor 286, and the other terminal is connected to pulsed voltage source 120. Transformer 104 includes primary winding 108 and secondary winding 110. The primary winding 108 is electromagnetically coupled to the secondary winding 110 of the transformer 104 such that the primary winding 108 is configured to electromagnetically induce an induced pulse voltage on the secondary winding 110. The primary winding 108 is connected in parallel to a resistor 286. The differentiator 282 forms a closed circuit with the pulsed voltage source 120. The device 280 is implemented for differential voltage / current sensing of the pulsed voltage source 120 (or an equivalent, such as a switching device). The device 280, including the voltage source 120 and peripheral elements such as the external load 122, is collectively designated 270 in FIG. 7 .

[0033] The voltage source 120 is configured to apply a pulsed, time-varying (input) signal (not shown) to the differentiator 282, which is configured and operable to receive the pulsed, time-varying input signal and differentiate it to produce a differentiated signal (not shown). The transformer 104 is configured and operable to up-convert (i.e., step-up convert) the differentiated signal received by the primary winding 108, thereby converting V O The device 280 generates an upconverted differential signal (not shown) output at output terminals 1161, 1162, designated (t). The device 280 enables detection and / or measurement of the upconverted differential signal at output terminals 1161, 1162 by measuring an output voltage indicative of a pulsed voltage source. Alternatively, the device 280 may be coupled to a sensing device 122 (e.g., resistor R L The resistors 12 and 13 allow for the detection and / or measurement of current through the resistors 12 and 13 (resistive loads shown in FIG. 1).

[0034] Thus, according to another embodiment of the disclosed technique, there is provided a resonant circuit having a capacitor that forms an LC circuit with a primary winding of a transformer. The resonant circuit forms a closed circuit with a pulsed voltage source. The primary winding of the transformer induces an induced pulsed voltage on the secondary winding of the transformer such that the induced pulsed voltage is indicative of the pulsed voltage source. To further elaborate on this embodiment, reference is now made to FIGS. 8A, 8B, and 8C. FIG. 8A is a schematic diagram of an apparatus (generally designated 300) utilizing a resonant circuit for differential voltage-current sensing of a pulsed voltage source, constructed and operative in accordance with another embodiment of the disclosed technique. FIG. 8B is a schematic diagram of a basic example equivalent circuit (generally designated 330) of the resonant circuit of the apparatus of FIG. 8A. FIG. 8C is a schematic diagram of a characteristic plot (generally designated 350) of an input time-dependent pulse signal of the pulsed voltage source. Apparatus 300 (FIG. 8A) is essentially the same as the corresponding components in apparatus 100 (FIG. 4A), except for the omission of resistor 112 (from apparatus 300) and the inclusion of capacitor C4 in apparatus 300 (FIG. 8A). Apparatus 300 is implemented for differential voltage-current sensing of a pulsed voltage source (or equivalent, such as a switching device).

[0035] Specifically, device 300 includes a transformer 304 and a resonant circuit 306. Transformer 304 includes a primary winding 308 and a secondary winding 310, the primary and secondary windings being electromagnetically coupled to one another. Similar to device 100 (FIG. 4), primary winding 308 includes an N P The secondary winding 310 of the device 300 includes N windings and has a first inductance L3 (which is non-zero). S8A , the resonant circuit 306 includes two windings and a second inductance L4 (which may be zero). The resonant circuit 306 includes at least one (electrically) capacitive element, represented by a capacitor 305, coupled to the primary winding 308, thereby forming a parallel LC circuit. The main inductive component in the resonant circuit 306 is therefore common to the primary winding 308 of the transformer 304. The at least one capacitive element is typically embodied in the form of a capacitor. Alternatively, the at least one capacitive element is embodied as multiple capacitors coupled together (not shown) to form an effective capacitor. The apparatus 300 further includes input terminals 3141, 3142 and output terminals 3161, 3162, as shown in FIG. 8A . The apparatus 300 receives a pulsed external time-varying voltage source 320 ("pulsed voltage source") (V S (t) and the internal resistance during the "on" state (i.e., R S 8A ). The resonant circuit 306, specifically the parallel coupled pair of capacitor 305 and primary winding 308, is configured to form a closed circuit with pulsed voltage source 320, as shown in FIG. 8A . The primary winding 308 is configured and operable to electromagnetically induce an induced pulsed voltage on the secondary winding 310 such that the induced pulsed voltage is indicative of the pulsed voltage source 320, as will be described in more detail below. Furthermore, the device 300, specifically the secondary inductor 310, is configured and operable to couple to an external load 322 (R L ), i.e., a "sensing load" through which a current can flow and which current can be detected and measured. Apparatus 300, including peripheral elements such as voltage source 320 and external load 322, is shown collectively at 302 in FIG. 8A. Note that the at least one capacitive element is shown as capacitor 305, which may be embodied as a single capacitor, multiple capacitors (series and / or parallel), or an equivalent device exhibiting capacitance consistent with the principles of the disclosed techniques.

[0036] Pulsed voltage source 320 is configured to apply a pulsed signal 324 to resonant circuit 306 to form a closed circuit with pulsed voltage source 320. FIG. 8C shows a characteristic plot 350 of the input time-dependent pulsed signal of pulsed voltage source 320. Pulsed signal 324 is a time-varying signal having a waveform including rising and falling edges. Without loss of generality, a square pulse with a 50% duty cycle was selected for pulsed signal 324 to demonstrate the techniques of the present disclosure. The principles of the techniques of the present disclosure are equally applicable to other types of pulsed signals with different duty cycles (e.g., a rectangular pulse with a 20% duty cycle).

[0037] 8B shows a basic model of an equivalent circuit 330 of the resonant circuit 306 (FIG. 8A) of the apparatus 300. The equivalent circuit 330 includes an inductor 332 that models the non-zero inductance of the primary winding 308 (FIG. 8A), a capacitor 334 that models the capacitance of the capacitor 305 (FIG. 8A), a resistor 336 that models the electrical resistance of the primary winding 308, and a resistor 338 that models the series resistance of the resonant circuit 306 (FIG. 8A). The resistors 336, 338 represent real-world dissipative (electrical resistance) elements of the resonant circuit 306.

[0038] When the time-dependent pulse signal 324 of the pulsed voltage source 320 is applied, the resulting time-dependent voltage drop across the inductor 308 (total V d The output voltage (t), generally designated 360, of the transformer according to a first model configuration of the resonant circuit of the apparatus of FIG. 8A. FIG. 8E is a characteristic plot of the time-dependent voltage drop in the transformer primary winding according to a second model configuration of the resonant circuit of the apparatus of FIG. 8A. FIG. 8F is a characteristic plot of the output voltage as a function of time (designated generally 380) of the apparatus of FIG. 8A, in accordance with an embodiment of the disclosed technique.

[0039] Without loss of generality, two example configurations of a resonant circuit are described for the purposes of illustrating the principles of the disclosed techniques. Referring to Figure 8D, plot 360 shows the voltage drop (V) across the primary winding as a function of time in response to an applied time-dependent pulse signal 324 (Figure 8C) for a first model configuration (example) of resonant circuit 306 of apparatus 300 (Figure 8A) as modeled by equivalent circuit 330 (Figure 8B). d1 3 shows a characteristic waveform 362 of the resonant circuit 306 (denoted as (t)). According to a first model configuration of the resonant circuit 306, the value of the resistor 336 has a relatively small resistance (0.1 Ω) to produce a slight dissipative effect in the resonant circuit 306. Other exemplary values ​​of the components are: capacitor 334 = 10 μF, inductor 332 = 0.2 μH, and resistor 338 = 0 Ω. The frequency of the applied (periodic) pulse signal 324 is selected to match the resonant frequency f0 of the resonant circuit 306. The resonant frequency f0 is given by:

number

[0040] According to another generally preferred alternative, L3 and C3 are selected such that the period T0=1 / F0 is greater than the pulse width of the applied (periodic) pulse signal 324. The maximum voltage drop (V d1max ) is generally greater than the maximum voltage (V Smax ) and other factors such as the rise time and fall time of the pulse voltage signal 324.

[0041] According to the second example model configuration, resonant circuit 306 exhibits a greater dissipative effect, which causes resistor 336 to assume a value an order of magnitude larger (i.e., 1 Ω) compared to the first example model configuration of resonant circuit 306. FIG. 8E is a schematic diagram of a characteristic plot of a time-dependent waveform 372 of the voltage drop across the primary winding of the transformer (generally designated 360) according to a second example model configuration of the resonant circuit of the device of FIG. 8A. In this second example model configuration, aside from the change in the value of resistor 336, all of the other component values, including the resonant frequency, remain unchanged. The greater dissipative effect of resistor 336 generally corresponds to a change in V d1max The maximum voltage drop (V d2max This affects the

[0042] In response to a time-varying current flowing through the primary winding 308 (i.e., having a non-zero inductance, L3≠0), the primary winding 308 is configured and operable to generate a varying magnetic field that propagates through the electromagnetically coupled secondary inductor 310, which generates a time-varying output voltage V' between the output terminals 3161, 3162. O (t) generates an induced pulse output voltage V' O (t) is the input pulse voltage source V S (t) shows the maximum output voltage V' Omax (i.e., V' Omax =max{V O (t)}) is (ideally) calculated by the following formula:

number

[0043] The devices of the disclosed technology, specifically device 100 (FIG. 4A), device 200 (FIG. 6A), device 280 (FIG. 7), and device 300 (FIG. 8A), are each configured and operative to be used with various auxiliary output detection and measurement circuits and devices configured and operative to at least sense, adjust, or modify output signals from device 100, device 200, device 280, and device 300, respectively, to generate data at least related to or indicative of the pulsed voltage source. Reference is now made to FIGS. 9A, 9B, and 9C. FIG. 9A is a schematic diagram of one implementation of an auxiliary output detection and measurement comparison circuit (generally designated 400) configured for use with devices of the disclosed technology. FIG. 9B is a schematic diagram of another implementation of an auxiliary output peak / envelope detection circuit (generally designated 420) configured for use with devices of the disclosed technology. FIG. 9C is a schematic diagram of a further implementation of an auxiliary output comparator peak detector (generally designated 440) configured for use with apparatus of the disclosed techniques.

[0044] Referring to FIG. 9A, implementation 400 includes a pulsed voltage source 402 (similar to voltage source 120 (FIG. 4A), or similar to MOSFET 152 (FIG. 5), or similar to a shunt resistor) input to any one of devices 100 (FIG. 4A), 200 (FIG. 6A), and 300 (FIG. 8A) (referred to herein as "100 / 200 / 300"), and a sense (load) resistor R L 404 and a comparator 406. The pulse voltage source 402 provides a pulse source (or a single pulse) to the input of any one of the devices 100, 200, 280, and 300, and outputs V1 out9A shows the pulsed voltage source 402 of the aforementioned embodiment of each device. As shown in FIG. 9A, a sense resistor 404 is coupled to the output terminal of any one of the devices 100, 200, 280, and 300, and one terminal of a comparator 406 is coupled to one of the output terminals (and the sense resistor 404). The comparator 406 is configured to calculate an output voltage V1 out is the reference voltage V ref. Compared to V1 out and V ref. Output V1 depends on the comparison with aux-out For example, V1 out >V ref. In the case of V1 aux-out =V1 out and V1 out <V ref. In the case of V1 aux-out =V ref. (i.e., alternatively, V1 aux-out =0). Alternatively, comparator 406 is configured and operable to compare between the currents (not shown). Thus, implementation 400 may indicate whether the output of any one of apparatus 100, apparatus 200, apparatus 280, and apparatus 300 is greater than or less than a reference value (e.g., a threshold value, a user-selected value, etc.). Comparator 406 may be implemented by an operational amplifier, a dedicated integrated comparator, a comparison circuit including transistors, an integrated circuit (IC), etc.

[0045] 9B shows an implementation 420 that includes a pulsed voltage source 422 (similar to pulsed voltage source 402 (FIG. 9A)) that is input to one of devices 100, 200, 280, and 300, a buffer 424, a diode 426, a capacitor 428, a resistor 430, and a switch 432. Pulsed voltage source 422 provides a pulse source to the input of any one of devices 100, 200, 280, and 300, and its output V2 out indicates the pulsed voltage source 402 of the above-described embodiment of the respective device. Output V2 outis an input to buffer 424. Buffer 424 is connected to the anode terminal of diode 426. The cathode terminal of diode 426 is connected to one terminal of capacitor 428, the other terminal of capacitor 428 being connected to ground. Switch 432 is configured and operable to switch resistor 430 on and off. When switch 432 is closed, an RC circuit is formed from capacitor 428 and resistor 430. Buffer 424 generally provides isolation (e.g., by providing isolation between a high input impedance level and a low output impedance level) from its output to its input V2 out Buffer 424 provides an output signal (not shown) that is input to diode 426, which rectifies the output signal, the output of which charges capacitor 428. When switch 432 is open, diode 426 and capacitor 428 are configured and operable as a peak (voltage) detector such that when capacitor 428 is charged to a peak voltage, output V2 aux-out V is held at that voltage peak. Diode 426 conducts only when capacitor 428 is forward biased to charge up to the new peak (taking into account the diode forward voltage drop, e.g., 0.6 V, depending on the diode). aux-out When the output voltage peak (at the cathode) becomes greater than the input voltage (at the anode), diode 426 becomes reverse biased and does not conduct current from capacitor 428 towards the input, thereby holding or maintaining the output at the same output voltage peak (i.e., minus the diode forward voltage drop). Alternatively, diode 426 may be reversed in polarity, thereby functioning as a negative voltage peak detector (not shown) in conjunction with unpolarized capacitor 428.

[0046] When switch 432 is closed, diode 426, capacitor 428, and resistor 430 are configured and operable as an envelope detector, the output of which is V2 aux-outdecreases according to a characteristic RC time constant that determines the decay time when the input voltage (at the anode) falls below the output voltage. Conversely, when the input voltage rises above the output voltage (taking into account the diode's forward voltage drop), diode 426 becomes forward biased and V aux-out increases, resulting in the output being the envelope of the input signal.

[0047] 9C shows an implementation 440 that includes a pulsed voltage source 442 (similar to pulsed voltage source 422 (FIG. 9B)) input to one of devices 100, 200, 280, and 300, a buffer 444, a comparator 446, a switch 448, and a capacitor 450. Pulsed voltage source 442 provides a pulse source to the input of any one of devices 100, 200, 280, and 300, and its output V3 out indicates the pulsed voltage source 422 in the above-described embodiment of the respective device. Output V3 out is input to a buffer 444 and one terminal of a comparator 446. The other terminal of the comparator 446 is connected to a reference voltage V ref When switch 448 is closed, the output of buffer 444 is controlled using output V3 aux-out , and the other terminal of capacitor 450 is grounded. Implementation 440 includes a sample-and-hold (i.e., equivalently "follow-hold") circuit (i.e., having a "sample" mode and a "hold" mode). Because there is no dependency on resistors that affects the decay time, implementation 440 exhibits a relatively faster voltage envelope tracking response than implementation 420 (FIG. 9B). In sample mode, switch 448 is closed and buffer 444 charges capacitor 450, discharges capacitor 450, equalizes the capacitor voltage, or reduces V3 out In hold mode, switch 448 is open and capacitor 450 retains its charge (i.e., it may eventually discharge, typically due to leakage current). Comparator 446 compares voltage V3 out and V ref.and V ref. , and the selected value of . Alternatively, other auxiliary devices such as a voltage analyzer, a current analyzer, a voltmeter, an ammeter, or general voltage / current sensing devices may be coupled to at least one output terminal (not shown) of apparatus 100, apparatus 200, apparatus 280, and apparatus 300.

[0048] The devices 100, 200, 280, and 300 of the disclosed technology can be used to quickly detect short-circuit conditions or events in a monitored device. Generally, modern wide bandgap semiconductor power transistors, in contrast to their silicon (Si) counterparts, have limited ability to withstand avalanche modes of operation. The devices 100, 200, 280, and 300 of the disclosed technology can be incorporated into protection circuits with very short reaction times, thereby enabling power transistor-based power electronics systems to be preserved and protected in the event of a short-circuit event. Specifically, the input V IN The rise time of the pulse signal and the corresponding output induced pulse voltage signal V OUT Due to the fast response between the output and the load, the output induced pulse voltage may indicate a short circuit condition or event (i.e., characterized by a sudden (rapid) increase in current and a corresponding drop (rapid) in voltage).

[0049] Reference is now made further to FIG. 10, which is a schematic block diagram of a method (generally designated 500) for detecting and / or measuring a pulsed voltage source in accordance with the principles of the disclosed technique. Method 500 begins at step 502, in which a pulse signal of the pulsed voltage source is differentiated via a differentiator having an inductive component that is a winding of a transformer, thereby generating a differentiated signal. Referring to FIGS. 4A, 4B, and 4C, the pulse signal V of pulsed voltage source 120 (FIG. 4A) is S (t) is differentiated via a differentiator 106 having an inductive component, which is a winding (primary winding) 108 of a transformer 104 (FIGS. 4A and 4B), thereby producing a differentiated signal (Vd According to an alternative procedure (not shown in FIG. 10), the pulsed signal of the pulsed voltage source is differentiated via a differentiator having an RC circuit that is separate from (and mutually exclusive in terms of components with) the transformer. Referring to FIG. 7, differentiator 282 differentiates the pulsed signal of pulsed voltage source 120, thereby generating a differentiated signal V d Generate (t).

[0050] In step 504, the differential signal is upconverted via a transformer, thereby generating an upconverted differential signal. Referring to Figures 4A, 4B, and 4C, the differential signal (V d (t)) is upconverted (“stepped up”) via transformer 104 (FIGS. 4A and 4B), thereby producing an upconverted differential signal V O 4A, 4B, and 4C are generated. Specifically, FIG. 4C shows the output voltage V as a function of time plotted along with a plot of the time-dependent input pulse signal 142 of the pulsed voltage source 120. O A characteristic plot (generally designated 144) of (t) is shown.

[0051] In step 506, the upconverted differential signal is detected and / or measured, where the upconverted differential signal is indicative of a pulsed voltage source. Referring to Figures 4A, 4B, and 4C, the upconverted differential signal V O (t) detects or measures the voltage drop across the resistive load 122 (R 1 ) coupled to the output terminals 1161, 1162 of the device 200 by detecting or measuring the voltage drop across the resistive load 122 representing the pulsed voltage source, as well as the current flowing through the resistive load 122 representing the pulsed voltage source. L) (i.e., a "sense load," a "sense resistor," or a "resistive load," which are all synonymous terms). With further reference to FIGS. 9A, 9B, and 9C, an auxiliary output detect measure compare circuit 400 (FIG. 9A) is used to detect and measure pulsed voltage source 402. Alternatively, an auxiliary output peak / envelope detector 420 (FIG. 9B) is used to detect pulsed voltage source 422. Yet alternatively, an auxiliary output comparator peak detector 440 (FIG. 9C) is used to measure and detect pulsed voltage source 442.

[0052] Those skilled in the art will appreciate that the technology of the present disclosure is not limited to what has been particularly shown and described above. Rather, the scope of the technology of the present disclosure is defined only by the following claims.

Claims

1. 1. A measuring device for measuring a pulsed current flowing through a transistor, the transistor exhibiting an on-resistance, the measuring device comprising: a transformer having a primary winding electromagnetically coupled to a secondary winding, the primary winding having a first inductance; a differentiator including at least one resistive element coupled at one end to the primary winding and at the other end to one of two input terminals of the measuring device, thereby forming a closed circuit together with the transistor, the differentiator being configured to electromagnetically induce an induced pulse voltage on the secondary winding; Equipped with The induced pulse voltage is indicative of the pulse current and correlated to the on-resistance value, and input terminals of the measurement device are connected to the drain and source terminals of the transistor, respectively.

2. 2. The measurement device of claim 1, wherein the at least one resistive element is a resistor coupled in series with the primary winding.

3. The measurement device of claim 1 , wherein the first inductance is non-zero.

4. 10. The measurement device of claim 1, wherein the secondary winding has an output terminal configured for coupling to an external resistive load.

5. 2. The measuring device according to claim 1, wherein the maximum value of the induced pulse voltage is proportional to a turns ratio between the secondary winding and the primary winding.

6. 6. The measurement device of claim 5, wherein the maximum value is 5 volts.

7. 5. The measurement device of claim 4, wherein the external resistive load allows measurement of a current flowing therethrough.

8. 2. The measurement device of claim 1, wherein the transistor is a metal oxide semiconductor field effect transistor (MOSFET).

9. 2. The measurement device of claim 1, further comprising a direct current (DC) decoupler between the at least one resistive element and the transistor, the DC decoupler configured to block at least one time-invariant DC component of the pulsed current.

10. 10. The measurement device of claim 9, wherein the DC decoupler is a capacitor coupled in series with the at least one resistive element.

11. 10. The measurement device of claim 9, wherein the DC decoupler is configured to provide high voltage (HV) protection for the measurement device.

12. 10. The measurement device of claim 9, wherein the peak-to-peak voltage of the induced pulse voltage exceeds 1 volt.

13. The secondary winding is A comparator; a peak / envelope detector; a comparator peak detector; A voltage analyzer, a current analyzer; A voltmeter and An ammeter, 10. The measurement device of claim 1, further comprising an output terminal configured to couple to at least one auxiliary device selected from the list consisting of:

14. 1. A measurement method for measuring a pulse current flowing through a transistor, the transistor exhibiting an on-resistance, the measurement method comprising: adjusting the switching of the pulsed current through the transistor; differentiating the pulse signal of the pulse current via a differentiator having an inductive component which is a winding of a transformer to generate a differentiated signal; upconverting the differential signal via the transformer to generate an upconverted differential signal; measuring the upconverted differential signal, the upconverted differential signal being indicative of the pulse current and correlated to the on-resistance value; , including, a measurement method.

15. 15. The method of claim 14, wherein the winding is a primary winding of the transformer.

16. 15. The method of claim 14, wherein the upconversion has an upconversion factor proportional to a turns ratio of the transformer.

17. 17. The method of claim 16, wherein the maximum value of the induced pulse voltage is proportional to the turns ratio.

18. 18. The method of claim 17, wherein the maximum value is 5 volts.

19. 15. The method of claim 14, wherein the measurement is a measurement of current flowing through a sensing device coupled to a secondary winding of the transformer.

20. blocking at least one time-varying direct current (DC) component of the pulsed current before differentiating the pulse signal of the pulsed current. The measurement method of claim 14 further comprising:

21. 21. The method of claim 20, wherein the disconnecting provides high voltage (HV) protection to the device.

22. 21. The method of claim 20, wherein the peak-to-peak voltage of the induced pulse voltage is greater than 1 volt.

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