Synchronous motor control device, electric vehicle, and synchronous motor control method

Abstract

A synchronous motor control device according to the present invention is provided with: a torque command value conversion unit that converts a torque command value for a synchronous motor to a second torque command value gradually increasing in a region including the peak torque of the synchronous motor; a difference calculation unit that calculates a difference between the torque of the synchronous motor and the second torque command value; a voltage phase control unit that controls a voltage phase angle so that the difference becomes smaller; and a power conversion unit that converts DC power to AC power on the basis of the voltage phase angle and outputs the converted AC power to the synchronous motor. When a code of the torque command value is inverted from that at a current operation point of the synchronous motor, the difference calculation unit corrects the difference by calculating the second torque command value on the basis of a prescribed correction value.

Description

Technical Field

[0001] The present invention relates to a synchronous motor control device, an electric vehicle, and a synchronous motor control method.

Background Art

[0002] DC power is converted into AC power by a power conversion unit, and a rectangular wave voltage is applied to a synchronous motor by voltage phase control to rotationally drive the synchronous motor. According to this voltage phase control, the output in the high-speed rotation range of the synchronous motor can be improved, and the number of switching operations in the power conversion unit can be reduced, suppressing switching loss. And, in order to improve the voltage utilization rate of the synchronous motor, 1-pulse control in which the number of pulses in one cycle of the rectangular wave voltage is reduced is used. In 1-pulse control, torque control is performed by operating the voltage phase based on the torque of the synchronous motor and the torque command value.

[0003] Patent Document 1 discloses a motor control device including a voltage command value calculation unit that calculates a voltage command value used for voltage phase control according to a torque command value, a phase calculation unit that calculates a phase command value of an applied voltage according to the torque command value, a first estimation unit that estimates the torque generated in the motor as a first torque based on a current command value, a second estimation unit that estimates the torque generated in the motor as a second torque based on the current value flowing through the motor, and a correction unit that corrects the phase command value to calculate a voltage command value so that the deviation between the first torque and the second torque becomes small.

Prior Art Documents

Patent Documents

[0004]

Patent Document 1

Summary of the Invention

Problems to be Solved by the Invention

[0005] Figure 17 shows the relationship between the voltage phase angle θv and the output torque. Here, θv is set to 0 degrees with respect to the q-axis. Up to the maximum (minimum) torque, the torque increases (decreases) monotonically with respect to the voltage phase angle θv. However, beyond the maximum (minimum) torque, the torque begins to decrease (increase). Therefore, it is necessary to control the torque within a range where the voltage phase angle θv does not exceed the maximum (minimum) torque, and it is necessary to limit the torque command value (or voltage phase angle θv) by providing a margin from the maximum (minimum) torque. Consequently, in methods such as Patent Document 1, where the voltage phase angle θv is manipulated based on estimated torque calculated from the current value, the torque that the motor can actually output is limited compared to the torque that the motor is originally capable of outputting. Therefore, this problem can be solved by manipulating the voltage phase angle θv using a physical quantity that increases monotonically even near the maximum (minimum) torque. For example, the d-axis current id or d-axis magnetic flux Φd can be used. Figure 18 shows the relationship between the voltage phase angle θv and the d-axis magnetic flux Φd, and Figure 19 shows the relationship between the voltage phase angle θv and the d-axis current id. In the case of d-axis magnetic flux Φd and d-axis current id, the voltage phase angle θv increases monotonically until it reaches 180 degrees (-180 degrees), so it operates stably even near the maximum (minimum) torque. The problem that this invention aims to solve is that when manipulating the voltage phase angle based on torque command values ​​such as d-axis magnetic flux Φd and d-axis current id, if the sign of the torque command changes, the deviation cannot be calculated correctly, overshoot occurs, and it becomes impossible to achieve the first-order lag response as designed. [Means for solving the problem]

[0006] The synchronous motor control device according to the present invention includes: a torque command value conversion unit that converts a torque command value to a synchronous motor into a second torque command value that is gradually increasing in a region including the peak torque of the synchronous motor; a difference calculation unit that calculates the difference between the torque of the synchronous motor and the second torque command value; a voltage phase control unit that controls the voltage phase angle so that the difference becomes small; and a power conversion unit that converts DC power into AC power based on the voltage phase angle and outputs the converted AC power to the synchronous motor. The difference calculation unit corrects the difference by calculating the second torque command value based on a predetermined correction value when the sign of the torque command value reverses from the current operating point of the synchronous motor. The synchronous motor control method according to the present invention is a synchronous motor control method in a synchronous motor control device equipped with a power conversion unit that converts DC power to AC power based on a voltage phase angle and outputs the converted AC power to a synchronous motor, wherein the torque command value to the synchronous motor is converted to a second torque command value that is gradually increasing in a region including the peak of the torque of the synchronous motor, the difference between the torque of the synchronous motor and the second torque command value is calculated, the voltage phase angle is controlled so that the difference becomes smaller, and the difference is corrected by calculating the second torque command value based on a predetermined correction value when the sign of the torque command value reverses from the current operating point of the synchronous motor. [Effects of the Invention]

[0007] According to the present invention, even if the sign of the torque command changes, no overshoot occurs, and a first-order lag response as designed can be achieved. [Brief explanation of the drawing]

[0008] [Figure 1] Figure 1 is a block diagram of the synchronous motor control device in the first embodiment. [Figure 2] Figure 2 is a block diagram of the control unit in the first embodiment. [Figure 3] Figure 3 is a block diagram of the difference calculation unit in the first embodiment. [Figure 4] Figure 4 shows the torque operating point on a coordinate system represented by the d-axis magnetic flux and q-axis magnetic flux in the first embodiment. [Figure 5] Figure 5 is a block diagram of the second difference calculation unit in the first embodiment. [Figure 6] Figures 6(A), 6(B), 6(C), 6(D), and 6(E) are graphs showing the response waveforms when the first embodiment is applied. [Figure 7] Figure 7 is a block diagram showing a modified example of the second difference calculation unit in the first embodiment. [Figure 8] Figure 8 shows an example of a lookup table in a modified version of the first embodiment. [Figure 9] Figure 9 shows the torque operating point on a coordinate system represented by the d-axis magnetic flux and q-axis magnetic flux in the comparative example. [Figure 10] Figure 10 is a block diagram showing the second difference calculation unit in the comparative example. [Figure 11] Figures 11(A), 11(B), 11(C), 11(D), and 11(E) are graphs showing the response waveforms in comparative examples. [Figure 12] Figure 12 is a block diagram of the difference calculation unit in the second embodiment. [Figure 13] Figure 13 shows the torque operating point on a coordinate system represented by the d-axis current and q-axis current in the second embodiment. [Figure 14] Figure 14 is a block diagram of the second difference calculation unit 302 in the second embodiment. [Figure 15] Figure 15 is a block diagram showing a modified example of the second difference calculation unit in the second embodiment. [Figure 16] Figure 16 shows an example of a lookup table in the second embodiment. [Figure 17] Figure 17 is a graph showing the relationship between the voltage phase angle and the torque output from a synchronous motor. [Figure 18] Figure 18 is a graph showing the relationship between voltage phase angle and d-axis magnetic flux. [Figure 19] FIG. 19 is a graph showing the relationship between the voltage phase angle and the d-axis current. [Figure 20] FIG. 20 is a configuration diagram of an electric vehicle in the third embodiment. MODE FOR CARRYING OUT THE INVENTION

[0009] Hereinafter, embodiments of the present invention will be described with reference to the drawings. The following description and drawings are examples for explaining the present invention, and for the sake of clarity of explanation, omissions and simplifications are made as appropriate. The present invention can be implemented in various other forms. Unless otherwise particularly limited, each component may be in a single or plural number.

[0010] The positions, sizes, shapes, ranges, etc. of the respective components shown in the drawings may not represent the actual positions, sizes, shapes, ranges, etc. in order to facilitate understanding of the invention. Therefore, the present invention is not necessarily limited to the positions, sizes, shapes, ranges, etc. disclosed in the drawings.

[0011] [First Embodiment] FIG. 1 is a block configuration diagram of the synchronous motor control device 1 in the present embodiment. The synchronous motor control device 1 converts DC power from the DC power supply 2 into AC power and drives the synchronous motor 3 with the converted AC power.

[0012] The synchronous motor (PMSM: Permanent Magnet Synchronous Motor) 1 has, for example, a rotating field type configuration in which a permanent magnet is provided on the rotor and armature windings are provided on the stator. The synchronous motor 3 operates by the interaction between the magnetic flux generated from the permanent magnet provided on the rotor (not shown) and the magnetic field generated by the current of each phase flowing through the three-phase windings fixed to the armature. Note that the synchronous motor 3 is not limited to a permanent magnet synchronous motor, and any synchronous motor such as a wound field synchronous motor may be used. The synchronous motor 3 is a drive source of the electric vehicle, and the rotation of the synchronous motor 3 propels the electric vehicle.

[0013] The synchronous motor 3 is equipped with a magnetic pole position sensor 31. The magnetic pole position sensor 31 has the function of detecting the magnetic pole position of the rotor of the synchronous motor 3 as a phase relative to the stator. The magnetic pole position sensor 31 outputs the detected rotor phase θ, which is input to the control unit 100 of the synchronous motor control device 1. The magnetic pole position sensor 31 can be configured as a resolver, rotary encoder, absolute encoder, etc.

[0014] The DC power supply 2 supplies DC power to the power conversion unit 400 of the synchronous motor control device 1. The DC power supply 2 is, for example, a rechargeable battery.

[0015] The synchronous motor control device 1 comprises a control unit 100, a voltage detector 200, a gate drive circuit 300, a power conversion unit 400, and a current detector 500. The control unit 100 generates a gate signal GS based on a torque command value T*, etc., provided by a higher-level controller (not shown) of the electric vehicle, and performs torque control of the synchronous motor 3. Details of the generation of the gate signal GS will be described later.

[0016] The voltage detector 200 is connected to the high-potential wiring 133 and the low-potential wiring 134 of the DC power supply 2, and detects the potential difference between them. Since this potential difference is usually a high voltage, for example, 100V or more, the voltage detector 200 converts it to a low voltage that the control unit 100 can detect, and outputs the value of this converted DC voltage Vdc to the control unit 100.

[0017] The gate drive circuit 300 converts the gate signal GS generated by the control unit 100 into the potential necessary to switch the switching elements 114, 115, 116, 117, 118, and 119 in the power conversion unit 400 on and off, and outputs it to the power conversion unit 400.

[0018] The power conversion unit 400 comprises a smoothing capacitor 138 and switching elements 114 to 119 that constitute an inverter with three phase upper and lower arms. The smoothing capacitor 138 suppresses the pulsation of the DC voltage generated by the switching operation of the switching elements 114 to 119. The power conversion unit 400 operates the switching elements 114 to 119 according to the gate signal GS described later, converting the DC power supplied from the DC power supply 2 into AC power to drive the synchronous motor 3.

[0019] The inverter includes switching elements 114, 115, 116, 117, 118, and 119, and freewheeling diodes 120, 121, 122, 123, 124, and 125. Switching elements 114 to 119 are, for example, Si-IGBTs and include a gate terminal, a collector terminal, and an emitter terminal. Freewheeling diodes 120 to 125 are connected between the collector terminal and emitter terminal of each switching element 114 to 119. When the collector terminal of switching elements 114 to 119 is at a higher potential than the emitter terminal, freewheeling diodes 120 to 125 allow current to flow through them, preventing a high reverse voltage from being applied to the switching elements 114 to 119. However, the inverter is not limited to a combination of Si-IGBTs and freewheeling diodes, and may be constructed using other semiconductor elements.

[0020] The switching of switching elements 114 to 119 is performed by the gate signal GS connected to the gate terminal of each switching element 114 to 119.

[0021] The emitter terminal of switching element 114 and the collector terminal of switching element 115 are connected to each other, and this connection point is connected to the U-phase wiring 105, allowing a U-phase current iu to flow. The emitter terminal of switching element 116 and the collector terminal of switching element 117 are connected to each other, and this connection point is connected to the V-phase wiring 106, allowing a V-phase current iv to flow. The emitter terminal of switching element 118 and the collector terminal of switching element 119 are connected to each other, and this connection point is connected to the W-phase wiring 107, allowing a W-phase current iw to flow. The collector terminals of switching elements 114, 116, and 118 are connected to each other and connected to the high-potential wiring 133. In addition, the emitter terminals of switching elements 115, 117, and 119 are connected to each other and connected to the low-potential wiring 134.

[0022] The current detector 500 detects the current flowing through each phase wiring 105, 106, and 107, and outputs the detected values ​​of each phase current iu, iv, and iw to the control unit 100.

[0023] The control unit 100 receives a torque command value T* from the higher-level controller, a DC voltage Vdc from the voltage detector 200, phase currents iu, iv, and iw from the current detector 500, and rotor phase θ from the magnetic pole position sensor 31. The control unit 100 refers to the DC voltage Vdc, phase currents iu, iv, and iw, and rotor phase θ, and generates a gate signal GS that turns the switching elements 114, 115, 116, 117, 118, and 119 on and off at the appropriate timing according to the torque command value T*. This gate signal GS controls the phase currents iu, iv, and iw flowing through the windings of the synchronous motor 3, thereby achieving torque control of the synchronous motor 3. The gate signal GS is a PWM signal (pulse width modulation signal) in which each phase current iu, iv, and iw are sinusoidal signals with a 120-degree phase difference from each other. As will be explained in more detail later, the PWM signal is generated based on rectangular wave-shaped three-phase voltage command values ​​Vu*, Vv*, and Vw*, which are based on the voltage phase angle θv and rotor phase θ.

[0024] In this embodiment, the control unit 100 uses one-pulse control, which reduces the number of pulses in one period of the rectangular wave three-phase voltage command values ​​Vu*, Vv*, and Vw* to one, in order to improve the voltage utilization rate of the synchronous motor 3. Since the voltage amplitude is fixed in one-pulse control, voltage phase control is performed. Voltage phase control is performed by manipulating the voltage phase angle θv based on the torque of the synchronous motor 3 and the torque command value to control the torque.

[0025] Figure 2 is a block diagram of the control unit 100 in the first embodiment. The control unit 100 includes a torque command value conversion unit 201, a three-phase to two-phase conversion unit 202, a magnetic flux calculation unit 203, a difference calculation unit 204, a velocity calculation unit 205, a voltage phase control unit 206, a square wave generation unit 207, and a PWM gate signal generation unit 208.

[0026] The control unit 100 controls the torque by supplying three-phase currents iu, iv, and iw to the synchronous motor 3. In this case, the control unit 100 internally transforms the three-phase fixed coordinates and uses a so-called current vector control method, which controls the motor with two-phase rotating coordinates synchronized with the rotor phase represented by the d-axis and q-axis.

[0027] The torque command value conversion unit 201 receives the torque command value T*, the DC voltage Vdc, and the rotational angular velocity ω as inputs, and calculates the d-axis current command value id* and the q-axis current command value iq* from these inputs. Furthermore, the torque command value conversion unit 201 calculates the d-axis magnetic flux command value Φd* and the q-axis magnetic flux command value Φq* by referring to a lookup table or the like based on the calculated d-axis current command value id* and q-axis current command value iq*.

[0028] Furthermore, the magnetic flux limit value Φlim is calculated based on the following equation (1). The magnetic flux limit value Φlim represents the amplitude of the d-axis magnetic flux and the q-axis magnetic flux, and is calculated by the following equation (1), ignoring the voltage drop across the motor winding resistance R. Φlim=sqrt(Φd^2+Φq^2)=Vdc×Kv / ω ···(1) Here, Kv is the voltage utilization coefficient, which is 2 / π in the case of 1-pulse control. The rotational angular velocity ω is calculated by the velocity calculation unit 205 based on the rotor phase θ. sqrt is an abbreviation for square root.

[0029] The torque command value conversion unit 201 converts the torque command value T*, which is the first torque command value to the synchronous motor 3, into a second torque command value that is gradually increasing in region A near the torque peak of the synchronous motor 3. An example of the second torque command value is the d-axis magnetic flux command value Φd* or the d-axis current command value id*. The reason for using a second torque command value that is gradually increasing in the region including the torque peak of the synchronous motor 3 will be explained later. The converted d-axis magnetic flux command value Φd* and q-axis magnetic flux command value Φq*, the d-axis current command value id* and q-axis current command value iq*, and the calculated magnetic flux limit value Φlim are output to the difference calculation unit 204.

[0030] The three-phase to two-phase conversion unit 202 converts the values ​​of each phase current iu, iv, and iw from the current detector 500 to the d-axis and q-axis based on the rotor phase θ information, and outputs the d-axis current id and q-axis current iq to the difference calculation unit 204 and the magnetic flux calculation unit 203.

[0031] The magnetic flux calculation unit 203 calculates the d-axis magnetic flux Φd and the q-axis magnetic flux Φq using a lookup table or the like based on the d-axis current id and the q-axis current iq, and outputs them to the difference calculation unit 204.

[0032] The difference calculation unit 204 receives the d-axis current command value id* and q-axis current command value iq*, the d-axis current id and q-axis current iq as inputs, as well as the d-axis magnetic flux command value Φd* and q-axis magnetic flux command value Φq*, the d-axis magnetic flux Φd, the q-axis magnetic flux Φq, and the magnetic flux limit value Φlim. Based on these input values, it calculates the first difference ΔT1, which is the difference between the torque of the synchronous motor 3 and the first torque command value, and the second difference ΔT2, which is the difference between the torque of the synchronous motor 3 and the second torque command value. If the sign of the torque command value reverses from the current operating point of the synchronous motor 3, a predetermined correction value, such as the magnetic flux limit value Φlim, is added to the second torque command value to calculate the second difference ΔT2. Details will be described later. The calculated first difference ΔT1 and second difference ΔT2 are output to the voltage phase control unit 206.

[0033] The voltage phase control unit 206 outputs a voltage phase angle θv such that the difference becomes small, i.e., zero, based on the first difference ΔT1 or the second difference ΔT2. Based on the rotational angular velocity ω, the voltage phase control unit 206 determines, for example, using a weighted average, which is the region A near the peak of the first torque, which is the torque command value T*. Then, using the second difference ΔT2 in region A and the first difference ΔT1 in region B, it performs feedback control so that the differences in each region become zero, and outputs the voltage phase angle θv. The feedback control is for example, PI control. The voltage phase control unit 206 may also perform feedback control using only the second difference ΔT2. In this case, the difference calculation unit 204 calculates only the second difference ΔT2.

[0034] The rectangular wave generation unit 207 calculates and outputs rectangular wave-shaped three-phase voltage command values ​​Vu*, Vv*, and Vw* using the SIGN function or the like based on the voltage phase angle θv and rotor phase θ. The PWM gate signal generation unit 208 calculates the PWM duty cycle based on the three-phase voltage command values ​​Vu*, Vv*, and Vw*. Then, taking into account dead time and other factors, it generates six gate signals GS, which are PWM signals, from the PWM duty cycle and outputs them to the gate drive circuit 300.

[0035] Figure 3 is a block diagram of the difference calculation unit 204 in the first embodiment. The difference calculation unit 204 includes a first difference calculation unit 301, a second difference calculation unit 302, and a low-pass filter 303.

[0036] The first difference calculation unit 301 takes the d-axis current command value id* and q-axis current command value iq*, the d-axis magnetic flux command value Φd* and q-axis magnetic flux command value Φq*, the d-axis current id and q-axis current iq, the d-axis magnetic flux Φd and q-axis magnetic flux Φq as input and outputs the first torque T1, the first torque command value T1*, and the first difference ΔT1. The first torque T1, the first torque command value T1*, and the first difference ΔT1 are calculated by the following equations (2), (3), and (4), respectively. T1 = Φd × iq - Φq × id ... (2) T1*=Φd*×iq*-Φq*×id* ···(3) ΔT1 = T1 - T1 ... (4)

[0037] The low-pass filter 303 is a low-pass filter that simulates a current control response, and takes the q-axis magnetic flux command value Φq* as input and outputs the filtered q-axis magnetic flux command value ΦqLPF*. However, the output to the second difference calculation unit 302 may be the q-axis magnetic flux command value Φq* before filtering.

[0038] The second difference calculation unit 302 takes the first torque T1, the first torque command value T1*, the d-axis magnetic flux Φd, the d-axis magnetic flux command value Φd*, and the filtered q-axis magnetic flux command value ΦqLPF* as inputs and calculates the second difference ΔT2. Details will be described later with reference to Figure 5.

[0039] Now, with reference to Figure 4, we will explain the change in the sign of the torque. Figure 4 shows the torque operating point on a coordinate system represented by the d-axis magnetic flux and q-axis magnetic flux in this embodiment. Figure 4 shows the transition of the operating point when the torque sign changes from positive to negative. The current torque, represented by the d-axis magnetic flux Φd, is the operating point C, and the target torque, represented by the d-axis magnetic flux command value Φd*, is the operating point D. Furthermore, the d-axis magnetic flux Φd0 represents the d-axis magnetic flux Φd at the point where the q-axis magnetic flux Φq is 0.

[0040] During single-pulse control, the voltage amplitude is fixed, and the magnetic flux amplitude is also fixed. Therefore, the operating point moves along the circumference of the magnetic flux limiting circle Φ (=sqrt(Φd^2+Φq^2)), as shown in Figure 4. In other words, since the voltage amplitude is fixed during single-pulse control, the amplitude of the magnetic flux is also fixed. Therefore, the magnetic flux moves along the circumference of an equimagnetic flux circle. If the distance traveled along this equimagnetic flux circle can be simulated, the movement distance ΔΦd of the operating point can be calculated correctly.

[0041] When the torque command value changes in a stepwise manner from positive to negative, as shown in Figure 4, the current operating point C, indicated by the d-axis magnetic flux Φd, moves along the circumference of the magnetic flux limiting circle Φ, via the d-axis magnetic flux Φd0, to the operating point D indicated by the d-axis magnetic flux command value Φd*. Therefore, the movement distance ΔΦd of the operating point is expressed by the following equation (5). ΔΦd=Φd0-Φd+Φd0-Φd*=2Φd0-Φd*-Φd ···(5)

[0042] The second difference ΔT2 is expressed by equation (6) using ΔΦd. ΔT2 = ΔΦd × ΦqLPF* ... (6) By applying equation (5) to equation (6), the second difference ΔT2 when the torque sign changes can be calculated using the following equation (7). ΔT2=(2Φd0-Φd*-Φd)ΦqLPF* ···(7)

[0043] Since the d-axis magnetic flux Φd0 is the d-axis magnetic flux Φd when the q-axis magnetic flux Φq=0, substituting Φd=Φd0 and Φq=0 into equation (1), the magnetic flux limit Φlim can be calculated using the following equation (8). Φlim=sqrt(Φd0^2+0^2)=Φd0 ···(8)

[0044] Therefore, in this embodiment, when the torque sign changes from positive to negative or from negative to positive (T1 × T1 * < 0), the difference is corrected by calculating the second difference ΔT2 using the following equation (9). When the torque sign does not change (T1 × T1 * >= 0), the second difference ΔT2 is calculated using the following equation (10). ΔT2=(2Φlim-Φd*-Φd)×ΦqLPF* ···(9) ΔT2=(Φd*-Φd)×ΦqLPF* (10)

[0045] Figure 5 is a block diagram of the second difference calculation unit 302 in the first embodiment. The second difference calculation unit 302 is configured to calculate the second difference ΔT2 based on equation (9) when T1 × T1 * < 0, and based on equation (10) when T1 × T1 * > = 0.

[0046] The multiplier 312 doubles the magnetic flux limit value Φlim, and the adder / subtractor 322 subtracts the d-axis magnetic flux command value Φd* from this doubled magnetic flux limit value Φlim, outputting the d-axis corrected magnetic flux command value Φd** to one side of the switch 332. The other side of the switch 332 receives the d-axis magnetic flux command value Φd* as input.

[0047] The multiplier 342 multiplies the first torque T1 by the first torque command value T1*. The sign determiner 352 determines the sign of this multiplication result and outputs it to the switch 332. Note that the sign determiner 352 may use other determination methods, such as checking whether the signs of the first torque T1 and the first torque command value T1* match, rather than relying on the multiplication result from the multiplier 342. The switch 332, upon receiving the determination from the sign determiner 352, outputs the d-axis corrected magnetic flux command value Φd** if the sign of the torque command value has reversed. If the sign of the torque command value has not reversed, it outputs the d-axis magnetic flux command value Φd*. In other words, during the period when the sign of the torque command value has reversed, the switch 332 outputs the d-axis corrected magnetic flux command value Φd** instead of the d-axis magnetic flux command value Φd*.

[0048] The adder / subtractor 362 subtracts the d-axis magnetic flux Φd from the d-axis magnetic flux command value Φd* or d-axis corrected magnetic flux command value Φd** output from the switch 332 to calculate the movement distance ΔΦd of the operating point. This movement distance ΔΦd is multiplied by the filtered q-axis magnetic flux command value ΦqLPF* in the multiplier 372 and output as the second difference ΔT2.

[0049] In other words, it is determined whether the sign of the torque command value has reversed relative to the current operating point of the synchronous motor 3. If the sign has changed, the second difference ΔT2 is calculated using equation (9), otherwise using equation (10). This makes it possible to achieve a first-order lag response even when the sign of the torque command value changes.

[0050] Figures 6(A), 6(B), 6(C), 6(D), and 6(E) are graphs showing the response waveforms when this embodiment is applied. Figure 6(A) shows the d-axis current id and the d-axis current command value id*, Figure 6(B) shows the q-axis current iq and the q-axis current command value iq*, Figure 6(C) shows the first torque T1 and the first torque command value T1*, Figure 6(D) shows the d-axis magnetic flux Φd and the d-axis magnetic flux command value Φd*, and Figure 6(E) shows the phase current. In each graph, the horizontal axis represents time, and the vertical axis represents values ​​normalized with the maximum value set to 1. In the graphs shown in Figures 6(A) to 6(D), the command value is shown as a dotted line, and the response waveform of the command value is shown as a solid line. In the graph shown in Figure 6(E), the U-phase current iu is shown as a solid line, the V-phase current iv as a dotted line, and the W-phase current iw as a dotted line with short dot spacing.

[0051] Each response waveform, as shown in Figure 6(C), is an example where the first torque T1, which is the current operating point, is negative, and the first torque command value T1* changes in a step-like manner from positive to negative. In this case, during the period Tn in which the sign of the torque command value is reversed, the d-axis corrected magnetic flux command value Φd** (see Figure 5) is output instead of the d-axis magnetic flux command value Φd*, as shown in Figure 6(D). Note that in Figure 6(D), the dotted line shows the d-axis magnetic flux command value Φd*, but the dotted line during the period Tn in which the sign of the torque command value is reversed shows the d-axis corrected magnetic flux command value Φd**.

[0052] As shown in Figure 6(C), the first torque T1 in response to the first torque command value T1* exhibits a first-order lag response. As shown in Figure 6(B), the q-axis current iq also exhibits a first-order lag response. Furthermore, as shown in Figure 6(A), the d-axis current id approaches 0A initially, but then responds to the target current, the d-axis current command value id*, without oscillation. This indicates that it operates along the circumference of the magnetic flux limiting circle Φ. Also, as shown in Figure 6(E), the amplitude of the phase current does not increase during the response. In this way, even when the sign of the torque command value changes, the response does not become oscillating, and a first-order lag response can be achieved. Cases where this embodiment is not applied will be described later with reference to Figure 11.

[0053] Figure 7 is a block diagram showing a modified example of the second difference calculation unit 302 in the first embodiment. The difference from the second difference calculation unit 302 shown in Figure 5 is that the d-axis magnetic flux Φd0 is used as the input to the doubler 312. The same reference numerals are used for parts that are the same as in Figure 5 for the sake of simplicity in explanation.

[0054] In equation (8), we show a method of calculating Φd0 using the magnetic flux limit value Φlim calculated in equation (1). This is because, in the high rotational speed region, the voltage drop across the motor winding resistance is considered to be sufficiently small compared to the speed electromotive force, and can be approximated as Φd0 ≈ Φlim as in equation (8).

[0055] If the voltage drop across the motor winding resistance R cannot be ignored, the magnetic flux limit Φlim in equation (1) is expressed by the following equation (11). Φlim = sqrt((R × id / ω - Φq)^2 +(R×iq / ω+Φd)^2) ···(11)

[0056] The d-axis magnetic flux Φd0 is the d-axis magnetic flux Φd when the q-axis magnetic flux Φq=0 (iq=0). Substituting Φd=Φd0, Φq=0, and iq=0 into equation (11) gives equation (12). The d-axis current id0 represents the d-axis current id at the point where the d-axis magnetic flux Φd0 is. Φd0=sqrt(Φlim^2-(R×id0 / ω)^2)···(12)

[0057] The second difference ΔT2 can be calculated by substituting the d-axis magnetic flux Φd0 expressed in equation (12) into equation (7). However, directly implementing equation (12) requires calculating the d-axis current id0, and it is difficult to calculate this in real time considering the nonlinearity of the motor inductance, etc. Therefore, the d-axis magnetic flux Φd0 is calculated by referencing the lookup table 382 based on the magnetic flux limit value Φlim and DC voltage Vdc calculated in equation (1), taking into account the motor winding resistance R.

[0058] As shown in Figure 7, the lookup table 382 outputs the d-axis magnetic flux Φd0 corresponding to the input magnetic flux limit value Φlim and DC voltage Vdc. Details of the lookup table 382 will be described later. The d-axis magnetic flux Φd0 is input to the multiplier 312, and the second difference ΔT2 is calculated as described below with reference to Figure 5.

[0059] Figure 8 shows an example of a lookup table 382 in a modified version of the first embodiment. The x-axis represents the magnetic flux limit value Φlim, the y-axis represents the DC voltage Vdc, and the z-axis represents the d-axis magnetic flux Φd0. In equation (12), if the magnetic flux limit Φlim is constant, when the DC voltage Vdc is high, the rotational angular velocity ω is large, and the effect of the voltage drop in the second term on the right-hand side becomes small, so the d-axis magnetic flux Φd0 is almost equal to the magnetic flux limit Φlim. Conversely, when the DC voltage Vdc is small, the rotational angular velocity ω becomes small, and the effect of the voltage drop becomes large, so the d-axis magnetic flux Φd0 is small relative to the magnetic flux limit Φlim. Therefore, as shown in Figure 8, when the DC voltage Vdc is low, the change in the d-axis magnetic flux Φd0 is large with respect to the DC voltage Vdc, and when the DC voltage Vdc is high, it is almost constant.

[0060] Figure 9 shows the torque operating point on a coordinate system represented by the d-axis magnetic flux and q-axis magnetic flux in a comparative example. This comparative example illustrates a case where this embodiment is not applied, and is shown to deepen the understanding of this embodiment. Figure 9, similar to Figure 4, shows the transition of the operating point when the torque sign changes from positive to negative. The current torque, represented by the d-axis magnetic flux Φd, is the operating point C, and the target torque, represented by the d-axis magnetic flux command value Φd*, is the operating point D. Furthermore, the d-axis magnetic flux Φd0 represents the d-axis magnetic flux Φd at the point where the q-axis magnetic flux Φq is 0.

[0061] As described with reference to Figure 4, during single-pulse control, the voltage amplitude is fixed, and the magnetic flux moves along the equiflux circle indicated by the magnetic flux limiting circle Φ (=sqrt(Φd^2+Φq^2)). Therefore, if the distance traveled along the equiflux circle can be simulated, the movement distance ΔΦd of the operating point can be calculated correctly.

[0062] When the torque command value changes in a stepwise manner from positive to negative, as shown in Figure 4, for example, the current operating point C, indicated by the d-axis magnetic flux Φd, moves along the circumference of the magnetic flux limiting circle Φ, via the d-axis magnetic flux Φd0, to the operating point D indicated by the d-axis magnetic flux command value Φd*. In order to correctly calculate the travel distance ΔΦd, it is necessary to use equation (6), as already mentioned. However, in comparative examples to which this embodiment is not applied, the travel distance ΔΦd is simply calculated as shown in equation (13) below, regardless of the change in the torque sign. ΔΦd = Φd * - Φd ... (13) In other words, in the comparative example, the displacement distance ΔΦd of the operating point passing along the circumference of the equiflux circle cannot be calculated correctly.

[0063] The second difference ΔT2 is then expressed by equation (14) from equation (6) above. ΔT2=(Φd*-Φd)ΦqLPF* ···(14)

[0064] Figure 10 is a block diagram showing the second difference calculation unit 302 in the comparative example. This comparative example is the block diagram for the comparative example described with reference to Figure 9. Figure 10 shows the configuration for calculating the second difference ΔT2 based on equation (14). The d-axis magnetic flux Φd is subtracted from the d-axis magnetic flux command value Φd* by the adder / subtractor 362 to calculate the movement distance ΔΦd of the operating point. This movement distance ΔΦd is multiplied by the filtered q-axis magnetic flux command value ΦqLPF* by the multiplier 372 and output as the second difference ΔT2.

[0065] In the comparative examples shown in Figures 9 and 10, regardless of the change in the torque sign, the difference between the d-axis magnetic flux Φd and the d-axis magnetic flux command value Φd* is simply calculated as shown in equation (13), resulting in a small calculation of the second difference ΔT2. As a result, the expected step response cannot be achieved when the sign of the torque command value changes.

[0066] Figures 11(A), 11(B), 11(C), 11(D), and 11(E) are graphs showing the response waveforms in the comparative examples. These show the cases of the comparative examples shown in Figures 9 and 10. Figure 11(A) shows the d-axis current id and the d-axis current command value id*, Figure 11(B) shows the q-axis current iq and the q-axis current command value iq*, Figure 11(C) shows the first torque T1 and the first torque command value T1*, Figure 11(D) shows the d-axis magnetic flux Φd and the d-axis magnetic flux command value Φd*, and Figure 11(E) shows the phase current. In each graph, the horizontal axis represents time, and the vertical axis represents values ​​normalized with the maximum value set to 1. In the graphs shown in Figures 11(A) to 11(D), the command value is shown as a dotted line, and the response waveform of the command value is shown as a solid line. In the graph shown in Figure 11(E), the U-phase current iu is shown as a solid line, the V-phase current iv as a dotted line, and the W-phase current iw as a dotted line with short dot spacing.

[0067] Each response waveform, as shown in Figure 11(C), is an example of a situation where the first torque T1, which is the current operating point, is negative, and the first torque command value T1* changes in a step-like manner from positive to negative. In this embodiment, as shown in Figure 6(D), a d-axis magnetic flux command value Φd* calculated based on a predetermined correction value is input for a predetermined period of time in response to a change in the torque sign. On the other hand, in this comparative example, such control is not implemented.

[0068] As shown in Figure 11(C), the first torque T1 in response to the first torque command value T1* exhibits an oscillatory response. As shown in Figure 11(B), the q-axis current iq also exhibits an oscillatory response. Furthermore, as shown in Figure 11(A), the oscillation of the d-axis current id increases, and then it responds to the target current, which is the d-axis current command value id*. Also, as shown in Figure 11(E), the amplitude of the phase current increases during the response. This indicates that there is a risk of overcurrent. Thus, when the sign of the torque command value changes, undershoot (overshoot) occurs in the first torque T1 and the q-axis current iq, and a first-order lag response is not achieved.

[0069] According to the first embodiment, when the sign of the torque command reverses from the current operating point of the synchronous motor 3, the difference between the torque of the synchronous motor 3 and the command value (d-axis magnetic flux command value Φd*) can be correctly calculated, thereby suppressing overcurrent and undershoot (overshoot) and achieving a first-order lag response. As a result, stable control of the synchronous motor becomes possible.

[0070] [Second Embodiment] In the first embodiment, an example was described in which the d-axis magnetic flux command value Φd* was used as the second torque command value. In the second embodiment, an example was described in which the d-axis current command value id* was used as the second torque command value. The block diagrams of the synchronous motor control device 1 shown in Figure 1 and the control unit 100 shown in Figure 2 in the first embodiment are the same in the second embodiment.

[0071] Figure 12 is a block diagram of the difference calculation unit 204 in this embodiment. The difference calculation unit 204 comprises a first difference calculation unit 301, a second difference calculation unit 302, and a low-pass filter 303. The first difference calculation unit 301 and the low-pass filter 303 are the same as in the first embodiment. The second difference calculation unit 302 is different from that of the first embodiment.

[0072] The second difference calculation unit 302 takes the first torque T1, the first torque command value T1*, the d-axis current command value id*, the q-axis current command value iq*, and the filtered q-axis magnetic flux command value ΦqLPF* as inputs and calculates the second difference ΔT2. Details will be described later with reference to Figure 14.

[0073] Figure 13 shows the torque operating point on a coordinate system represented by the d-axis current and q-axis current. Figure 13 shows the transition of the operating point when the torque sign changes from positive to negative. The current torque, represented by the d-axis current id, is the operating point C, and the target torque, represented by the d-axis current command value id*, is the operating point D. Furthermore, the d-axis current id0 represents the d-axis current id when the q-axis current iq is 0.

[0074] As already mentioned, since the voltage amplitude is fixed during single-pulse control, the operating point moves along the magnetic flux limiting ellipse Φidq. The magnetic flux limiting ellipse Φidq is the magnetic flux limiting circle Φ (=Φlim) represented by coordinates expressed by the d-axis current and q-axis current, and can be expressed by the following equation (15) if the voltage drop across the winding resistance R of the synchronous motor 3 is sufficiently small compared to the speed electromotive force. Φidq = Φlim =sqrt((-Lq×iq)^2+(Ld×id+Φm)^2) ···(15) Here, Ld is the d-axis inductance, Lq is the q-axis inductance, and Φm is the magnetic flux coefficient of the magnet.

[0075] When the torque command value changes in a stepwise manner from positive to negative, the current operating point C moves along the magnetic flux limiting ellipse, via the d-axis current id0, to the operating point D indicated by the d-axis current command value id*. Considering this movement, if the torque sign changes from positive to negative or negative to positive (T1 × T1* < 0), the difference is corrected by calculating the second difference ΔT2 using equation (16). If the torque sign does not change (T1 × T1* >= 0), the second difference ΔT2 is calculated using equation (17). However, the q-axis current iq may be used instead of ΦqLPF, or the value obtained by applying a low-pass filter that simulates the current control response to the q-axis current iq may be used. ΔT2=(2id0-id*-id)×ΦqLPF* ···(16) ΔT2=(id*-id)×ΦqLPF* ···(17) Here, the d-axis current id0 can be expressed as equation (18) by substituting id=id0 and iq=0 into equation (15) and rearranging it. id0=(Φlim-Φm) / Ld ···(18)

[0076] In general, the synchronous motor 3 used in electric vehicles is greatly affected by magnetic saturation, and Φm and Ld change with the current. Therefore, in this embodiment, the d-axis current id0 is referenced in a lookup table based on Φlim calculated by equation (1).

[0077] Figure 14 is a block diagram of the second difference calculation unit 302 in the second embodiment. The second difference calculation unit 302 is configured to calculate the second difference ΔT2 based on equation (16) when T1 × T1 * < 0, and based on equation (17) when T1 × T1 * >= 0.

[0078] The lookup table 392 outputs the d-axis current id0 corresponding to the input magnetic flux limit value Φlim. The d-axis current id0 is input to the multiplier 312.

[0079] The multiplier 312 doubles the d-axis current id0, and the adder / subtractor 322 subtracts the d-axis current command value id* from this doubled d-axis current id0, outputting the d-axis corrected current command value id** to one side of the switch 332. The other side of the switch 332 receives the d-axis current command value id* as input.

[0080] The multiplier 342 multiplies the first torque T1 by the first torque command value T1*. The sign determiner 352 determines the result of this multiplication and outputs it to the switch 332. The switch 332, upon receiving the determination from the sign determiner 352, outputs the d-axis correction current command value id** if the sign of the torque command value has reversed. If the sign of the torque command value has not reversed, it outputs the d-axis current command value id*. In other words, during the period when the sign of the torque command value has reversed, the switch 332 outputs the d-axis correction current command value id** instead of the d-axis current command value id*.

[0081] The adder / subtractor 362 subtracts the d-axis current from the d-axis current command value id* or d-axis correction current command value id** output from the switch 332 to calculate the movement distance Δid of the operating point. This movement distance Δid is multiplied by the filtered q-axis magnetic flux command value ΦqLPF* in the multiplier 372 and output as the second difference ΔT2.

[0082] In other words, it is determined whether the sign of the torque command value has reversed relative to the current operating point of the synchronous motor 3. If the sign has changed, the second difference ΔT2 is calculated using equation (16); otherwise, it is calculated using equation (17). This makes it possible to achieve a first-order lag response even when the sign of the torque command value changes.

[0083] Figure 15 is a block diagram showing a modified example of the second difference calculation unit 302 in the second embodiment. In the second difference calculation unit 302 shown in Figure 14, only Φlim was used to reference the d-axis current id0. In the modified example shown in Figure 15, the voltage drop across the winding resistance R of the synchronous motor 3 is taken into consideration, and the d-axis current id0 is referenced using the magnetic flux limit value Φlim and the DC voltage Vdc. The same reference numerals are used for parts identical to those in Figure 14 for simplified explanation.

[0084] When the voltage drop across the motor winding resistance R is taken into account in equation (15), the following equation (19) is obtained. Φidq=Φlim=sqrt((R×id / ω-Lq×iq)^2 +(R×iq / ω+Ld×id+Φm)^2) ···(19) Substituting id=id0 and iq=0 into equation (19) yields equation (20). Φlim = sqrt((R × id0 / ω)^2 +(Ld×id0+Φm)^2) ···(20)

[0085] In equation (20), if the magnetic flux limit Φlim is constant, when the DC voltage Vdc is high, the rotational angular velocity ω is large, and the effect of the voltage drop in the first term on the right-hand side becomes small, so the d-axis current id0 is almost the same as in equation (18). Conversely, when the DC voltage Vdc is small, the rotational angular velocity ω becomes small, and the effect of the voltage drop becomes large.

[0086] Figure 16 shows an example of a lookup table 392' in the second embodiment. The x-axis represents the magnetic flux limit value Φlim, the y-axis represents the DC voltage Vdc, and the z-axis represents the d-axis current id0. As shown in Figure 16, when the DC voltage Vdc is low, the change in the d-axis current id0 is large with respect to the DC voltage Vdc, and when the DC voltage Vdc is high, the axis current id0 remains almost constant.

[0087] According to the second embodiment, when the sign of the torque command reverses from the current operating point of the synchronous motor 3, the difference between the torque of the synchronous motor 3 and the command value (d-axis current command value id*) can be correctly calculated, thereby suppressing overcurrent and undershoot (overshoot) and achieving a first-order lag response. As a result, stable control of the synchronous motor becomes possible.

[0088] [Reasons for using the second torque command value] The reason for using a second torque command value, which is gradually increasing in the region including the torque peak of the synchronous motor 3, as the torque command value T* is explained below. Figure 17 is a graph showing the relationship between the voltage phase angle θv and the torque output from the synchronous motor 3. The horizontal axis represents the voltage phase angle θv, which is set to 0 degrees with respect to the q-axis. The vertical axis represents torque.

[0089] Up to the maximum torque, the torque increases monotonically with respect to the voltage phase angle θv. However, beyond the maximum torque, the torque begins to decrease. Similarly, up to the minimum torque, the torque decreases monotonically with respect to the voltage phase angle θv. However, beyond the minimum torque, the torque begins to increase. Therefore, it is necessary to control the torque within a range where the voltage phase angle θv does not exceed the maximum or minimum torque, and it is necessary to limit the torque command value (or voltage phase angle θv) by providing a margin from the maximum or minimum torque. For this reason, when manipulating the voltage phase angle θv based on actual torque or estimated torque calculated from current values, the torque that can actually be used is limited compared to the torque that the synchronous motor 3 is originally capable of outputting.

[0090] To solve this, one can consider manipulating the voltage phase angle θv using a physical quantity that increases monotonically even near the maximum (minimum) torque, i.e., a second torque command value. For example, the d-axis magnetic flux command value Φd* or the d-axis current command value id* can be used.

[0091] Figure 18 is a graph showing the relationship between the voltage phase angle θv and the d-axis magnetic flux Φd. Figure 19 is a graph showing the relationship between the voltage phase angle θv and the d-axis current id. As shown in Figures 18 and 19, in the case of d-axis magnetic flux Φd and d-axis current id, the voltage phase angle θv increases monotonically from 0 degrees to 180 degrees (-180 degrees), making stable torque control possible even near the maximum (minimum) torque.

[0092] However, when controlling the voltage phase angle θv based on the d-axis magnetic flux Φd and d-axis current id, if the sign of the torque command reverses from the current operating point of the synchronous motor 3, the difference between the torque of the synchronous motor 3 and the command value is not calculated correctly, resulting in unstable control of the synchronous motor 3. In this embodiment, as already mentioned, when the sign of the torque command reverses from the current operating point of the synchronous motor 3, the difference between the torque of the synchronous motor 3 and the command value can be correctly calculated and corrected, preventing overshoot, achieving a first-order lag response as designed, and enabling stable control of the synchronous motor 3.

[0093] In the first and second embodiments, the d-axis magnetic flux command value Φd* and the d-axis current command value id* were described as examples of the second torque command value. However, the second torque command value may not be limited to the d-axis magnetic flux command value Φd* or the d-axis current command value id*, but may also be a value that gradually increases or decreases in the region including the maximum or minimum torque shown in the graph of Figure 17, or a combination of these values. Furthermore, in the first and second embodiments, an example was described in which the difference is corrected by calculating the second torque command value based on the magnetic flux limit value. However, other correction values ​​may be used instead of the magnetic flux limit value. Other correction values ​​may be parameters that represent the movement distance ΔΦd of the operating point as described with reference to Figure 4, or in some cases, constants corresponding to the characteristics of the synchronous motor 3. In short, any value that corrects the difference to a larger value when the sign of the torque command is reversed is sufficient.

[0094] [Third Embodiment] Figure 20 is a diagram showing the configuration of the electric vehicle 1000 in the third embodiment. The synchronous motor control device 1 shown in Figure 20 is the same synchronous motor control device 1 described in the first and second embodiments. The synchronous motor control device 1 converts DC power from the DC power supply 2 into AC power to drive the synchronous motor 3.

[0095] The synchronous motor 3 is connected to the transmission 601. The transmission 601 is connected to the drive shaft 603 via the differential gear 602 and supplies power to the wheels 604. Alternatively, the transmission 601 may be omitted and the motor may be directly connected to the differential gear 602, or a synchronous motor 3 and a synchronous motor control device 1 may be applied to the front wheels and rear wheels respectively.

[0096] In automobiles, torque at high speeds is related to acceleration performance, for example, on highways. In particular, on uphill roads, if torque at high speeds is insufficient, it becomes impossible to increase speed, so there is a strong demand for improved torque at high speeds. Also, from the perspective of improving energy efficiency, there is a demand to make the synchronous motor 3 smaller and lighter, and a synchronous motor control device 1 that can realize the maximum torque that the synchronous motor 3 can output is important. Similarly, in railways, which are also moving objects like automobiles, acceleration performance during high-speed running is equally important as it affects the travel time between stations. Furthermore, even if the sign of the torque command reverses from the current operating point of the synchronous motor 3, the difference between the torque of the synchronous motor 3 and the command value is correctly corrected to stabilize the control of the synchronous motor 3. In electric vehicle 1000, scenarios in which the sign of the torque command changes include vibration damping control and gear changes. By applying the synchronous motor control device 1 described in the first and second embodiments as the synchronous motor control device 1, it is possible to improve acceleration performance and control stability at high speeds in automobiles and railways.

[0097] In the first and second embodiments, the synchronous motor control device 1 was described with reference to a block diagram. Of these block diagrams, some or all of the blocks, excluding the power conversion unit 400, may be implemented by a processor (e.g., CPU, GPU) and a program executed by the processor. In this case, the processor may be the main processing unit, as the program is executed by the processor and performs defined processing using memory resources (e.g., memory) and / or interface devices (e.g., communication ports) as appropriate. Similarly, the main processing unit that executes the program may be a controller, device, system, computer, or node having a processor. The main processing unit that executes the program may be an arithmetic unit and may include a dedicated circuit (e.g., FPGA or ASIC) that performs specific processing.

[0098] A program may be installed from its program source into a device such as a computer. The program source may be, for example, a program distribution server or a computer-readable storage medium. If the program source is a program distribution server, the program distribution server includes a processor and storage resources for storing the program to be distributed, and the processor of the program distribution server may distribute the program to other computers. Furthermore, in the following description, two or more programs may be implemented as a single program, or one program may be implemented as two or more programs.

[0099] According to the embodiments described above, the following effects and advantages can be obtained. (1) The synchronous motor control device 1 includes a torque command value conversion unit 201 that converts a torque command value T* to the synchronous motor 3 into a second torque command value (d-axis magnetic flux command value Φd*, or d-axis current command value id*) that is gradually increasing in the region including the torque peak of the synchronous motor 3; a difference calculation unit 204 that calculates the difference (second difference ΔT2) between the torque of the synchronous motor 3 and the second torque command value; a voltage phase control unit 206 that controls the voltage phase angle θv so that the difference becomes small; and a power conversion unit 400 that converts DC power to AC power based on the voltage phase angle θv and outputs the converted AC power to the synchronous motor 3. The difference calculation unit 204 corrects the difference (second difference ΔT2) by calculating the second torque command value based on a predetermined correction value (magnetic flux limit value Φlim) when the sign of the torque command value T* reverses from the current operating point of the synchronous motor 3. As a result, overshoot does not occur even when the sign of the torque command changes, and a first-order lag response as designed can be achieved.

[0100] (2) The synchronous motor control method is a synchronous motor control method in a synchronous motor control device 1 equipped with a power conversion unit 400 that converts DC power to AC power based on a voltage phase angle θv and outputs the converted AC power to a synchronous motor 3. The method converts the torque command value T* to the synchronous motor 3 into a second torque command value (d-axis magnetic flux command value Φd*, or d-axis current command value id*) that is gradually increasing in the region including the torque peak of the synchronous motor 3, calculates the difference (second difference ΔT2) between the torque of the synchronous motor 3 and the second torque command value, controls the voltage phase angle θv so that the difference becomes smaller, and corrects the difference (second difference ΔT2) by calculating the second torque command value based on a predetermined correction value (magnetic flux limit value Φlim) when the sign of the torque command value reverses from the current operating point of the synchronous motor 3. As a result, overshoot does not occur even when the sign of the torque command changes, and a first-order lag response as designed can be achieved.

[0101] The present invention is not limited to the embodiments described above, and other forms conceivable within the scope of the technical concept of the present invention are also included within the scope of the present invention, as long as they do not impair the features of the present invention. Furthermore, configurations combining the embodiments and modifications described above are also possible. [Explanation of symbols]

[0102] 1... Synchronous motor control device, 2... DC power supply, 3... Synchronous motor, 31... Magnetic pole position sensor, 100... Control unit, 114-119... Switching elements, 120-125... Freewheeling diodes, 133... High-potential wiring, 134... Low-potential wiring, 105, 106, 107... Phase wiring, 200... Voltage detector, 201... Torque Command value conversion unit, 202... Three-phase to two-phase conversion unit, 203... Magnetic flux calculation unit, 204... Difference calculation unit, 205... Speed ​​calculation unit, 206... Voltage phase control unit, 207... Square wave generation unit, 208... PWM gate signal generation unit, 300... Gate drive circuit, 382, ​​392, 392'... Lookup table, 400... Power conversion unit, 500... • Current detector, 601...Transmission, 602...Differential gear, 603...Drive shaft, 604...Wheel, 1000...Electric vehicle, T*...Torque command value, ω...Rotational angular velocity, iu, iv, iw...Phase current, GS...Gate signal, Vdc...DC voltage, θv...Voltage phase angle, id*...d-axis current command value iq*...q-axis current command value, Φd*...d-axis magnetic flux command value, Φq*...q-axis magnetic flux command value, id...d-axis current, iq...q-axis current, Φd...d-axis magnetic flux, Φq...q-axis magnetic flux, Φlim...magnetic flux limit value, ΦqLPF*...filtered q-axis magnetic flux command value, ΔT1...first difference, ΔT2...second difference, Vu*, Vv*, Vw*...three-phase voltage command values.

Claims

1. A torque command value conversion unit converts a torque command value to a synchronous motor into a second torque command value that is gradually increasing in the region including the torque peak of the synchronous motor, A difference calculation unit that calculates the difference between the torque of the synchronous motor and the second torque command value, A voltage phase control unit controls the voltage phase angle so that the aforementioned difference becomes smaller, The system includes a power conversion unit that converts DC power to AC power based on the voltage phase angle and outputs the converted AC power to the synchronous motor, The difference calculation unit is a synchronous motor control device that corrects the difference by calculating the second torque command value based on a predetermined correction value when the sign of the torque command value is reversed from the current operating point of the synchronous motor.

2. In the synchronous motor control device according to claim 1, The torque command value conversion unit generates the d-axis magnetic flux command value and the q-axis magnetic flux command value as the second torque command value. The difference calculation unit calculates the difference using the d-axis magnetic flux and q-axis magnetic flux based on the current of the synchronous motor, and the d-axis magnetic flux command value and the q-axis magnetic flux command value. Furthermore, the difference calculation unit corrects the difference by calculating the d-axis magnetic flux command value based on the correction value when the sign of the torque command value is reversed from the current operating point of the synchronous motor.

3. In the synchronous motor control device according to claim 1, The torque command value conversion unit generates the d-axis current command value and the q-axis current command value as the second torque command value. The difference calculation unit calculates the difference based on the d-axis current and q-axis current based on the current of the synchronous motor, and the d-axis current command value and the q-axis current command value. Furthermore, the difference calculation unit corrects the difference by calculating the d-axis current command value based on the correction value when the sign of the torque command value is reversed from the current operating point of the synchronous motor.

4. In the synchronous motor control device according to claim 2 or claim 3, A synchronous motor control device in which the correction value is a magnetic flux limit value calculated based on the DC voltage supplied to the power conversion unit and the rotational angular velocity of the synchronous motor.

5. In the synchronous motor control device according to claim 1, The difference calculation unit determines, based on the sign of the torque command value, that the sign of the torque command value has reversed from the current operating point of the synchronous motor, by the sign of the result of multiplying the torque of the synchronous motor by the second torque command value.

6. A synchronous motor control device according to any one of claims 1 to 3, The system comprises a synchronous motor controlled by the synchronous motor control device, An electric vehicle powered by the aforementioned synchronous motor.

7. A synchronous motor control method in a synchronous motor control device, comprising a power conversion unit that converts DC power to AC power based on a voltage phase angle and outputs the converted AC power to a synchronous motor, wherein The torque command value to the synchronous motor is converted into a second torque command value that is gradually increasing in the region including the torque peak of the synchronous motor. The difference between the torque of the synchronous motor and the second torque command value is calculated. The voltage phase angle is controlled so that the difference becomes smaller. A synchronous motor control method that corrects the difference by calculating the second torque command value based on a predetermined correction value when the sign of the torque command value reverses from the current operating point of the synchronous motor.

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