Power conversion device and air-conditioning device

Abstract

A power conversion device (100) comprises: an inverter (3) that converts DC power into AC power and supplies the AC power to a motor (5); and a control unit (4) that generates a switching signal for the switching elements (31a to 31c, 32a to 32c) of the inverter (3). A control unit (4) performs two-phase modulation in which switching operations of switching elements (31a, 32a, 31b, 32b, or 31c, 32c) of one phase among three phases are sequentially stopped, and inserts a three-phase modulation period for performing a three-phase switching operation immediately before and / or immediately after the timing at which a phase in which a switching operation is stopped during two-phase modulation on the basis of a current flowing into and out of an inverter (3) shifts from a switching period to a switching-stopped period, and also immediately before and / or immediately after the timing at which a phase in which the switching operation is stopped shifts from a switching-stopped period to a switching period.

Description

Power conversion device and air conditioning device

[0001] The present disclosure relates to a power conversion device and an air conditioner that include an inverter that converts DC power into AC power and supplies the AC power to a three-phase load.

[0002] PWM (Pulse Width Modulation) drive is generally used as a drive method for each switching element provided in an inverter. Common modulation methods for sinusoidal modulation using PWM include "three-phase modulation" and "two-phase modulation." When using three-phase modulation and two-phase modulation together, three-phase modulation is basically used, and when switching loss needs to be reduced, it is often the case that three-phase modulation is switched to two-phase modulation.

[0003] An inverter has legs consisting of upper and lower elements (hereinafter referred to as "upper and lower elements") connected in series. The drive signals for driving the upper and lower elements of the inverter include a period called dead time during which the upper and lower elements are simultaneously turned off to prevent a leg short circuit that would occur if the upper and lower elements were simultaneously turned on. From the perspective of a three-phase load, the dead time corresponds to a disturbance voltage. For example, if the three-phase load is a motor, the dead time can cause ripples in the motor current and motor torque, which can adversely affect noise and vibration.

[0004] To prevent this adverse effect, Non-Patent Document 1 below discloses a technique for reducing ripples in the motor current and motor torque by superimposing a correction voltage on the voltage command for each phase based on the carrier frequency used to generate the DC bus voltage, motor current, and drive signal. This correction is called dead time correction.

[0005] Patent Document 1 below discloses a control method for an inverter circuit in a three-phase voltage-type inverter that obtains three-phase AC voltage from a DC power source. The method selects, depending on the rotation speed, either bottom-pinned two-phase modulation or top-pinned two-phase modulation, which reduces the maximum value of leakage current in the low-speed range, and top-pinned two-phase modulation, which ensures speed stability in the high-speed range. The bottom-pinned two-phase modulation is a modulation method in which the voltage amplitude command for each phase is set to its minimum value every 120 degrees, which is 1 / 3 of one electrical angle cycle, and the bottom element of the top and bottom elements of the inverter is maintained in an on-state for a period of 120 degrees. The top-pinned two-phase modulation is a modulation method in which the positive and negative polarities of the bottom-pinned two-phase modulation are reversed, i.e., the top element of the top and bottom elements of the inverter is maintained in an on-state for a period of 120 degrees. The top-pinned two-phase modulation is a method in which both bottom-pinned two-phase modulation and top-pinned two-phase modulation are alternately performed every 60 degrees within one electrical angle cycle.

[0006] Japanese Patent Application Laid-Open No. 2006-217673

[0007] Hidehiko Sugimoto and two others, "Theory and Design Practice of AC Servo Systems," Sogo Electronics Publishing, pp. 54-58

[0008] When three-phase modulation is performed, dead time correction is performed by superimposing a correction voltage on the voltage command for each phase. On the other hand, when two-phase modulation is performed, dead time correction is not performed during switching pause periods when switching operation is paused. Therefore, when three-phase modulation and two-phase modulation are used together, during the three-phase modulation periods immediately before and after the switching pause periods when two-phase modulation is performed, the voltage operation margin, which is the margin for voltage superposition, is small due to the influence of minimum pulse width constraints for protecting the switching elements or ensuring current detection function, making it difficult to perform the desired dead time correction. This has resulted in a problem of insufficient suppression of ripple in the motor current and motor torque.

[0009] The present disclosure has been made in view of the above, and aims to provide a power conversion device that can sufficiently suppress ripples in motor current and motor torque even when a drive method that combines three-phase modulation and two-phase modulation is adopted.

[0010] In order to solve the above-mentioned problems and achieve the object, a power conversion device according to the present disclosure includes an inverter that converts DC power into AC power and supplies the power to a three-phase load, and a control unit that generates switching signals for a plurality of three-phase switching elements included in the inverter and outputs the switching signals to the inverter. The control unit performs two-phase modulation to sequentially suspend switching operations of switching elements of one of the three phases, and inserts, based on current flowing in and out of the inverter, a three-phase modulation period in which switching operations are performed for all three phases at least one of immediately before and after a timing at which a phase whose switching operation is suspended transitions from a switching period to a switching suspend period during the two-phase modulation and at least one of immediately before and after a timing at which a phase whose switching operation is suspended transitions from a switching suspend period to a switching period.

[0011] According to the power conversion device of the present disclosure, even when a drive method that uses both three-phase modulation and two-phase modulation is adopted, it is possible to sufficiently suppress ripples in the motor current and motor torque.

[0012] FIG. 1 is a diagram illustrating a basic configuration and basic functions of a power conversion device according to the first embodiment; FIG. 2 is a diagram illustrating another example configuration having the basic functions of the power conversion device shown in FIG. 1; FIG. 3 is a block diagram illustrating a basic function relating to generation of a switching signal in a control unit according to the first embodiment; FIG. 4 is a diagram illustrating a problem in the prior art; FIG. 5 is a diagram illustrating a second three-phase voltage modulated wave generated inside the control unit according to the first embodiment; FIG. 6 is a diagram illustrating an example of a characteristic table referenced inside the control unit according to the first embodiment; FIG. 7 is a diagram illustrating a relationship between a u-phase Td correction value generated inside the control unit according to the first embodiment and a u-phase current; FIG. 1 is a block diagram of a control unit that realizes the modulation method selection control described above. FIG. 2 is a diagram showing an example of waveforms of phase current and q-axis current by conventional two-phase modulation. FIG. 3 is a diagram showing an example of waveforms of phase current and q-axis current when a power conversion device according to embodiment 1 is used. FIG. 4 is a diagram showing an example of waveforms of phase current and q-axis current when a power conversion device according to embodiment 1 is used. FIG. 5 is a diagram showing an example of waveforms of a three-phase modulation insertion period inserted by control according to embodiment 2. FIG. 6 is a diagram showing an example of waveforms of a three-phase modulation insertion period inserted by control according to embodiment 2. FIG. 7 is a diagram showing an example of waveforms of a three-phase modulation insertion period inserted by control according to embodiment 3.

[0013] A power conversion device and an air conditioning device according to embodiments of the present disclosure will be described in detail below with reference to the accompanying drawings.

[0014] First Embodiment Fig. 1 is a diagram illustrating the basic configuration and basic functions of a power conversion device 100 according to a first embodiment. In Fig. 1, the power conversion device 100 is connected between a commercial power supply 1 and a motor 5. The commercial power supply 1 is an example of an AC power supply. The motor 5 is a three-phase motor mounted on a three-phase load. When the power conversion device 100 is mounted on an air conditioning system, the three-phase load corresponds to a compressor that compresses a refrigerant, a fan that blows air to a heat exchanger that exchanges heat with the refrigerant, and the like.

[0015] The power conversion device 100 includes a converter 2, an inverter 3, and a control unit 4. The converter 2 rectifies the power supply voltage applied from the commercial power supply 1 and outputs the rectified voltage to the inverter 3. If the converter 2 has a boost function, the converter 2 boosts the power supply voltage and outputs the boosted voltage to the inverter 3. That is, the converter 2 rectifies the power supply voltage applied from the commercial power supply 1 and, if necessary, boosts the power supply voltage.

[0016] The inverter 3 is connected to the output terminal of the converter 2 by electrical wiring 6a and 6b. The electrical wiring 6a and 6b are also called DC buses. The electrical wiring 6a is a high-potential DC bus, and the electrical wiring 6b is a low-potential DC bus.

[0017] The inverter 3 has switching elements 31a, 31b, 31c, 32a, 32b, and 32c with freewheeling diodes connected in anti-parallel. In the inverter 3, the switching elements 31a to 31c are the upper elements described above, and the switching elements 32a to 32c are the lower elements described above. In the inverter 3, the switching elements 31a to 31c and 32a to 32c are turned on or off under the control of the control unit 4, and the inverter 3 converts the DC power output from the converter 2 into AC power having a desired amplitude and phase and supplies it to the motor 5. Note that while FIG. 1 illustrates a case where the switching elements 31a to 31c and 32a to 32c are IGBTs (insulated gate bipolar transistors), MOSFETs (metal oxide semiconductor field effect transistors) may be used instead of the IGBTs. In the case of a MOSFET, a parasitic diode is built in due to its structure, so a configuration in which the diode is not connected in antiparallel may be adopted.

[0018] The inverter 3 also has shunt resistors 33a, 33b, and 33c for detecting the current flowing through each phase of the inverter 3. The shunt resistor 33a is connected between the switching element 32a, which is the lower element, and the low-potential side electrical wiring 6b. The shunt resistors 33b and 33c are connected in a similar manner. The detected values ​​of the shunt resistors 33a to 33c are input to the control unit 4. The control unit 4 calculates the voltages detected by the shunt resistors 33a to 33c and converts them into currents to determine the current flowing through each phase of the inverter 3. Once the current flowing through each phase of the inverter 3 is determined, the three-phase output current output from the inverter 3 to the motor 5 can also be determined.

[0019] The control unit 4 also receives a detected value of the bus voltage Vdc from the converter 2. The bus voltage Vdc is the voltage between the electrical wiring 6a and 6b, which are DC buses. The bus voltage Vdc may be the voltage across a smoothing capacitor (not shown in FIG. 1 ) that smoothes the output voltage of the converter 2.

[0020] The control unit 4 generates switching signals for the three-phase switching elements 31a to 31c and 32a to 32c provided in the inverter 3 based on the current flowing through each phase of the inverter 3 and the bus voltage Vdc, and outputs the signals to the inverter 3. The on and off operations of the switching elements 31a to 31c and 32a to 32c are controlled by the switching signals.

[0021] Furthermore, in the power conversion device 100, the configuration and arrangement of each part shown in the basic configuration in Fig. 1 are one example, and the configuration and arrangement of each part are not limited to the example shown in Fig. 1. The power conversion device 100 according to the first embodiment may be configured, for example, as shown in Fig. 2. Fig. 2 is a diagram showing another example configuration having the basic functions of the power conversion device 100 shown in Fig. 1.

[0022] In Fig. 2, shunt resistors 33a to 33c are omitted, while shunt resistor 34 is inserted in electrical wiring 6b. While Fig. 1 shows a configuration known as a three-shunt system, Fig. 2 shows a configuration known as a one-shunt system. In the one-shunt system, the current flowing through each phase of inverter 3 is detected based on the timing at which switching elements 31a to 31c, 32a to 32c are turned on or off. Note that in the one-shunt system, the method of detecting the current flowing through each phase of inverter 3 based on the detection value of shunt resistor 34 is well known, and further description thereof will be omitted here.

[0023] Moreover, the power conversion device 100 according to the first embodiment may be configured, for example, as shown in Fig. 3. Fig. 3 is a diagram showing yet another example configuration having the basic functions of the power conversion device 100 shown in Fig. 1.

[0024] 3, the shunt resistors 33a to 33c are omitted, and current detectors 35a and 35b are inserted in the electrical wiring 7 connecting the inverter 3 and the motor 5. Each of the current detectors 35a and 35b detects the current for one phase of the three-phase output current that is the output current of the inverter 3. The detected values ​​of the current detectors 35a and 35b are input to the control unit 4. The control unit 4 calculates the current for the remaining one phase based on the detected values ​​of the currents for any two phases detected by the current detectors 35a and 35b.

[0025] Typical examples of the current detectors 35a and 35b include an ACCT (Alternating Current Transformer) that can detect only AC components and a DCCT (Direct Current Transformer) that can detect both DC and AC components, but any detector that can detect three-phase output current may be used.

[0026] Fig. 4 is a block diagram illustrating basic functions related to generation of a switching signal in the control unit 4 according to the first embodiment. The control unit 4 according to the first embodiment generates a switching signal by using a combination of three-phase modulation and two-phase modulation, and the control unit 4 has an internal functional block configuration as shown in Fig. 4 for implementing this control. Specifically, the control unit 4 includes a modulation method selection unit 41, a modulated wave generation unit 42, a Td correction unit 43, a PWM modulation unit 44, and a Td addition unit 45. Note that "Td" in the Td correction unit 43 and the Td addition unit 45 means dead time.

[0027] When a positive modulation factor command Vk and a voltage phase θ are given, the modulated wave generating unit 42 generates a first three-phase voltage modulated wave Vu1 expressed by the following equation (1): ＊ , Vv1 ＊ , Vw1 ＊ Generate.

[0028] Vu1 ＊ =Vk×cosθ Vv1 ＊ =Vk×cos(θ-2 / 3π) Vw1 ＊ =Vk×cos(θ−4 / 3π)…(1)

[0029] First three-phase voltage modulation wave Vu1 ＊ , Vv1 ＊ , Vw1 ＊ originally corresponds to a desired voltage to be output from the inverter 3, and is generated based on a voltage command output from a higher-level control system (not shown). The voltage phase θ is the phase of the three-phase output voltage, which is the output voltage of the inverter 3, and is the phase when the rotation of the motor 5 is viewed in terms of electrical angle.

[0030] The modulation method selection unit 41 selects and instructs a modulation method based on the modulation factor command Vk and the voltage phase θ. When three-phase modulation is instructed, the modulation wave generation unit 42 generates a first three-phase voltage modulation wave Vu1 ＊ , Vv1 ＊ , Vw1 ＊ is converted into a second three-phase voltage modulation wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ Furthermore, when two-phase modulation is instructed, the modulated wave generating unit 42 outputs a second three-phase voltage modulated wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ and outputs it to the Td correction unit 43.

[0031] Vu2 ＊ = Vu1 ＊ -Vcom Vv2 ＊ = Vv1 ＊ -Vcom Vw2 ＊ = Vw1 ＊ -Vcom...(2)

[0032] In the above equation (2), Vcom is a three-phase common signal. As shown in the above equation (2), the second three-phase voltage modulation wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ is the first three-phase voltage modulation wave Vu1 ＊ , Vv1 ＊ , Vw1 ＊ Since the line voltages between the phases are maintained, the three-phase common signal Vcom is generated by subtracting the three-phase common signal Vcom of the same value from the three-phase common signal Vcom. When the underlay two-phase modulation disclosed in Patent Document 1 is performed, the three-phase common signal Vcom can be calculated, for example, by the following equation (3):

[0033] Vcom=min(Vu1 ＊ , Vv1 ＊ , Vw1 ＊ ) + 1 … (3)

[0034] In the above formula (3), min(Vu1 ＊ , Vv1 ＊ , Vw1 ＊ ) is the first three-phase voltage modulation wave Vu1 ＊ , Vv1＊ , Vw1 ＊ This is a function that obtains the minimum value among

[0035] The Td correction unit 43 calculates a second three-phase voltage modulation wave Vu2 based on the three-phase output currents iu, iv, and iw, the bus voltage Vdc, and the carrier frequency fc. ＊ , Vv2 ＊ , Vw2 ＊ The third three-phase voltage modulation wave Vu3 obtained by correcting ＊ , Vv3 ＊ , Vw3 ＊ The Td correction unit 43 is equipped with a current characteristic when the dead time Td is regarded as a disturbance voltage. The Td correction unit 43 generates the second three-phase voltage modulation wave Vu2 by referring to the current characteristic according to the values ​​of the three-phase output currents iu, iv, and iw. ＊ , Vv2 ＊ , Vw2 ＊ Correct the following.

[0036] The PWM modulation unit 44 internally generates a carrier signal of the commanded carrier frequency fc and outputs the third three-phase voltage modulation wave Vu3 ＊ , Vv3 ＊ , Vw3 ＊ The Td adding unit 45 compares the carrier signal with the switching signal SW1 and generates a switching signal SW1 based on the magnitude relationship between the two signals. This switching signal SW1 is the signal before the dead time Td is added. The Td adding unit 45 generates a switching signal SW by adding the dead time Td to the switching signal SW1 and outputs the switching signal SW to the inverter 3.

[0037] Next, the problems of the prior art will be described. Fig. 5 is a diagram for explaining the problems of the prior art. The operation waveforms in Fig. 5 are those during two-phase modulation, and the solid line, dashed line, and thick dashed line respectively represent the u-phase voltage modulation wave Vu2 ＊ , v-phase voltage modulation wave Vv2 ＊ and w-phase voltage modulation wave Vw2 ＊ The thick solid line represents the u-phase current iu of the three-phase output currents iu, iv, and iw. The horizontal axis represents the electrical angle phase angle, and the vertical axis represents the voltage value of each modulated wave or the current value of the u-phase current iu. The voltage values ​​on the vertical axis are normalized to ±1, and the values ​​on the vertical axis correspond to the modulation factor. This also applies to Figures 6, 8, 9, and 15-19, which will be described later.

[0038] In the example of FIG. 5, when the electrical phase angle is in the range of 120 to 240 degrees, the u-phase voltage modulation wave Vu2 ＊ Since the first limiter value Limit1 has been reached, the switching operation of the u-phase is paused. In FIG. 5, this range of electrical phase angles is represented by "Yu." In this paper, the range of electrical phase angles in which the switching operation is paused when two-phase modulation is performed is referred to as the "switching pause period," and the period outside the switching pause period, i.e., the range of electrical phase angles in which the switching operation is not paused, is referred to as the "switching period." This switching period is the three-phase modulation period in which three-phase modulation is performed.

[0039] In Figure 5, attention is focused on periods Xa and Xb enclosed by thick dashed rectangular frames. Period Xa is a three-phase modulation period immediately preceding a u-phase switching pause period Yu during which two-phase modulation is performed. Period Xb is a three-phase modulation period immediately following a u-phase switching pause period Yu during which two-phase modulation is performed. As explained in the section [Problem to be Solved by the Invention], there is a minimum pulse width constraint when generating switching signal SW. The minimum pulse width is set to protect switching elements 31a to 31c and 32a to 32c and to ensure the current detection function.

[0040] In FIG. 5, the u-phase voltage modulation wave Vu2 ＊ The area surrounded by the rectangular frames in the periods Xa and Xb is hatched. The size of the hatched area represents the voltage manipulation margin for Td correction. The larger the area of ​​the hatched area, the larger the voltage manipulation margin. Although the voltage manipulation margin can be increased by extending the periods Xa and Xb, this approach does not solve the problem. This is because the closer to the u-phase switching pause period Yu, the greater the u-phase voltage modulation wave Vu2 ＊This is because the difference between the first limiter value Limit1 and the first limiter value Limit2 becomes small, and the voltage control margin rapidly decreases, making it difficult to perform sufficient Td correction. Therefore, the periods immediately before and after the u-phase switching pause period Yu are periods in which a voltage control error exists that prevents sufficient Td correction. On the other hand, during the u-phase switching pause period Yu, there is no switching operation and no Td correction is performed, so no voltage control error exists. Therefore, at the timing of the transition from period Xa to u-phase switching pause period Yu and from u-phase switching pause period Yu to period Xb, the voltage control error changes stepwise, causing current ripple and torque ripple.

[0041] Therefore, in the first embodiment, the second three-phase voltage modulation wave Vu2 is set to reduce the voltage control error near the boundary between the period Xa and the u-phase switching stop period Yu, and near the boundary between the u-phase switching stop period Yu and the period Xb. ＊ , Vv2 ＊ , Vw2 ＊ The specific calculation method is as follows:

[0042] First, the three-phase common signal Vcom is generated using the following equation (4) instead of the above equation (3).

[0043] Vcom=min(Vu1 ＊ , Vv1 ＊ , Vw1 ＊ ) +(1-Δduty)(Δduty>0)…(4)

[0044] In the above equation (4), Δduty is the amount of shift for shifting the first limiter value Limit1 in the voltage direction. ＊ , Vv2 ＊ , Vw2 ＊ is calculated by substituting the three-phase common signal Vcom obtained by the above equation (4) into the above equation (2). For convenience of explanation, the above equation (2) is rewritten as equation (5).

[0045] Vu2 ＊ = Vu1 ＊ -Vcom Vv2 ＊ = Vv1＊ -Vcom Vw2 ＊ = Vw1 ＊ -Vcom... (5) (reprinted)

[0046] FIG. 6 shows the second three-phase voltage modulation wave Vu2 generated inside the control unit 4 according to the first embodiment. ＊ , Vv2 ＊ , Vw2 ＊ 6 is a diagram illustrating the second three-phase voltage modulation wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ The waveform for one electrical angle cycle is shown. Because the shift amount Δduty is set to Δduty > 0, the lower limit value is shifted from "-1" (=Limit1) in the positive voltage direction by +Δduty. In this paper, the limiter value shifted in the voltage direction by the shift amount Δduty is denoted as "Limit2" and referred to as the "second limiter value." The relationship between the first limiter value Limit1, the second limiter value Limit2, and the shift amount Δduty is "Limit2 = Limit1 + Δduty." The appropriate range for Δduty will be described later.

[0047] In the first embodiment, the Td corrector 43 calculates the third three-phase voltage modulated wave Vu3 using the following equation (6): ＊ , Vv3 ＊ , Vw3 ＊ Generate.

[0048] Vu3 ＊ = Vu2 ＊ +Vtd_u Vv3 ＊ = Vv2 ＊ +Vtd_v Vw3 ＊ = Vw2 ＊ +Vtd_w…(6)

[0049] In the above equation (6), Vtd_u, Vtd_v, and Vtd_w are Td correction values ​​for the u-phase, v-phase, and w-phases. The u-phase Td correction value Vtd_u, the v-phase Td correction value Vtd_v, and the w-phase Td correction value Vtd_w can be calculated by referring to a characteristics table such as that shown in FIG. 7. FIG. 7 is a diagram showing an example of a characteristics table referred to within the control unit 4 according to the first embodiment. The horizontal axis of FIG. 7 represents the absolute values ​​of the instantaneous values ​​of the three-phase output currents iu, iv, and iw, and the vertical axis represents the absolute value |Vtd| of the Td correction value Vtd.

[0050] The Td correction unit 43 uses the absolute values ​​of the instantaneous values ​​of the three-phase output currents iu, iv, and iw as arguments to determine the absolute value |Vtd| of the Td correction value Vtd by referring to the characteristics table in Fig. 7. Furthermore, the Td correction unit 43 generates the Td correction values ​​Vtd_u, Vtd_v, and Vtd_w using the following equation (7).

[0051] Vtd_u=|Vtd|×sign(iu) Vtd_v=|Vtd|×sign(iv) Vtd_w=|Vtd|×sign(iw) …(7)

[0052] In the above equation (7), sign(iu) is a function that obtains the sign of the instantaneous value of the u-phase current iu, and takes on any of the values ​​"1", "0", and "-1". The same is true for sign(iv) and sign(iw). The three-phase output currents iu, iv, and iw can be obtained in any of the power conversion devices 100 shown in FIGS. 1 to 5.

[0053] 8 is a diagram showing the relationship between the u-phase Td correction value Vtd_u generated inside the control unit 4 according to the first embodiment and the u-phase current iu. In FIG. 8, the u-phase Td correction value Vtd_u is indicated by a solid line, and the u-phase current iu is indicated by a dashed line. As shown in FIG. 8, the direction of correction of the u-phase Td correction value Vtd_u is reversed depending on the sign of the instantaneous value of the u-phase current iu. The other v-phase Td correction values ​​Vtd_v and Vtd_w have a similar relationship.

[0054] Fig. 9 is a diagram illustrating the relationship between the first, second, and third three-phase voltage modulated waves generated inside the control unit 4 according to embodiment 1. In Fig. 9, the same waveforms and elements as those in Figs. 5 and 6 are denoted by the same reference numerals.

[0055] In FIG. 9, for simplicity, the first three-phase voltage modulation wave Vu1 ＊ , Vv1 ＊ , Vw1 ＊ and the third three-phase voltage modulation wave Vu3 ＊ , Vv3 ＊ , Vw3 ＊ Regarding the u-phase voltage modulation wave Vu1 ＊ , Vu3 ＊ Only the

[0056] 9 in comparison with Fig. 5, the periods Xa and Xb are changed to periods Xa' and Xb', respectively, in Fig. 9. Also, the u-phase switching pause period Yu is changed to a u-phase switching pause period Yu' in Fig. 9.

[0057] Comparing the u-phase switching pause period Yu and the u-phase switching pause period Yu', the u-phase switching pause period Yu' is shorter than the u-phase switching pause period Yu. This is because the period Xa shown in FIG. 5 includes only the period immediately before the timing at which the u-phase switching operation transitions from a switching period to a switching pause period, whereas the period Xa' shown in FIG. 9 also includes the period immediately after the timing at which the u-phase switching operation transitions from a switching period to a switching pause period. Similarly, the period Xb shown in FIG. 5 includes only the period immediately after the timing at which the u-phase switching operation transitions from a switching pause period to a switching period, whereas the period Xb' shown in FIG. 9 also includes the period immediately before the timing at which the u-phase switching operation transitions from a switching pause period to a switching period. As a result, the relationship Yu'<Yu holds between the u-phase switching pause period Yu and the u-phase switching pause period Yu'.

[0058] As shown in the above equation (6), the u-phase voltage modulation wave Vu3 ＊ , the u-phase voltage modulation wave Vu2＊ As shown in FIG. 7 , the u-phase Td correction value Vtd_u is a voltage corresponding to the absolute value of the instantaneous values ​​of the three-phase output currents iu, iv, and iw (in the case of the u-phase, the u-phase current iu). Therefore, the u-phase voltage modulation wave Vu2 ＊ is equal to or less than the second limiter value Limit2, which is the lower limit value. ＊ The waveform of the u-phase voltage modulation wave Vu3 is a waveform that gradually bottoms out as shown in FIG. 9. By appropriately setting the magnitude of the shift amount Δduty in accordance with the magnitude of the u-phase Td correction value Vtd_u, the u-phase voltage modulation wave Vu3 ＊ changes up and down across -1 at a gentle slope. ＊ The period during which the first limiter value Limit1 is less than the first limiter value Limit2 is the period during which no switching operation occurs in the u-phase.

[0059] Therefore, it is possible to appropriately perform Td correction in the periods Xa' and Xb' according to the first embodiment shown in Fig. 9. Furthermore, in the periods Xa' and Xb' according to the first embodiment, it is possible to reduce step-like changes in the voltage control error at the timing of transition from the period Xa to the u-phase switching suspension period Yu and from the u-phase switching suspension period Yu to the period Xb, thereby making it possible to suppress current ripple and torque ripple.

[0060] 9 describes the u-phase, but the same description can be given for the other v- and w-phases. The voltage and current waveforms of the u-, v-, and w-phases are symmetrical waveforms with a phase angle difference of 120 degrees, and it goes without saying that the same description can be given. The following description will also only be given for the u-phase.

[0061] As is clear from the above description, according to the power conversion device of embodiment 1, three-phase modulation periods in which all three phases are subjected to switching operation are inserted immediately before and after the timing at which the phase in which the switching operation is suspended during two-phase modulation transitions from the switching period to the switching suspension period. Also, three-phase modulation periods in which all three phases are subjected to switching operation are inserted immediately before and after the timing at which the phase in which the switching operation is suspended during two-phase modulation transitions from the switching suspension period to the switching period.

[0062] Next, an appropriate method for setting the shift amount Δduty will be described. First, in a control using both three-phase modulation and two-phase modulation, in order to reduce the change in voltage control error at the timing when the switching period and the switching pause period are switched to each other, the u-phase voltage modulation wave Vu3 after Td correction is set as shown in FIG. ＊ It is preferable that the u-phase voltage modulation wave Vu2 gradually falls below the first limiter value Limit1. ＊ , Vu3 ＊ Vu3 ＊ = Vu2 ＊ +Vtd_u, it is desirable to determine the shift amount Δduty in relation to the Td correction value Vtd_u. Therefore, when the maximum value of the absolute value |Vtd_u| of the Td correction value Vtd_u under the operating conditions is Vtd_peak, the shift amount Δduty is set to satisfy the following equation (8).

[0063] Vtd_peak×0.3<Δduty<Vtd_peak×0.7…(8)

[0064] 7, the absolute value |Vtd_u| and maximum value Vtd_peak of the Td correction value Vtd_u have characteristics that vary depending on the magnitude of the u-phase current iu. Therefore, setting the shift amount Δduty based on the above equation (8) is equivalent to setting it depending on the effective value of the motor current output from the inverter 3 to the motor 5. For this reason, the shift amount Δduty may be set as shown in the following equation (9).

[0065] Δduty=Ktd_I×(motor current effective value) (9) where Ktd_I: proportionality coefficient

[0066] Furthermore, when the motor 5 drives a fluid load such as a fan or a compressor, the load torque is approximately proportional to the square of the rotational speed of the motor 5. Therefore, the effective motor current value when driving such a fan or compressor increases as the rotational speed increases. Therefore, when the three-phase load is a fluid load such as a fan or a compressor, instead of calculating the shift amount Δduty as a motor current-dependent characteristic as in equation (9) above, it is possible to calculate it by converting the description to a rotational speed characteristic. A specific concept is shown in FIG. 10 . FIG. 10 is a diagram for explaining the method for setting the shift amount Δduty in embodiment 1.

[0067] The upper part of Fig. 10 shows the change characteristics of the motor current effective value in response to the rotation speed. Therefore, it is desirable to set the shift amount Δduty in accordance with the change characteristics of the motor current effective value, as shown in the lower part of Fig. 10. Generally, the rotation speed is less likely to change suddenly than the current, so setting the shift amount Δduty in response to the rotation speed has the advantage of enabling stable Td correction.

[0068] FIG. 11 is a diagram illustrating another method for setting the shift amount Δduty in the first embodiment. The change characteristics of the effective motor current shown in the upper part of FIG. 11 are the same as those shown in the upper part of FIG. 10. In FIG. 10, the shift amount Δduty is set in accordance with the change characteristics of the effective motor current. However, the characteristics for determining the set value may be switched depending on the range of the rotational speed. In the example shown in the lower part of FIG. 11, the range of the rotational speed is set as follows: a low speed range from zero to a first speed, a medium speed range from the first speed to a second speed, and a high speed range above the second speed, including the maximum rotational speed.

[0069] The low-speed range, where the rotation speed is low, is an operating condition frequently used when starting the inverter 3, and the voltage applied to the motor 5 is also low. For this reason, in this low-speed range, it is desirable to always use three-phase modulation in order to minimize the disturbance voltage that occurs when switching between two-phase modulation and three-phase modulation. Therefore, as shown in Figure 11, the shift amount Δduty is set to a large value, at least approximately 0.5. By setting the shift amount Δduty to such a value, it is possible to set it so that no switching pause period is inserted, regardless of what Td correction value Vtd is superimposed.

[0070] Furthermore, in the high-speed range where the rotation speed is high, the induced voltage of the motor is high and the influence of the disturbance voltage caused by the dead time Td is small. Therefore, the influence of the current ripple and torque ripple is small even with conventional two-phase modulation. It is also desirable to shorten the three-phase modulation period to reduce switching loss. Therefore, as shown in Figure 11, the shift amount Δduty is set to zero. Setting the shift amount Δduty to zero results in a control operation equivalent to conventional two-phase modulation, making it possible to improve the operating efficiency of the inverter 3 by reducing switching loss.

[0071] In addition, in the medium speed range where the rotation speed is medium, the shift amount Δduty is connected by an exponentially decreasing curve characteristic so that the change in the shift amount Δduty between the low speed range and the high speed range is smooth. Note that Fig. 11 is just an example, and any curve, including a straight line, may be used as long as the change in the shift amount Δduty is smooth.

[0072] FIG. 12 is a block diagram of the control unit 4 that realizes the modulation method selection control described with reference to FIGS. 10 and 11 . In FIG. 12 , components that are the same as or equivalent to those in FIG. 4 are designated by the same reference numerals. Comparing FIG. 12 with FIG. 4 , the modulation method selection unit 41 is replaced with a modulation method selection unit 41A in FIG. The modulation method selection unit 41A receives, in addition to the modulation factor command Vk and the voltage phase θ, the motor current effective value Irms or the rotation speed Rrot from a higher-level control system (not shown). In addition to the functions of the modulation method selection unit 41, the modulation method selection unit 41A also has the additional function of calculating a shift amount Δduty based on the motor current effective value Irms or the rotation speed Rrot. The modulation method selection unit 41A sets the shift amount Δduty in accordance with the change characteristics of the motor current effective value Irms, as described with reference to FIG. 10 , for example, and outputs the shift amount Δduty to the modulated wave generation unit 42. Alternatively, as described with reference to FIG. 11, the modulation method selection unit 41A determines the rotation speed region from the rotation speed Rrot, sets the shift amount Δduty according to the rotation speed region, and outputs the shift amount Δduty to the modulated wave generation unit 42. The modulated wave generation unit 42 generates the second three-phase voltage modulated wave Vu2 in accordance with an instruction from the modulation method selection unit 41A. ＊ , Vv2 ＊ , Vw2 ＊ Although not shown in FIG. 12 , the Td correction values ​​Vtd_u, Vtd_v, and Vtd_w for each phase calculated by the Td correction unit 43 may be reflected in the calculation of the shift amount Δduty performed by the modulation method selection unit 41A.

[0073] FIG. 13 shows example waveforms of phase currents and q-axis currents obtained by conventional two-phase modulation. FIG. 14 shows example waveforms of phase currents and q-axis currents obtained by using the power conversion device 100 according to embodiment 1. In each diagram, the upper row shows the phase current, and the lower row shows the q-axis current. The phase current is the current of any one of the three-phase output currents iu, iv, and iw. The q-axis current is a current component that contributes to motor torque when the phase current is transformed into a rotating rectangular coordinate system. The waveforms in FIGS. 13 and 14 are the result of driving the inverter 3 at a constant output frequency of 50 Hz electrical angle while intentionally setting the response of the current control system low in order to understand the influence of disturbance voltages due to dead time Td and the influence of switching periods and non-switching periods in two-phase modulation. The larger the torque ripple generated in the motor 5, the greater the pulsation of the q-axis current.

[0074] FIG. 13 shows the operating waveforms when the shift amount Δduty is set to 0. The phase current is distorted, and the q-axis current exhibits pulsation at a frequency three times the electrical angle of 50 Hz. On the other hand, FIG. 14 shows that the distortion of the phase current is suppressed, and the pulsation at three times the frequency is also suppressed. Therefore, by using the power conversion device 100 according to the first embodiment, current ripple can be suppressed. This suppresses torque ripple generated in the motor 5, and also suppresses noise caused by torque ripple. Furthermore, by using the power conversion device 100 according to the first embodiment, current ripple can be suppressed, thereby suppressing losses in the wiring resistance and winding resistance caused by the current ripple.

[0075] As described above, the power conversion device according to the first embodiment includes an inverter that converts DC power into AC power and supplies it to a three-phase load, and a control unit that generates switching signals for a plurality of three-phase switching elements included in the inverter and outputs the signals to the inverter. The control unit performs two-phase modulation to sequentially suspend the switching operation of the switching elements of one of the three phases, and inserts three-phase modulation periods in which all three phases are switched on, based on the current flowing in and out of the inverter, immediately before and after the timing at which the phase whose switching operation is suspended during the two-phase modulation transitions from a switching period to a switching suspend period and immediately before and after the timing at which the phase whose switching operation is suspended transitions from a switching suspend period to a switching period. This control reduces step-like changes in the voltage control error at the timings at which the switching period transitions from the switching period to the switching suspend period and from the switching suspend period to the switching period. This makes it possible to sufficiently suppress ripples in the motor current and motor torque, even when a drive method that combines three-phase modulation and two-phase modulation is adopted.

[0076] In the power conversion apparatus configured as described above, the control unit generates a first three-phase voltage modulated wave based on a voltage command output from a higher-level control system, and calculates a three-phase common signal for setting a switching pause period while maintaining a line voltage value for the generated first three-phase voltage modulated wave. The control unit also calculates a second three-phase voltage modulated wave by superimposing the three-phase common signal on the first three-phase voltage modulated wave. The control unit further generates a switching signal based on a third three-phase voltage modulated wave obtained by correcting an error caused by a dead time imparted to the switching signal for the second three-phase voltage modulated wave. The three-phase common signal can be calculated based on a first difference, which is the difference between a first three-phase voltage modulated wave of a value close to a predetermined first limiter value among the first three-phase voltage modulated waves and the first limiter value. When performing the process of inserting the three-phase modulation period, the three-phase common signal can be calculated based on a second difference which is the difference between a first three-phase voltage modulation wave whose value is close to the first limiter value and a second limiter value whose absolute value is smaller than the first limiter value. When the difference between the first limiter value and the second limiter value is set to a third difference, this third difference can be determined based on at least one of an effective value of a three-phase output current which is the output current of the inverter, or a voltage-current phase difference which is the phase difference between a three-phase output voltage which is the output voltage of the inverter and the three-phase output current.

[0077] When the three-phase load is a motor, the control unit may perform three-phase modulation without performing two-phase modulation under operating conditions where the rotational speed of the motor is below a predetermined threshold. On the other hand, the control unit may not insert a three-phase modulation period under operating conditions where the rotational speed of the motor exceeds the predetermined threshold. Furthermore, under operating conditions where the rotational speed of the motor is below the predetermined threshold, the control unit may determine the value of the third difference based on the value of a disturbance voltage caused by dead time, thereby setting a three-phase modulation period in which each phase is switched for the entire period. Furthermore, under operating conditions where the rotational speed of the motor exceeds the predetermined threshold, the control unit may not insert a three-phase modulation period for the entire period by setting the third difference to zero.

[0078] Second Embodiment The magnitude of the disturbance voltage caused by the dead time Td and the Td correction value Vtd, which is the correction voltage value thereof, are generated according to the magnitude of the instantaneous values ​​of the three-phase output currents iu, iv, and iw, as described above. In addition, the positive and negative polarities of the disturbance voltage and the Td correction value Vtd depend on the polarities of the three-phase output currents iu, iv, and iw, and therefore the remaining margin of the modulation wave operation width also depends on the magnitude and polarity of the three-phase output currents iu, iv, and iw. Therefore, the second three-phase voltage modulation wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ Near the first limiter value Limit1, which is the lower limit of the first limiter value Limit2, the instantaneous values ​​of the three-phase output currents iu, iv, and iw are large, and the second three-phase voltage modulation wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ When the polarity of the Td correction value Vtd, which is the correction voltage value, is the same as the polarity of the Td correction value Vtd, the remaining margin of the modulation wave operation width becomes small, and the risk of residual disturbance voltage due to insufficient Td correction increases.

[0079] On the other hand, to maintain the switching loss suppression effect, which is an inherent benefit of two-phase modulation, it is desirable to make the two-phase modulation period as long as possible. In consideration of these risks and benefits, in the second embodiment, two-phase modulation and three-phase modulation are switched multiple times within a period of 0 to 360 electrical degrees. In this way, by switching from two-phase modulation to three-phase modulation during a switching period immediately before or after a switching pause period of two-phase modulation, a voltage manipulation margin for the three-phase voltage modulation wave is provided.

[0080] Specifically, the control unit 4 switches between two-phase modulation and three-phase modulation multiple times within a period of 0 to 360 electrical degrees based on at least one of the voltage-current phase difference, the current values ​​of the three-phase output currents iu, iv, and iw, the Td correction value Vtd, and the modulation factor. When switching between two-phase modulation and three-phase modulation, a three-phase modulation period is inserted immediately before or after the switching timing, depending on at least one of the current-voltage phase difference, the current values ​​of the three-phase output currents iu, iv, and iw, and the Td correction value Vtd.

[0081] In FIG. 5, even during the switching period, the second three-phase voltage modulation wave Vu2＊ , Vv2 ＊ , Vw2 ＊ In order to ensure that the Td correction functions even during periods Xa and Xb close to the first limiter value Limit1, which is the lower limit value of the Td correction value Vtd, the second three-phase voltage modulation wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ However, in order to maintain the line voltage of the inverter output, the second three-phase voltage modulation wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ The second three-phase voltage modulation wave Vu2 of all phases, not just the phase close to the first limiter value Limit1, ＊ , Vv2 ＊ , Vw2 ＊ By performing such a shift operation, the second three-phase voltage modulation wave Vu2 of all phases can be shifted upward. ＊ , Vv2 ＊ , Vw2 ＊ None of these will fall below the first limiter value Limit1, and there will be no "bottom-stuck" phases. This is equivalent to inserting a period of three-phase modulation in which all three phases perform switching operations. In order to maintain the effect of suppressing switching loss, which is the original benefit of two-phase modulation, it is desirable to make the period in which three-phase modulation is inserted as short as possible. For this reason, in periods Xa and Xb, the second three-phase voltage modulation wave Vu2 ＊ , Vv2 ＊ , Vw2 ＊ When the phase angle condition is reached where there is sufficient margin for voltage manipulation, it is desirable to immediately return to two-phase modulation.

[0082] As described above, with the aim of ensuring a minimum modulation wave operation width and inserting a minimum three-phase modulation period, the three-phase modulation insertion period X3in is set based on at least one of the voltage-current phase difference, the current values ​​of the three-phase output currents iu, iv, and iw, the Td correction value Vtd, and the modulation factor, and during the three-phase modulation insertion period X3in, the three-phase common signal Vcom is given as shown in the following equation (10).

[0083] (Three-phase modulation insertion period) Vcom = min (Vu1＊ , Vv1 ＊ , Vw1 ＊ ) + (1 - Δduty) (other than the three-phase modulation insertion period) Vcom = min (Vu1 ＊ , Vv1 ＊ , Vw1 ＊ ) + 1 … (10)

[0084] 15 and 16 are first and second diagrams illustrating the three-phase modulation insertion period X3in inserted by the control according to embodiment 2. In Fig. 15 and Fig. 16, the same waveforms and elements as those in Fig. 9 are denoted by the same reference numerals.

[0085] 15 and 16, for the sake of simplicity, the first three-phase voltage modulation wave Vu1 ＊ , Vv1 ＊ , Vw1 ＊ and for the three-phase output currents iu, iv, and iw, the u-phase voltage modulation wave Vu1 ＊ 15 shows only the u-phase current iu and the u-phase voltage modulation wave Vu1. ＊ 16 shows an example in which the u-phase current iu is advanced in phase with respect to the u-phase voltage modulation wave Vu1 ＊ 15 and 16, the three-phase modulation insertion period X3in is indicated by a rectangular frame enclosed by a thick solid line.

[0086] In the first embodiment, periods Xa' and Xb' are provided before and after the u-phase switching pause period Yu in conventional two-phase modulation so as to provide a suitable margin for voltage manipulation. Although not specifically mentioned in the explanation of FIG. 9, the periods Xa' and Xb' shown in FIG. 9 basically have the same phase angle width. On the other hand, in the example of the second embodiment shown in FIG. 15, the periods Xa'' and Xb'' shown in the thick dashed rectangular frame do not necessarily satisfy Xa''=Xb''. As shown in FIG. 15, when the u-phase current iu is equal to the u-phase voltage modulation wave Vu1 ＊If the phase is advanced relative to the u-phase current iu, the polarity of the u-phase Td correction value Vtd_u is reversed at the zero-crossing timing of the u-phase current iu, eliminating the need to provide a voltage manipulation margin after the zero-crossing. Therefore, three-phase modulation may be switched to two-phase modulation at that timing. In the example of FIG. 15 , because three-phase modulation is switched to two-phase modulation at the zero-crossing timing, the relationship between period Xa″ and period Xb″ is Xa″>Xb″.

[0087] On the other hand, as shown in FIG. 16, the u-phase current iu is modulated by the u-phase voltage modulation wave Vu1 ＊ When the phase is delayed with respect to the u-phase current iu, the polarity of the u-phase Td correction value Vtd_u is reversed at the timing of the zero crossing of the u-phase current iu. ＊ The polarity of the u-phase Td correction value Vtd_u is the same as the polarity of the u-phase Td correction value Vtd_u, reducing the voltage manipulation margin. For this reason, it is desirable to switch from two-phase modulation to three-phase modulation at the zero-crossing timing of the u-phase current iu. In the example of FIG. 16 where two-phase modulation is switched to three-phase modulation at the zero-crossing timing, the relationship between the period Xa''' and the period Xb''' is Xa''' < Xb'''.

[0088] Furthermore, the widths of the periods Xa", Xb" and Xa'", Xb'", which are the width of the three-phase modulation insertion period X3in, may be changed according to the Td correction value Vtd in addition to the voltage-current phase difference described above. The Td correction value Vtd depends on the magnitude of the current values ​​of the three-phase output currents iu, iv, iw. When the current values ​​of the three-phase output currents iu, iv, iw are small, the operating margin of the modulated wave may be small. Therefore, the widths of the periods Xa", Xb", Xa'", Xb'" can be shortened accordingly.

[0089] Furthermore, the voltage manipulation margin increases as the modulation factor decreases. Therefore, the widths of the periods Xa", Xb", Xa'", and Xb'" can be shortened accordingly. Furthermore, when switching between two-phase modulation and three-phase modulation, the timing of switching may be controlled based on the widths of the periods Xa", Xb", Xa'", and Xb'", i.e., the width of the three-phase modulation insertion period X3in.

[0090] As described above, in the power conversion device according to the second embodiment, the control unit controls the timing and length of the three-phase modulation period based on at least one of the current value of the three-phase output current of the inverter, the phase difference between the three-phase output voltage and the three-phase output current of the inverter, the voltage correction value for correcting an error due to dead time, and the modulation factor of the three-phase output voltage. This control makes it possible to eliminate excess voltage manipulation margin and set the truly necessary width of the three-phase modulation insertion period. This makes it possible to minimize the period in which three-phase modulation is inserted, thereby maintaining the effect of suppressing switching loss, which is an inherent benefit of two-phase modulation.

[0091] Embodiment 3. In embodiments 1 and 2, three-phase modulation periods in which all three phases are switched are inserted immediately before and after the timing when the phase in which switching operation is paused transitions from a switching period to a switching pause period during two-phase modulation, and immediately before and after the timing when the phase in which switching operation is paused transitions from a switching pause period to a switching period. On the other hand, for example, under operating conditions or applications with small current amplitude, the influence of disturbance voltages due to dead time Td is small, so even if a period with a small voltage control margin remains, the generation of current ripple is small. Therefore, the width of the three-phase modulation insertion period X3in can be further narrowed compared to embodiments 1 and 2. In embodiment 3, this implementation will be described with reference to FIGS. 17 and 18 .

[0092] 17 and 18 are first and second diagrams illustrating the three-phase modulation insertion period X3in inserted by the control according to embodiment 3. In Fig. 17 and Fig. 18, the same waveforms and elements as those in Fig. 15 and Fig. 16 are denoted by the same reference numerals.

[0093] 17 and 18, in order to avoid complexity, the first three-phase voltage modulation wave Vu1 ＊ , Vv1 ＊ , Vw1 ＊ and for the three-phase output currents iu, iv, and iw, the u-phase voltage modulation wave Vu1 ＊ 17 shows only the u-phase current iu and the u-phase voltage modulation wave Vu1. ＊18 shows an example in which the u-phase current iu is advanced in phase with respect to the u-phase voltage modulation wave Vu1 ＊ 17 and 18, the three-phase modulation insertion period X3in is indicated by a rectangular frame enclosed by a thick solid line.

[0094] In the example of Fig. 17, the three-phase modulation insertion period X3in is set only immediately before the timing at which the phase in which switching operation is suspended transitions from a switching period to a switching suspension period and only immediately before the timing at which the phase in which switching operation is suspended transitions from a switching suspension period to a switching period. Also, in the example of Fig. 18, the three-phase modulation insertion period X3in is set only immediately after the timing at which the phase in which switching operation is suspended transitions from a switching period to a switching suspension period and only immediately after the timing at which the phase in which switching operation is suspended transitions from a switching suspension period to a switching period. That is, when the three-phase output currents iu, iv, and iw are advanced in phase with respect to the three-phase output voltage, the three-phase modulation insertion period X3in is set only immediately before the timing at which the switching period transitions between the switching period and the switching suspension period. When the three-phase output currents iu, iv, and iw are delayed in phase with respect to the three-phase output voltage, the three-phase modulation insertion period X3in is set only immediately after the timing at which the switching period transitions between the switching period and the switching suspension period.

[0095] In the example of FIG. 17 , the three-phase modulation insertion period X3in is inserted only immediately before the timing of the transition between the switching period and the switching pause period, but this does not prevent the three-phase modulation insertion period X3in from being set immediately after the transition. In the example of FIG. 18 , the three-phase modulation insertion period X3in is set only immediately after the timing of the transition between the switching period and the switching pause period, but this does not prevent the three-phase modulation insertion period X3in from being set immediately before the transition. According to the control of the third embodiment, the three-phase modulation insertion period can be limited, thereby suppressing switching loss while suppressing current ripple and torque ripple. That is, by using the power conversion device according to the third embodiment, it is possible to suppress both current ripple and torque ripple and switching loss.

[0096] As described above, in the power conversion device according to the third embodiment, the control unit performs two-phase modulation, which sequentially pauses the switching operation of the switching elements of one of the three phases, and inserts a three-phase modulation period, in which all three phases are switched on, at least one of immediately before and after the timing at which the phase whose switching operation is paused during the two-phase modulation transitions from a switching period to a switching pause period, and at least one of immediately before and after the timing at which the phase whose switching operation is paused transitions from a switching pause period to a switching period. This control limits the insertion period of the three-phase modulation. This makes it possible to simultaneously suppress current ripple and torque ripple and switching loss.

[0097] Fourth Embodiment In order to maintain the line voltage of the three-phase inverter output, it is necessary to shift the modulation waves of all phases in the same direction, and this shift operation causes fluctuations in the neutral point potential of the motor 5. Fluctuations in the neutral point potential have the adverse effect of accelerating the progression of motor shaft electrolytic corrosion, so fluctuations in the neutral point potential are undesirable. Therefore, in the fourth embodiment, fluctuations in the neutral point potential are suppressed while ensuring the accuracy of the Td correction. The control according to the fourth embodiment will be described below with reference to FIG. 19. FIG. 19 is a diagram illustrating the three-phase modulation insertion period X3in inserted by the control according to the fourth embodiment.

[0098] To avoid complexity, FIG. 19 shows only the u-phase current iu of the three-phase output current. Also, in FIG. 19 , the three-phase modulation insertion period X3in is indicated by a thick solid rectangular frame. In FIG. 19 , to provide symmetry to the three-phase modulation insertion period X3in, the three-phase modulation insertion period X3in is set with an equal phase angle width Δθ, i.e., a phase angle width of 2Δθ, before and after the phase angle of 120 degrees at which the u-phase switching pause period Yu in conventional two-phase modulation begins. Similarly, the three-phase modulation insertion period X3in is set with a predetermined equal phase angle width Δθ, i.e., a phase angle width of 2Δθ, before and after the phase angle of 240 degrees at which the u-phase switching pause period Yu in conventional two-phase modulation ends. While FIG. 19 illustrates an example of insertion for the u-phase switching pause period Yu, similar insertion is also performed for the v- and w-phase switching pause periods. Therefore, for each of the uvw phases, equal three-phase modulation periods in which all three phases perform switching operations are inserted immediately before and after the timing at which the phase in which switching operation is paused transitions from a switching period to a switching pause period, and immediately before and after the timing at which the phase in which switching operation is paused transitions from a switching pause period to a switching period. According to the control of embodiment 4, the phase angle widths of the three-phase modulation periods inserted immediately before and after the timing at which the switching pause period starts and immediately before and after the timing at which the switching pause period ends are equal for all four, making it possible to suppress fluctuations in the neutral point potential while ensuring the accuracy of the Td correction.

[0099] As described above, according to the power conversion device of the fourth embodiment, three-phase modulation periods of equal phase angle width are inserted at four locations: immediately before and after the start of a switching pause period, and immediately before and after the end of a switching pause period. This makes it possible to suppress fluctuations in the neutral point potential while ensuring the accuracy of the Td correction. This makes it possible to suppress the progression of motor shaft electrolytic corrosion while ensuring the accuracy of the Td correction.

[0100] Embodiment 5. The Td correction value Vtd described in Embodiment 1 is a voltage corresponding to the magnitude and polarity of the three-phase output current, as shown in Figures 7 and 8. Therefore, the closer the phase of the three-phase voltage modulated wave and the phase of the Td correction value Vtd, which is synonymous with the phase of the three-phase output current, the more the symmetry of the three-phase voltage modulated wave is maintained. In order to more effectively suppress fluctuations in the neutral point potential, a voltage command is generated so that the load power factor for the three-phase output current is 1.

[0101] As a method for controlling the load power factor to 1, for example, the method described in Japanese Patent Laid-Open Publication No. 10-243700 can be used. Also, if the motor 5, which is a three-phase load, is, for example, a surface permanent magnet motor, a method based on well-known vector control can be used. Specifically, the three-phase output currents iu, iv, and iw detected by the method of FIG. 1 are converted into d- and q-axis currents Id and Iq on an orthogonal coordinate system that rotates in synchronization with the rotor of the motor 5, and a d-axis current command Id is calculated. ＊ and q-axis current command Iq ＊ Furthermore, the dq axis current command Id ＊ , Iq ＊ The voltage command for each axis is generated based on the difference between the d-axis current command Id and the coordinate-transformed detected current values ​​Id and Iq. ＊ By setting zero and controlling the d-axis current Id to zero, the load power factor can be more accurately controlled to 1. Therefore, if the motor 5 is a surface permanent magnet motor, the load power factor can be suitably controlled to 1 by using this type of vector control.

[0102] As described above, in the power conversion device according to the fifth embodiment, the control unit controls the load power factor to approach 1 in accordance with the three-phase output current, which is the output current of the inverter. By controlling the load power factor to approach 1, the phase difference between each phase in the three-phase output current and the three-phase output voltage approaches zero, so that the switching pause period due to two-phase modulation coincides with the period in which each phase current is large. Since the switching loss reduction effect is approximately proportional to the magnitude of the three-phase output current, it is possible to enhance the switching loss reduction effect in addition to the effects described in the first to fourth embodiments. Furthermore, when the three-phase load is a surface permanent magnet motor, the load power factor can be controlled strictly to 1 by controlling the d-axis current command to zero, thereby further enhancing the switching loss reduction effect.

[0103] Sixth Embodiment Figure 20 is a diagram showing an example of the configuration of an air conditioning apparatus 200 pertaining to a sixth embodiment. The air conditioning apparatus 200 pertaining to the sixth embodiment includes the power conversion apparatus 100 described in the first to fifth embodiments, a compressor 50, a fan motor 5b, a fan 52 driven by the fan motor 5b, and a refrigeration cycle 110. The compressor 50 includes a compressor motor 5a and a compression element 51 that compresses the refrigerant. The compressor motor 5a is the drive source of the compressor 50.

[0104] The power conversion device 100 has two inverters (not shown) that supply power to a compressor motor 5a that is the drive source for the compressor 50 and a fan motor 5b that is the drive source for the fan 52. At least one of the two inverters is the inverter 3 described in any of the first to fifth embodiments. The converter 2 described in any of the first to fifth embodiments is intended to output a rectified voltage, and may be common to the two inverters 3 or may be provided individually for each of the two inverters 3.

[0105] In the refrigeration cycle 110, a refrigerant circuit is formed by a compressor 50, a four-way valve 121, a heat source-side heat exchanger 122, a load-side heat exchanger 132, and an expansion device 131. The compressor 50 compresses the refrigerant, the heat source-side heat exchanger 122 and the load-side heat exchanger 132 exchange heat between the refrigerant, and a fan 52 blows air to the heat source-side heat exchanger 122. Regarding the components of the refrigeration cycle 110, FIG. 20 shows a configuration in which the four-way valve 121 and the heat source-side heat exchanger 122 are provided in the outdoor unit 120, and the expansion device 131 and the load-side heat exchanger 132 are provided in the indoor unit 130. Note that the configuration in FIG. 20 is an example, and the air conditioning apparatus 200 according to embodiment 6 is not limited to the configuration in FIG. 20 .

[0106] If any of the power conversion devices 100 according to embodiments 1 to 5 is applied to the air conditioning device 200 according to embodiment 6, it is possible to enjoy any of the effects described in embodiments 1 to 5. Specifically, the current ripple of at least one of the compressor motor 5a, which is the drive source of the compressor 50, and the fan motor 5b, which is the drive source of the fan 52, is suppressed, and therefore the torque ripple generated by the motor is suppressed. This makes it possible to suppress noise of the air conditioning device 200 caused by torque ripple and also suppress losses caused by current ripple, making it possible to improve the performance of the air conditioning device 200.

[0107] Finally, the hardware configuration for realizing the functions of the control unit 4 described above will be described with reference to the drawings in Fig. 21 and Fig. 22. Fig. 21 is a diagram showing an example of a hardware configuration for realizing the functions of the control unit 4 in embodiments 1 to 5. Fig. 22 is a diagram showing another example of a hardware configuration for realizing the functions of the control unit 4 in embodiments 1 to 5.

[0108] When realizing some or all of the functions of the control unit 4 in embodiments 1 to 5, the configuration can include a processor 300 that performs calculations, a memory 302 that stores programs read by the processor 300, and an interface 304 that inputs and outputs signals, as shown in Figure 21.

[0109] The processor 300 is an example of a computing unit. The processor 300 may be a computing unit called a microprocessor, a microcomputer, a central processing unit (CPU), or a digital signal processor (DSP). Examples of the memory 302 include non-volatile or volatile semiconductor memory such as random access memory (RAM), read-only memory (ROM), flash memory, erasable programmable read-only memory (EPROM), and electrically programmable read-only memory (EEPROM), as well as a magnetic disk, a flexible disk, an optical disk, a compact disk, a minidisk, and a digital versatile disk (DVD).

[0110] The memory 302 stores a program that executes the functions of the control unit 4 in the first to fifth embodiments. The processor 300 exchanges necessary information via the interface 304, executes the program stored in the memory 302, and refers to the table stored in the memory 302, thereby performing the above-mentioned processing. The calculation results by the processor 300 can be stored in the memory 302.

[0111] 22 can be used to realize part of the functions of the control unit 4 in the first to fifth embodiments. The processing circuit 303 may be a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. Information input to and output from the processing circuit 303 can be exchanged via an interface 304.

[0112] It is also possible that some of the processing in the control unit 4 is performed by the processing circuit 303 , and the processing that is not performed by the processing circuit 303 is performed by the processor 300 and the memory 302 .

[0113] The configurations shown in the above embodiments are merely examples, and may be combined with other known technologies, or different embodiments may be combined with each other. It is also possible to omit or modify parts of the configurations as long as they do not deviate from the gist of the invention.

[0114] 1 to 3, for example, the number of phases of the commercial power supply 1, which is an AC power supply, may be either single-phase or three-phase. The commercial power supply 1 and converter 2 operate as DC power supply sources that supply DC power to the inverter 3, but a DC power supply such as a battery may be used instead of the commercial power supply 1 and converter 2. Also, in FIG. 2, the modulation factor command Vk and voltage phase θ input to the control unit 4 are generated based on current feedback control, including well-known vector control, or feedforward control in a higher-level control system. Furthermore, although the first to fifth embodiments have been described assuming a bottom-pinned two-phase modulation scheme, a top-pinned two-phase modulation scheme in which the upper limit value is set to "1" and pinned to the upper limit value side may also be used.

[0115] 1 Commercial power supply, 2 Converter, 3 Inverter, 4 Control unit, 5 Motor, 5a Compressor motor, 5b Fan motor, 6a, 6b, 7 Electrical wiring, 31a to 31c, 32a to 32c Switching elements, 33a to 33c, 34 Shunt resistor, 35a, 35b Current detector, 41, 41A Modulation method selection unit, 42 Modulation wave generation unit, 43 Td correction unit, 44 PWM modulation unit, 45 Td addition unit, 50 Compressor, 51 Compression element, 52 Fan, 100 Power conversion device, 110 Refrigeration cycle, 120 Outdoor unit, 121 Four-way valve, 122 Heat source side heat exchanger, 130 Indoor unit, 131 Expansion device, 132 Load side heat exchanger, 200 Air conditioning device, 300 Processor, 302 Memory, 303 Processing circuit, 304 interface.

Claims

1. an inverter that converts DC power into AC power and supplies it to a three-phase load; a control unit that generates switching signals for a plurality of three-phase switching elements provided in the inverter and outputs the signals to the inverter; Equipped with The control unit performs two-phase modulation to sequentially pause the switching operation of the switching elements of one of the three phases, and inserts a three-phase modulation period in which all three phases are subjected to switching operation at least one of immediately before and immediately after the timing at which the phase in which the switching operation is paused during the two-phase modulation transitions from a switching period to a switching pause period, and at least one of immediately before and immediately after the timing at which the phase in which the switching operation is paused transitions from a switching pause period to a switching period, based on the current flowing in and out of the inverter. Power conversion device.

2. When a three-phase output current, which is an output current of the inverter, is advanced in phase with respect to a three-phase output voltage, which is an output voltage of the inverter, the control unit inserts the three-phase modulation period immediately after a timing at which a phase in which a switching operation is paused transitions from a switching period to a switching pause period, and immediately after a timing at which a phase in which a switching operation is paused transitions from a switching pause period to a switching period. The power conversion device according to claim 1 .

3. When a three-phase output current, which is an output current of the inverter, is delayed in phase with respect to a three-phase output voltage, which is an output voltage of the inverter, the control unit inserts the three-phase modulation period immediately before a timing at which a phase for which a switching operation is paused transitions from a switching period to a switching pause period and immediately before a timing at which a phase for which a switching operation is paused transitions from a switching pause period to a switching period. The power conversion device according to claim 1 .

4. The control unit performs two-phase modulation to sequentially pause the switching operation of the switching elements of one of the three phases, and inserts three-phase modulation periods in which all three phases perform switching operations immediately before and after the timing at which the phase whose switching operation is paused when performing the two-phase modulation transitions from a switching period to a switching pause period, and immediately before and after the timing at which the phase whose switching operation is paused transitions from a switching pause period to a switching period, based on the current flowing in and out of the inverter. The power conversion device according to claim 1 .

5. The control unit controls the timing and length of the three-phase modulation period based on at least one of a current value of a three-phase output current of the inverter, a phase difference between a three-phase output voltage of the inverter and the three-phase output current, a voltage correction value for correcting an error caused by dead time, and a modulation factor of the three-phase output voltage. The power conversion device according to claim 4.

6. Three-phase modulation periods having equal phase angle widths are inserted at four locations: immediately before and after the timing at which the switching pause period starts, and immediately before and after the timing at which the switching pause period ends. The power conversion device according to claim 4.

7. The control unit generates a first three-phase voltage modulated wave based on a voltage command output from a higher-level control system, calculates a three-phase common signal for setting the switching pause period while maintaining a line voltage value for the generated first three-phase voltage modulated wave, calculates a second three-phase voltage modulated wave by superimposing the three-phase common signal on the first three-phase voltage modulated wave, and further generates the switching signal based on a third three-phase voltage modulated wave obtained by correcting an error caused by a dead time imparted to the switching signal for the second three-phase voltage modulated wave. The power conversion device according to claim 1 .

8. The control unit calculates the three-phase common signal based on a first difference which is a difference between a first three-phase voltage modulated wave having a value close to a predetermined first limiter value among the first three-phase voltage modulated waves and the first limiter value, and when performing the process of inserting the three-phase modulation period, calculates the three-phase common signal based on a second difference which is a difference between the first three-phase voltage modulated wave having a value close to the first limiter value and a second limiter value whose absolute value is smaller than the first limiter value. The power conversion device according to claim 7.

9. A third difference, which is the difference between the first limiter value and the second limiter value, is determined based on at least one of an effective value of a three-phase output current, which is an output current of the inverter, or a phase difference between a three-phase output voltage, which is an output voltage of the inverter, and the three-phase output current. The power conversion device according to claim 8 .

10. the three-phase load is a motor; The control unit performs three-phase modulation without performing two-phase modulation under an operating condition in which the rotation speed of the motor is below a predetermined threshold. The power conversion device according to claim 1 .

11. the three-phase load is a motor; The control unit determines the value of the third difference based on the value of a disturbance voltage caused by the dead time under an operating condition in which the rotation speed of the motor is lower than a predetermined threshold, thereby setting a three-phase modulation period in which each phase is switched over for the entire period. The power conversion device according to claim 9.

12. the three-phase load is a motor; The control unit does not insert the three-phase modulation period under an operating condition in which the rotation speed of the motor exceeds a predetermined threshold. The power conversion device according to claim 1 .

13. the three-phase load is a motor; The control unit sets the third difference to zero under an operating condition in which the rotation speed of the motor exceeds a predetermined threshold, thereby preventing a three-phase modulation period from being inserted for the entire period. The power conversion device according to claim 9.

14. The control unit controls the load power factor to approach 1 according to the output current of the inverter. The power conversion device according to claim 1 .

15. the three-phase load is a surface permanent magnet motor, The control unit generates a current command that makes the load power factor 1 by vector control based on the output current of the inverter. The power converter according to claim 14.

16. The power conversion device according to any one of claims 1 to 15; a compressor that compresses a refrigerant; a heat exchanger for exchanging heat of the refrigerant; a fan that blows air into the heat exchanger; At least one of a motor that drives the compressor and a motor that drives the fan is driven by the power conversion device. Air conditioning equipment.