Control device for induction motor and electric vehicle

Abstract

The purpose of the present invention is to provide a control device of an induction motor and an electric vehicle capable of controlling a voltage saturation degree of a voltage command value of a primary winding in consideration of a frequency characteristic of a control system of a width of vibration of a voltage command value of the primary winding, which changes according to a rotation speed. This control device (1) for an induction motor calculates a voltage saturation degree (∆vorv) that is an excess degree of a width of a vibration (vas) of a voltage command value with respect to a power supply voltage (vdc), integrates a value obtained by multiplying the voltage saturation degree (∆vorv) by an integration gain (Kvi), calculates a correction value (Δφdr*) of a secondary magnetic flux and corrects a command value (dr*) of the secondary magnetic flux by the correction value (Δφdr*) of the secondary magnetic flux, calculates a command value (Δφdrcr*) of the secondary magnetic flux after the correction, and decreases the absolute value of the integration gain (Kvi) as a rotation speed (ωe) of a position of the secondary magnetic flux increases.

Description

Induction motor control device and electric vehicle

[0001] The present disclosure relates to a control device for an induction motor and an electric vehicle.

[0002] When the amplitude of the oscillation of the voltage command value of the primary winding exceeds the power supply voltage, causing voltage saturation, torque controllability deteriorates and the torque oscillation component increases due to the superposition of harmonic components on the applied voltage. Therefore, in the technology of Patent Document 1, the excess amplitude of the oscillation exceeding the power supply voltage is fed back to reduce the secondary magnetic flux command value, thereby reducing the d-axis current command value, reducing the amplitude of the voltage command value oscillation, and reducing voltage saturation. This type of control is called automatic flux-weakening control.

[0003] Japanese Patent Application Laid-Open No. 2000-92899

[0004] However, Patent Document 1 does not disclose a control system design that takes into consideration the frequency characteristics of the control system for the amplitude of oscillation of the voltage command value of the primary winding, and it is not easy to maintain responsiveness and stability when the operating state of the induction motor changes or when the electrical constants of the induction motor change. In particular, since the rotation speed changes significantly, the impact on the control system cannot be ignored.

[0005] Therefore, an object of the present disclosure is to provide an induction motor control device and an electric vehicle that can control the degree of voltage saturation of the voltage command value of the primary winding by taking into account the frequency characteristics of the control system of the amplitude of oscillation of the voltage command value of the primary winding, which changes depending on the rotation speed.

[0006] A first induction motor control device according to the present disclosure is a control device for an induction motor that controls, via an inverter, an induction motor having a stator provided with primary windings of multiple phases and a rotor provided with secondary windings, and includes: a current detection unit that detects currents flowing through the primary windings of multiple phases; a current coordinate conversion unit that converts the current detection values ​​of the primary windings of multiple phases into d-axis current detection values ​​determined in the direction of the secondary magnetic flux and into q-axis current detection values ​​determined in a direction leading 90 degrees in electrical angle from the d-axis, based on the position of secondary magnetic flux generated by the secondary winding; and a voltage command value calculation unit that changes the d-axis voltage command value and the q-axis voltage command value so that the d-axis current detection value approaches the d-axis current command value and the q-axis current detection value approaches the q-axis current command value, and converts the d-axis voltage command value and the q-axis voltage command value into multi-phase voltage command values ​​based on the position of the secondary magnetic flux. a switching control unit that turns on and off a plurality of switching elements of the inverter based on the plurality of phase voltage command values ​​to apply voltages to the plurality of phase primary windings; a magnetic flux command value calculation unit that calculates a secondary magnetic flux command value; a voltage saturation compensation unit that calculates a voltage saturation degree that is the degree of excess of an amplitude of oscillation of the plurality of phase voltage command values ​​relative to a power supply voltage of the inverter based on the d-axis voltage command value and the q-axis voltage command value or the plurality of phase voltage command values, integrates a value obtained by multiplying the voltage saturation degree by an integral gain to calculate a secondary magnetic flux correction value, corrects the secondary magnetic flux command value with the secondary magnetic flux correction value, and calculates the corrected secondary magnetic flux command value; a secondary magnetic flux estimation unit that estimates secondary magnetic flux generated by the secondary winding based on the d-axis current detection value; and a current command value calculation unit that changes the d-axis current command value and sets the q-axis current command value so that the estimated value of secondary magnetic flux approaches the corrected secondary magnetic flux command value, The voltage saturation compensator reduces the absolute value of the integral gain as the rotation speed of the position of the secondary magnetic flux increases.

[0007] A second induction motor control device according to the present disclosure is an induction motor control device that controls, via an inverter, an induction motor having a stator provided with primary windings of multiple phases and a rotor provided with secondary windings, and includes: a current detection unit that detects currents flowing through the primary windings of multiple phases; a current coordinate conversion unit that converts the current detection values ​​of the primary windings of multiple phases into d-axis current detection values ​​determined in the direction of the secondary magnetic flux and into q-axis current detection values ​​determined in a direction leading 90 degrees in electrical angle from the d-axis, based on the position of secondary magnetic flux generated by the secondary winding; and a voltage command value calculation unit that changes the d-axis voltage command value and the q-axis voltage command value so that the d-axis current detection value approaches a corrected d-axis current command value and the q-axis current detection value approaches the q-axis current command value, and converts the d-axis voltage command value and the q-axis voltage command value into multi-phase voltage command values ​​based on the position of the secondary magnetic flux. the inverter comprises: a switching control unit that turns on and off a plurality of switching elements of the inverter based on the plurality of phase voltage command values, thereby applying voltages to the plurality of phase primary windings; a current command value calculation unit that calculates a d-axis current command value and a q-axis current command value; and a voltage saturation compensation unit that calculates a voltage saturation degree, which is the degree of excess of an oscillation width of the plurality of phase voltage command values ​​relative to a power supply voltage of the inverter, based on the d-axis voltage command value and the q-axis voltage command value, or the plurality of phase voltage command values, calculates a d-axis current correction value by integrating a value obtained by multiplying the voltage saturation degree by an integral gain, and corrects the d-axis current command value with the d-axis current correction value to calculate the corrected d-axis current command value, wherein the voltage saturation compensation unit decreases the absolute value of the integral gain as the rotational speed of the position of the secondary magnetic flux increases.

[0008] The electric vehicle according to the present disclosure includes the induction motor, the inverter, and a control device for the induction motor, and a power transmission mechanism that transmits the driving force of the induction motor to wheels.

[0009] With the induction motor control device and electric vehicle according to the present disclosure, it is possible to change the integral gain so as to cancel out the component that varies in proportion to the rotational speed of the secondary magnetic flux position, which is included in the frequency characteristics of the open-loop transfer function of the control system for the oscillation width of the voltage command value, and it is possible to maintain the stability and responsiveness of the feedback control system for the oscillation width of the voltage command value even when the rotational speed changes.

[0010] 1 is a schematic configuration diagram of an induction motor and an induction motor control device according to embodiment 1. FIG. 2 is a schematic block diagram of the induction motor control device according to embodiment 1. FIG. 3 is a hardware configuration diagram of the induction motor control device according to embodiment 1. FIG. 4 is a block diagram of a current command value calculation unit according to embodiment 1. FIG. 5 is a block diagram of a voltage command value calculation unit according to embodiment 1. FIG. 6 is a block diagram of a voltage saturation compensation unit according to embodiment 1. FIG. 7 is a Bode plot of a closed loop frequency characteristic according to a comparative example. FIG. 8 is a Bode plot of a closed loop frequency characteristic according to embodiment 1. FIG. 9 is a schematic block diagram of an induction motor control device according to embodiment 2. FIG. 10 is a block diagram of a voltage saturation compensation unit according to embodiment 2. FIG. 11 is a Bode plot of a closed loop frequency characteristic according to embodiment 2. FIG. 12 is a schematic configuration diagram of an electric vehicle according to embodiment 3.

[0011] 1. Embodiment 1 An induction motor control device 1 (hereinafter simply referred to as the control device 1) according to embodiment 1 will be described with reference to the drawings. Fig. 1 is a schematic diagram of an induction motor 2 and the control device 1 according to this embodiment.

[0012] 1-1. Induction Motor The induction motor 2 includes a cylindrical stator 3 and a cylindrical rotor 4 disposed radially inside the stator 3. The stator 3 is provided with multi-phase primary windings. In this embodiment, three-phase primary windings Cu, Cv, and Cw, namely, U-phase, V-phase, and W-phase, are provided. The three-phase primary windings Cu, Cv, and Cw are star-connected. Multi-phase primary windings other than three-phase may also be provided. The three-phase primary windings may also be delta-connected. A secondary winding 5 is provided on the rotor 4. The secondary winding 5 is a cage conductor. A winding other than a cage conductor may also be provided as the secondary winding 5.

[0013] The induction motor 2 is equipped with a rotation sensor 16 that outputs an electric signal corresponding to the rotation of the rotor. The rotation sensor 16 is a Hall element, an encoder, a resolver, or the like. The output signal of the rotation sensor 16 is input to the control device 1.

[0014] 1-2. Inverter, etc. The inverter 20 is a power converter that performs power conversion between the DC power supply 10 and three-phase primary windings, and has a plurality of switching elements. The inverter 20 has three sets of series circuits (legs) that correspond to the primary windings of each of the three phases, each set having a high-potential side switching element 23H (upper arm) connected to the high-potential side of the DC power supply 10 and a low-potential side switching element 23L (lower arm) connected to the low-potential side of the DC power supply 10 connected in series. The connection point between the two switching elements in each series circuit for each phase is connected to the primary winding of the corresponding phase.

[0015] Specifically, in the series circuit of each phase, the collector terminal of the high potential side switching element 23H is connected to the high potential side electric wire 14, the emitter terminal of the high potential side switching element 23H is connected to the collector terminal of the low potential side switching element 23L, and the emitter terminal of the low potential side switching element 23L is connected to the low potential side electric wire 15. The connection point between the high potential side switching element 23H and the low potential side switching element 23L is connected to the primary winding of the corresponding phase.

[0016] The switching elements may be an insulated gate bipolar transistor (IGBT) with a diode 22 connected in anti-parallel, a field effect transistor (FET) with a diode connected in anti-parallel, a metal oxide semiconductor field effect transistor (MOSFET) having the function of a diode connected in anti-parallel, or a bipolar transistor with a diode connected in anti-parallel. The gate terminal of each switching element is connected to the control device 1. Each switching element is turned on or off by a control signal output from the control device 1.

[0017] A smoothing capacitor 12 is connected between the high-potential side electric wire 14 and the low-potential side electric wire 15. A power supply voltage sensor 13 is provided to detect the power supply voltage supplied from the DC power supply 10 to the inverter 20. The power supply voltage sensor 13 is connected between the high-potential side electric wire 14 and the low-potential side electric wire 15. An output signal from the power supply voltage sensor 13 is input to the control device 1.

[0018] The current sensor 17 outputs an electric signal corresponding to the current flowing through the primary winding of each phase. The current sensor 17 is provided on the electric wire of each phase connecting the series circuit of the switching elements to the winding. The output signal of the current sensor 17 is input to the control device 1. Note that the current sensor 17 may also be provided in the series circuit of each phase.

[0019] A chargeable and dischargeable power storage device (for example, a lithium ion battery, a nickel-metal hydride battery, or an electric double layer capacitor) is used as the DC power supply 10. The DC power supply 10 may be provided with a DC-DC converter, which is a DC power converter that steps up and down the power supply voltage.

[0020] 1-3. Control Device 1 The control device 1 controls the induction motor 2 via the inverter 20. As shown in FIG. 2 , the control device 1 includes a rotation detection unit 31, a power supply voltage detection unit 32, a current detection unit 33, a current coordinate conversion unit 34, a secondary magnetic flux estimation unit 36, a magnetic flux position estimation unit 35, a magnetic flux command value calculation unit 37, a voltage saturation compensation unit 38, a torque command value setting unit 39, a current command value calculation unit 40, a voltage command value calculation unit 41, and a switching control unit 42. The functions of the control device 1 are realized by processing circuits included in the control device 1. Specifically, as shown in FIG. 3 , the control device 1 includes, as processing circuits, a processing device 90 (computer) such as a CPU (Central Processing Unit), a storage device 91 that exchanges data with the processing device 90, an input circuit 92 that inputs external signals to the processing device 90, an output circuit 93 that outputs signals from the processing device 90 to the outside, and a communication circuit 94.

[0021] The arithmetic processing device 90 may include an application-specific integrated circuit (ASIC), an integrated circuit (IC), a digital signal processor (DSP), a field programmable gate array (FPGA), various logic circuits, various signal processing circuits, etc. Furthermore, the arithmetic processing device 90 may include a plurality of the same or different types of devices, each performing a different process. The storage device 91 may include a random access memory (RAM) configured to be able to read and write data from the arithmetic processing device 90, a read-only memory (ROM) configured to be able to read data from the arithmetic processing device 90, etc. The input circuit 92 is connected to various sensors, such as the power supply voltage sensor 13, the rotation sensor 16, and the current sensor 17, and includes an A / D converter and the like that inputs output signals from these sensors to the arithmetic processing device 90. The output circuit 93 is connected to electrical loads, such as a gate drive circuit that drives switching elements on and off, and includes a drive circuit and the like that outputs control signals from the arithmetic processing device 90 to these electrical loads. The communication circuit 94 is connected to the external control device 60 and performs communication therewith.

[0022] 2 provided in the control device 1 are realized by an arithmetic processing device 90 executing software (programs) stored in a storage device 91 such as a ROM, and cooperating with other hardware of the control device 1 such as the storage device 91, an input circuit 92, and an output circuit 93. Setting data used by the control devices 31 to 42, such as the conversion coefficient Kovr, each electric constant, the target response angular frequency ωvc, and the secondary magnetic flux command value φdr*, is stored in the storage device 91 such as a ROM. Each function of the control device 1 will be described in detail below.

[0023] 1-3-1. Rotation Detection Unit 31 The rotation detection unit 31 detects the rotational angular velocity ωre of the rotor in electrical angle (hereinafter also referred to as the rotational angular frequency ωre). The rotational angular velocity ωre of the rotor is a mechanical rotational angular velocity. In this embodiment, the rotation detection unit 31 detects the rotational angular velocity ωre of the rotor based on the output signal of the rotation sensor 16. Note that the rotation detection unit 31 may be configured to estimate the rotational angular velocity based on voltage, current information, etc. without using a rotation sensor (a so-called sensorless method). For example, an observer or the like is used.

[0024] 1-3-2. Power Supply Voltage Detector 32 The power supply voltage detector 32 detects the power supply voltage vdc supplied from the DC power supply 10 to the inverter 20. In this embodiment, the power supply voltage detector 32 detects the power supply voltage vdc based on the output signal of the power supply voltage sensor 13. Note that if the power supply voltage vdc is not variable and is a specified value, the power supply voltage sensor 13 may not be provided, and a preset power supply voltage setting value may be used instead of the detected value of the power supply voltage.

[0025] The current detection unit 33 detects the currents ius, ivs, and iws flowing through the three-phase primary windings. Based on the output signal of the current sensor 17, the current detection unit 33 detects the current ius flowing through the U-phase primary winding, the current ivs flowing through the V-phase primary winding, and the current iws flowing through the W-phase primary winding.

[0026] 1-3-4. Current Coordinate Transformation Unit 34 The current coordinate transformation unit 34 transforms the three-phase primary winding current detection values ​​ius, ivs, and iws into a d-axis current detection value ids and a q-axis current detection value iqs by performing a three-phase to two-phase transformation and a rotational coordinate transformation based on the secondary magnetic flux position θe estimated by the magnetic flux position estimator 35 (described later). The d-axis is defined as the direction of the secondary magnetic flux generated by the secondary winding (the direction of the secondary magnetic flux position), and the q-axis is defined as a direction 90 electrical degrees ahead of the d-axis. For example, Equation (16) (described later) is used. The d-axis current is also called the magnetization component current, and the q-axis current is also called the torque component current.

[0027] 1-3-5. Secondary Magnetic Flux Estimation Unit 36 ​​The secondary magnetic flux estimation unit 36 ​​estimates the secondary magnetic flux φdr generated by the secondary winding based on the d-axis current detection value ids. For example, Equation (12) described below using the electrical constants of the induction motor is used. The secondary magnetic flux φdr is the magnetic flux linkage that links with the primary winding.

[0028] 1-3-6. Flux Command Value Calculation Unit 37 The flux command value calculation unit 37 calculates a command value φdr* of the secondary magnetic flux generated by the secondary winding. For example, the flux command value calculation unit 37 calculates the secondary magnetic flux command value φdr* based on the rotational angular velocity ωre of the rotor. Specifically, when the rotational angular velocity ωre of the rotor is equal to or lower than the rated rotational angular velocity at which voltage saturation begins to occur, the secondary magnetic flux command value φdr* is set to a constant value, and when the rotational angular velocity ωre of the rotor is higher than the rated rotational angular velocity, the secondary magnetic flux command value φdr* is reduced from the constant value in inverse proportion to the rated rotational angular velocity.

[0029] As will be described later, the voltage saturation compensator 38 corrects the secondary magnetic flux command value φdr* with the secondary magnetic flux correction value Δφdr* to calculate the corrected secondary magnetic flux command value φdrcr*.

[0030] Torque command value setting unit 39 sets a torque command value T* to be output from the induction motor. The torque command value T* may be transmitted from an external control device or may be calculated by the torque command value setting unit 39.

[0031] The current command value calculation unit 40 changes the d-axis current command value ids* and sets the q-axis current command value iqs* so that the secondary magnetic flux estimate value φdr approaches the corrected secondary magnetic flux command value φdrcr*.

[0032] In this embodiment, as shown in FIG. 4 , the current command value calculation unit 40 performs PI control based on the deviation between the corrected secondary magnetic flux command value φdrcr* and the estimated secondary magnetic flux value φdr to calculate the d-axis current command value ids*′. Kpφ is a proportional gain, Kiφ is an integral gain, and 1 / s represents an integrator. Feedback control other than PI control may be used. A d-axis current limiter and a q-axis current limiter are provided to calculate the final d-axis current command value ids* and the q-axis current command value iqs* by upper-limiting the d-axis current command value ids*′ and the q-axis current command value iqs*′, respectively, so that the magnitudes of the current vectors of the final d-axis current command value ids* and the q-axis current command value iqs* do not exceed the maximum current values ​​allowed by the inverter 20. The integrator has an anti-windup function that holds the integral value when the d-axis current limiter imposes an upper limit.

[0033] In this embodiment, the current command value calculation unit 40 sets the q-axis current command value iqs*' based on the torque command value T* and the corrected secondary magnetic flux command value φdrcr*. As shown in the following equation, the current command value calculation unit 40 sets the q-axis current command value iqs*' proportional to the torque command value T* and inversely proportional to the corrected secondary magnetic flux command value φdrcr*. Here, Lr is the inductance of the secondary winding, Pn is the number of pole pairs, and M is the mutual inductance between the primary winding and the secondary winding. With this configuration, even if the corrected secondary magnetic flux command value φdrcr* decreases due to an increase in the secondary magnetic flux correction value Δφdr*, the q-axis current command value iqs*' is increased, so torque can be maintained.

[0034] The q-axis current command value iqs* may be set by a rotation speed control that makes the rotation speed of the rotor follow a target speed.

[0035] 1-3-9. Voltage command value calculation unit 41 The voltage command value calculation unit 41 changes the d-axis voltage command value vds* and the q-axis voltage command value vqs* so that the d-axis current detection value ids approaches the d-axis current command value ids* and the q-axis current detection value iqs approaches the q-axis current command value iqs*, and converts the d-axis voltage command value vds* and the q-axis voltage command value vqs* into three-phase voltage command values ​​vus*, vvs*, vws* based on the secondary magnetic flux position θe.

[0036] In this embodiment, as shown in FIG. 5 , the voltage command value calculation unit 41 performs PI control based on the deviation between the d-axis current command value ids* and the d-axis current detection value ids to calculate the d-axis voltage command value vds*'. Kpd is the proportional gain, Kid is the integral gain, and 1 / s represents an integrator. Furthermore, the voltage command value calculation unit 41 performs PI control based on the deviation between the q-axis current command value iqs* and the q-axis current detection value iqs to calculate the q-axis voltage command value vqs*'. Kpq is the proportional gain, Kiq is the integral gain, and 1 / s represents an integrator.

[0037] In this embodiment, the voltage command value calculation unit 41 calculates a d-axis decoupling voltage command value Δvds and a q-axis decoupling voltage command value Δvqs based on the d-axis current detection value ids, the q-axis current detection value iqs, and the rotational speed ωe of the position of the secondary magnetic flux. For example, equations (18) and (19) described below are used. The voltage command value calculation unit 41 adds the d-axis decoupling voltage command value Δvds to the d-axis voltage command value vds*' obtained by PI control to calculate a d-axis voltage command value vds*'' and adds the q-axis decoupling voltage command value Δvqs to the q-axis voltage command value vqs*' to calculate a q-axis voltage command value vqs*''.

[0038] A d-axis voltage limiter and a q-axis voltage limiter are provided to calculate the final d-axis voltage command value vds* and the q-axis voltage command value vqs* by limiting the upper limits of the d-axis voltage command value vds*'' and the q-axis voltage command value vqs*, respectively, so that the magnitudes of the voltage vectors of the final d-axis voltage command value vds* and the q-axis voltage command value vqs* do not exceed the maximum voltage value set corresponding to voltage saturation. The d-axis voltage integrator has an anti-windup function that holds the integral value when the d-axis voltage limiter imposes an upper limit. The q-axis voltage integrator has an anti-windup function that holds the integral value when the q-axis voltage limiter imposes an upper limit.

[0039] The voltage command value calculation unit 41 converts the d-axis voltage command value vds* and the q-axis voltage command value vqs* into three-phase voltage command values ​​vus*, vvs*, and vws* by performing fixed coordinate transformation and two-phase to three-phase transformation based on the position θe of the secondary magnetic flux. Known calculation formulas are used.

[0040] In order to reduce the amplitude of the three-phase voltage command values, the voltage command value calculation unit 41 may apply amplitude reduction modulation to the three-phase voltage command values, such as third-order harmonic superposition, min-max method (pseudo third-order harmonic superposition), two-phase modulation, and trapezoidal wave modulation, so that the line voltages do not change.

[0041] The switching control unit 42 controls the on / off of multiple switching elements included in the inverter 20 based on the three-phase voltage command values ​​vus*, vvs*, and vws*, and applies voltages to the three-phase primary windings. The switching control unit 42 uses a known carrier comparison PWM or space vector PWM.

[0042] When carrier comparison PWM is used, the switching control unit 42 compares the carrier wave with each of the three-phase voltage command values ​​vus*, vvs*, and vws*, and generates switching signals that turn multiple switching elements on and off based on the comparison results.

[0043] When space vector PWM is used, the switching control unit 42 generates a voltage command vector from the three-phase voltage command values ​​vus*, vvs*, and vws*, determines the output time distribution of seven basic voltage vectors in the PWM period based on the voltage command vector, and generates switching signals that turn each switching element on and off in the PWM period based on the output time distribution of the seven basic voltage vectors.

[0044] 1-3-11. Voltage Saturation Compensation Unit 38 1-3-11-1. Configuration of the Voltage Saturation Compensation Unit 38 <Issues Due to Voltage Saturation> When voltage saturation occurs, in which the amplitude VA of the three-phase voltage command values ​​exceeds the power supply voltage VDC, torque controllability deteriorates and the torque oscillation component increases due to the superposition of harmonic components on the applied voltage. Therefore, the excess of the amplitude VA of the oscillation that exceeds the power supply voltage VDC is fed back to reduce the secondary magnetic flux command value φDR*, thereby reducing the d-axis voltage command value VDS*, reducing the amplitude VA of the oscillation, and reducing voltage saturation. This type of control is called automatic flux-weakening control.

[0045] <Calculation of Secondary Magnetic Flux Correction Value Δφdr* Based on Voltage Saturation Level (Automatic Flux-Weakening Control)> The voltage saturation compensator 38 calculates a voltage saturation level Δvovr, which is the degree to which the oscillation amplitude vas of the three-phase voltage command values ​​relative to the power supply voltage vdc of the inverter 20 exceeds the oscillation amplitude vas, based on the d-axis voltage command value vds* and the q-axis voltage command value vqs*, or the three-phase voltage command values ​​vus*, vvs*, and vws*. The voltage saturation compensator 38 integrates the voltage saturation level Δvovr multiplied by an integral gain Kvi to calculate the secondary magnetic flux correction value Δφdr*. The voltage saturation compensator 38 then corrects the secondary magnetic flux command value φdr* with the secondary magnetic flux correction value Δφdr* to calculate the corrected secondary magnetic flux command value φdrcr*. FIG. 6 shows a block diagram of the voltage saturation compensator 38 according to this embodiment. The control for changing the d-axis current command value ids* in accordance with the voltage saturation degree Δvovr is called automatic flux-weakening control.

[0046] <Calculation of Voltage Saturation Value Δvovr> In the present embodiment, the voltage saturation compensator 38 uses the following equation to calculate the oscillation amplitude vas of the three-phase voltage command values ​​based on the d-axis voltage command value vds* and the q-axis voltage command value vqs* or the three-phase voltage command values ​​vus*, vvs*, and vws*, and then calculates the voltage saturation value Δvovr by subtracting the oscillation amplitude command value vas* from the oscillation amplitude vas. In the present embodiment, the oscillation amplitude command value vas* is set to the power supply voltage vdc. The oscillation amplitude vas is the difference between the maximum and minimum values ​​of the three-phase voltage command values ​​in the oscillation period. Here, Kovr is a conversion coefficient for converting the magnitude of the vector of the voltage command values ​​into the oscillation amplitude vas of the three-phase voltage command values. The conversion coefficient Kovr is changed depending on whether amplitude reduction modulation such as the above-mentioned third-order harmonic superposition is performed. In addition, taking into consideration allowable torque vibrations, etc., the value of the conversion coefficient Kovr may be adjusted so that the amplitude of vibration of the target three-phase voltage command value becomes a predetermined value that is above or below the power supply voltage vdc.

[0047] Alternatively, various parameters that represent the degree of voltage saturation, such as a modulation rate or a voltage utilization rate, may be used as the degree of voltage saturation.

[0048] The voltage saturation compensator 38 calculates the secondary magnetic flux correction value Δφdr* by integrating a value obtained by multiplying the voltage saturation level Δvovr by the integral gain Kvi and filtering the result with a frequency characteristic Gflt (described later), as shown in the following equation. If filtering is not performed, Gflt is set to a fixed value. Here, s is the Laplace operator, and 1 / s represents an integrator.

[0049] In this embodiment, as shown in the following equation, the voltage saturation compensator 38 subtracts the secondary magnetic flux correction value Δφdr* from the secondary magnetic flux command value φdr* to calculate the corrected secondary magnetic flux command value φdrcr*.

[0050] In addition, when the voltage saturation level Δvovr is calculated by subtracting the amplitude vas of oscillation from the power supply voltage vdc, the secondary magnetic flux correction value Δφdr* may be added to the secondary magnetic flux command value φdr* to calculate the corrected secondary magnetic flux command value φdrcr*. Furthermore, addition and subtraction may be arbitrarily changed, and the integral gain Kvi may be arbitrarily changed between positive and negative.

[0051] The voltage saturation compensation unit 38 does not perform an increasing correction by increasing the secondary magnetic flux command value φdr* using the secondary magnetic flux correction value Δφdr*, but performs a decreasing correction by decreasing the secondary magnetic flux command value φdr* using the secondary magnetic flux correction value Δφdr*.

[0052] In this embodiment, the secondary magnetic flux correction value Δφdr* is subtracted from the secondary magnetic flux command value φdr*, and therefore the voltage saturation compensator 38 uses a limiter to limit the lower limit of the secondary magnetic flux correction value Δφdr*' calculated by the integrator to 0, and calculates a secondary magnetic flux correction value Δφdr* that is equal to or greater than 0. Note that when the secondary magnetic flux correction value Δφdr* is added to the secondary magnetic flux command value φdr*, the voltage saturation compensator 38 may use a limiter to limit the upper limit of the secondary magnetic flux correction value Δφdr*' calculated by the integrator to 0, and calculate a secondary magnetic flux correction value Δφdr* that is equal to or less than 0.

[0053] The integrator has an anti-windup function that holds the integral value Δφdr*′ when the limiter limits the integral value Δφdr*′ to a lower limit (or an upper limit) of 0.

[0054] <Setting of Integral Gain Kvi> The voltage saturation compensator 38 decreases the absolute value of the integral gain Kvi as the rotational speed ωe of the position of the secondary magnetic flux increases.

[0055] According to this configuration, as shown in equation (37) described later, the integral gain Kvi can be changed so as to cancel out the component that changes in proportion to the rotational speed ωe and is included in the frequency characteristics of the open-loop transfer function of the feedback control system of the oscillation amplitude vas of the voltage command value. Therefore, even if the rotational speed ωe changes, the stability and responsiveness of the feedback control system of the oscillation amplitude vas of the voltage command value can be maintained.

[0056] In this embodiment, as shown in the following equation, the voltage saturation compensator 38 changes the absolute value of the integral gain Kvi in ​​inverse proportion to the rotational speed ωe of the position of the secondary magnetic flux. Here, ωvc is the target response angular frequency of the feedback control system (automatic flux-weakening control system) of the voltage saturation degree Δvovr, and is set to a desired value. Note that the following equation is based on the premise that the integral gain Kvi is a positive value, but if the integral gain Kvi is a negative value, it is sufficient to further multiply it by -1.

[0057] In addition, taking into consideration various adaptation factors, the absolute value of the integral gain Kvi does not have to be completely inversely proportional to the rotational speed ωe of the secondary magnetic flux position, but rather it is sufficient if it decreases as the rotational speed ωe of the secondary magnetic flux position increases.

[0058] The voltage saturation compensator 38 may vary the integral gain Kvi based on the torque and rotational speed of the induction motor. For example, the voltage saturation compensator 38 varies the target response angular frequency ωvc based on the torque and rotational speed of the induction motor. The responsiveness of the control system can be appropriately adjusted according to the operating point of the torque and rotational speed of the induction motor. Here, a torque command value T* is used as the torque of the induction motor. Furthermore, the mechanical rotational angular velocity ωre of the rotor or the rotational speed ωe of the secondary magnetic flux position is used as the rotational speed of the induction motor. For example, the voltage saturation compensator 38 calculates the target response angular frequency ωvc corresponding to the current torque and rotational speed of the induction motor by referring to a characteristic function (e.g., map data or an approximate expression) that defines a predetermined relationship between the torque and rotational speed of the induction motor and the target response angular frequency ωvc.

[0059] In this embodiment, the voltage saturation compensator 38 integrates the voltage saturation difference Δvovr multiplied by a gain and filtered to calculate the secondary magnetic flux correction value Δφdr*. The frequency characteristics of the filtering are set based on the electrical constants of the induction motor.

[0060] According to this configuration, as shown in equation (37) described later, filtering can be performed in consideration of the frequency characteristics of the transfer function corresponding to the electrical constants of the induction motor, which are included in the frequency characteristics of the open-loop transfer function of the feedback control system of the oscillation amplitude vas of the voltage command value, and the stability and responsiveness of the feedback control system of the oscillation amplitude vas of the voltage command value can be improved.

[0061] As the electrical constants of an induction motor, one or both of the inductance and resistance of the induction motor are used.

[0062] The frequency characteristics of the filter processing are set to the inverse function of the frequency characteristics simulating the frequency characteristics from the corrected secondary magnetic flux command value φdrcr* to the voltage saturation degree Δvovr, excluding the components that change according to the rotational speed ωe of the secondary magnetic flux position.

[0063] According to this configuration, as shown in equation (37) described later, it is possible to cancel out the frequency characteristics of the transfer function corresponding to the electrical constants of the induction motor, which are included in the frequency characteristics of the open-loop transfer function of the feedback control system of the oscillation amplitude vas of the voltage command value, and to improve the stability and responsiveness of the feedback control system of the oscillation amplitude vas of the voltage command value.

[0064] In this embodiment, the filter processing is set to the frequency characteristics of a first-order lag transfer function, and the time constant (cutoff angular frequency) and gain of the first-order lag are set based on the electrical constants of the induction motor.

[0065] For example, the filtering process has a frequency characteristic Gflt expressed by the transfer function of the following equation.

[0066] where Ls is the inductance of the primary winding, Lr is the inductance of the secondary winding, Rr is the resistance value of the secondary winding, M is the mutual inductance between the primary winding and the secondary winding, and s is the Laplace operator. Note that each constant of the first-order lag calculated using each electrical constant may be set to a value calculated in advance using each electrical constant.

[0067] This configuration can cancel out the frequency characteristic of the transfer function according to the electrical constants of the induction motor in equation (37). As a result, as shown in the following equation, the frequency characteristic of the closed loop from the power supply voltage vdc to the oscillation amplitude vas of the three-phase voltage command values ​​becomes a first-order lag having the target response angular frequency ωvc, and stable response characteristics can be obtained.

[0068] The voltage saturation compensator 38 may change the frequency characteristics of the filtering process based on the torque and rotational speed of the induction motor. For example, the voltage saturation compensator 38 changes the electrical constants or first-order lag constants used in the filtering process based on the torque and rotational speed of the induction motor. The frequency characteristics of the filtering process can be appropriately changed to match the electrical constants that change depending on the operating point of the torque and rotational speed of the induction motor. Here, a torque command value T* is used as the torque of the induction motor. Furthermore, the mechanical rotational angular velocity ωre of the rotor or the rotational speed ωe of the secondary magnetic flux position is used as the rotational speed of the induction motor. For example, the voltage saturation compensator 38 calculates the electrical constants or first-order lag constants corresponding to the current torque and rotational speed of the induction motor by referring to a characteristic function (e.g., map data or an approximate expression) that defines a predetermined relationship between the torque and rotational speed of the induction motor and the electrical constants or first-order lag constants.

[0069] Each electrical constant of the induction motor changes depending on the temperature of each part of the induction motor. Therefore, the voltage saturation compensator 38 may change the frequency characteristics of the filter processing depending on the temperature of the induction motor. For example, the resistance value Rr of the secondary winding changes depending on the temperature of the secondary winding. In an induction motor, current flows through the secondary winding, which generates heat and causes temperature changes. The voltage saturation compensator 38 may change the resistance value Rr of the secondary winding depending on the temperature of the secondary winding. Alternatively, the voltage saturation compensator 38 may directly change each first-order lag constant depending on the temperature of the secondary winding. For example, the voltage saturation compensator 38 refers to a characteristic function (e.g., map data or an approximate formula) that defines a predetermined relationship between the temperature of the secondary winding and the resistance value Rr of the secondary winding or each first-order lag constant, calculates the resistance value Rr of the secondary winding or each first-order lag constant corresponding to the current temperature of the secondary winding, and sets each constant of the filter processing. The temperatures of each part, such as the temperature of the secondary winding, are measured using a temperature sensor or estimated using a known estimation method.

[0070] FIG. 7 shows the frequency characteristics of a closed loop according to a comparative example. Unlike the present embodiment, the comparative example, like Patent Document 1, does not change the integral gain Kvi according to the rotational speed ωe of the secondary magnetic flux position, and no filtering is performed. FIG. 8 shows the frequency characteristics of a closed loop according to the present embodiment. In each diagram, the horizontal axis represents frequency. The vertical axis in the upper row represents gain [dB], representing vas / vdc. The vertical axis in the lower row represents phase, representing the phase difference between vdc and vas. These are frequency characteristics when the vibration amplitude command value vas* (power supply voltage vdc) is changed sinusoidally and the angular frequency of the sinusoidal wave is changed.

[0071] 7 relating to the comparative example, the gain is lower than 0 dB at low angular frequencies, and it is clear that the tracking operation is not satisfactory. On the other hand, in FIG. 8 relating to the present embodiment, as shown in equation (6), the frequency characteristics are first-order lag, and the gain is 0 dB at angular frequencies lower than the target response angular frequency ωvc, and it is clear that the tracking operation is satisfactory.

[0072] Theoretical Explanation The following provides a theoretical explanation of the configuration of the voltage saturation compensator 38. First, the symbols of the parameters are defined.

[0073] <Basic Equation> As is well known, the circuit equation for an induction motor is as follows:

[0074] When the formula (8) is rewritten as a state equation, it becomes the following formula.

[0075] <Slip Frequency Control> In this embodiment, so-called slip frequency control (slip frequency type vector control) is performed, and control is performed so that φqr = 0. At this time, the torque T is expressed by the following equation.

[0076] Extracting the third and fourth lines of equation (9) gives the following equation:

[0077] When φqr = 0, the secondary magnetic flux φdr on the d-axis is φdr' and the slip angle frequency ωse is ωse'. From equation (11), the secondary magnetic flux φdr' on the d-axis and the slip angle frequency ωse' are expressed as equations (12) and (13).

[0078] The secondary magnetic flux φdr' on the d-axis is found using equation (12), and the slip angular frequency ωse' can be found using this secondary magnetic flux φdr' on the d-axis using equation (13). Using the found slip angular frequency ωse', the rotational angular frequency ωe (rotational angular velocity) of the secondary magnetic flux position and the rotational phase angle θe (rotation angle) of the secondary magnetic flux position can be found using equations (14) and (15).

[0079] Using a rotating coordinate system of the d- and q-axes that rotates at a rotational phase angle θe of the position of the secondary magnetic flux obtained by equation (15), the currents ius, ivs, and iws of the three-phase secondary winding are converted into a d-axis current ids and a q-axis current iqs using the following equations.

[0080] In slip frequency control, control is performed using equations (12) to (16). As a result, the conditions for φqr = 0 on the dq coordinates are met, and therefore equation (9) can be expressed as equation (17).

[0081] <Current Feedback Control> As shown in equations (18) and (19), a decoupling compensation term that decouples the d-axis and q-axis is added to the d-axis and q-axis voltages vds' and vqs' before decoupling compensation, resulting in the d-axis and q-axis voltages vds and vqs. As a result, as shown in equation (20), the d-axis current ids becomes a first-order lag with respect to the d-axis voltage vds' before decoupling compensation, and the q-axis current iqs becomes a first-order lag with respect to the q-axis voltage vqs' before decoupling compensation.

[0082] The d-axis and q-axis voltages vds' and vqs' before decoupling compensation are calculated by performing PI control on the current deviation between the current command value and the current, as shown in equations (21) and (22), respectively. As a result, as shown in equations (23) and (24), the d-axis current ids exhibits a first-order lag with respect to the d-axis current command value ids*, and the q-axis current iqs exhibits a first-order lag with respect to the q-axis current command value iqs*. Here, ωcc is the target response angular frequency of the current with respect to the current command value, and the larger ωcc is, the better the responsiveness becomes.

[0083] The q-axis current can be stably controlled with a first-order lag response characteristic as shown in equation (24). Therefore, as can be seen from equation (10), if the secondary magnetic flux φdr on the d-axis can be stably controlled, the torque T can be stably controlled.

[0084] <Feedback control of secondary magnetic flux> As shown in equation (25), the d-axis secondary magnetic flux φdr can be controlled by PI control of the deviation between the d-axis secondary magnetic flux φdr' estimated using equation (12) and the secondary magnetic flux command value φdr*.

[0085] Here, by setting the proportional gain Kφp as in equation (26) and the integral gain Kφi as in equation (27), it can be seen that the d-axis secondary magnetic flux φdr' can be stably controlled with a first-order lag response characteristic as shown in equation (28). ωφc is the target response angular frequency of the d-axis secondary magnetic flux φdr' with respect to the secondary magnetic flux command value φdr*, and the larger ωcc is, the better the response becomes.

[0086] <Automatic Flux-Weakening Control According to a Comparative Example> When the amplitude vas of the three-phase voltage command values ​​of the primary winding exceeds the power supply voltage vdc, the controllability of the d- and q-axis current control and secondary magnetic flux control deteriorates, and harmonic components are superimposed on the applied voltage, increasing the torque oscillation component. Therefore, the excess of the amplitude vas exceeding the power supply voltage vdc is compensated for by reducing the secondary magnetic flux command value φdr* through feedback, thereby reducing the d-axis voltage command value vds*, reducing the amplitude vas, and reducing voltage saturation. This type of control is called automatic flux-weakening control.

[0087] The automatic flux-weakening control according to the comparative example is shown in equation (29). Proportional control is not performed, and Kvp is set to 0. Unlike the present embodiment, in the comparative example, the integral gain Kvi is not changed according to the rotation speed ωe, but is set to the target response angular frequency ωvc, and no filtering is performed on the voltage deviation. Here, vas* is a command value for the amplitude of vibration, and in the present embodiment, it is set to the power supply voltage vdc.

[0088] This explains why the secondary magnetic flux φdr cannot be well controlled in accordance with the secondary magnetic flux command value φdr* when controlled as in equation (29). Assuming that slip frequency control controls φqr to 0 in equation (8), ignoring the q-axis current iqs as it has no effect on the stability of the automatic flux-weakening control in equation (29), and ignoring the voltage drop due to the resistance value Rs of the primary winding as being sufficiently small, equations (30) and (31) are obtained.

[0089] Below, the response when Equation (29) is applied to Equation (30) and Equation (31) is derived. By substituting Equation (31) into Equation (30) and ignoring the second-order differential term, Equation (32) and Equation (33) are obtained.

[0090] Since voltage saturation occurs in a region where the rotational speed ωe is higher than the rated rotational speed, if it is assumed that vds<<vqs when ωe is large, the amplitude vas of the voltage command value oscillation can be approximated by equation (34), where Kovr is a conversion coefficient for converting the magnitude of the vector of the voltage command value into the amplitude vas of the oscillation of the three-phase voltage command values, as shown in equation (2).

[0091] Substituting equation (28) into equation (34) gives equation (35).

[0092] By replacing equation (35) with equation (29) that takes into account the automatic flux-weakening control, equation (36) is obtained.

[0093] Equation (36) is feedback control of the amplitude vas of the voltage command value, and the open-loop transfer function Gop is expressed as in equation (37).

[0094] The target response angular frequency ωvc of the automatic flux-weakening control system is set to about 1 / 5 of the target response angular frequency ωφc of the secondary flux feedback control system. Therefore, when ωφc / (s+ωφc)≈1 is approximated, Equation (37) can be approximated as Equation (38).

[0095] As shown in equation (38), the automatic flux-weakening control according to the comparative example includes in its open-loop transfer function a term for ωe that changes according to the rotational speed ωe, and a proportional term and an integral term that are affected by the electrical constants of the induction motor. Therefore, the feedback characteristics of the automatic flux-weakening control according to the comparative example are affected by the rotational speed ωe and change depending on the electrical constants of the induction motor.

[0096] <Automatic Flux-Weakening Control According to This Embodiment> Next, the automatic flux-weakening control according to this embodiment is shown in equation (39). Proportional control is not performed, and Kvp=0 is set.

[0097] When the open-loop transfer function is found using the same procedure as in the automatic flux-weakening control according to the comparative example described above, it is expressed as in equation (40). In equation (38) of the comparative example, the terms multiplied by ωvc / s and corresponding to the rotational speed ωe and the electrical constants of the induction motor are cancelled out.

[0098] From equation (40), the closed-loop transfer function of the automatic flux-weakening control system is expressed by equation (41), which shows that it is not affected by the rotational speed ωe or the electrical constants of the induction motor. Furthermore, since the closed-loop transfer function of the automatic flux-weakening control system is a first-order lag, it is possible to obtain stable response characteristics without overshoot or oscillation. In this embodiment, the command value vas* of the amplitude of oscillation is set to the power supply voltage vdc.

[0099] 2. Second Embodiment An induction motor 2 and a control device 1 according to a second embodiment will be described. Description of components similar to those of the first embodiment will be omitted. The basic configuration of the induction motor 2 and the control device 1 according to this embodiment is similar to that of the first embodiment, but differs from the first embodiment in that the d-axis current correction value Δids* is calculated based on the voltage saturation degree Δvovr. Figure 9 is a schematic block diagram of the control device 1 according to this embodiment.

[0100] In this embodiment, the current command value calculation unit 40 calculates a d-axis current command value ids* and a q-axis current command value iqs*.

[0101] As will be described later, the d-axis current command value ids* is corrected by subtracting the d-axis current correction value Δids*, so the secondary magnetic flux estimate value φdr decreases by the change Δφdrid in the secondary magnetic flux corresponding to the d-axis current correction value Δids*. In this state, if the d-axis current command value ids* is calculated so that the secondary magnetic flux estimate value φdr approaches the secondary magnetic flux command value φdr*, the d-axis current command value ids* will be increased so as to cancel out the d-axis current correction value Δids*.

[0102] Therefore, the current command value calculation unit 40 calculates the change in secondary magnetic flux Δφdrid due to the d-axis current correction value Δids* based on the d-axis current correction value Δids*, as shown in the following equation, and subtracts the change in secondary magnetic flux Δφdrid from the secondary magnetic flux command value φdr* to calculate the corrected secondary magnetic flux command value φdrcr*.

[0103] The current command value calculation unit 40 changes the d-axis current command value ids* and sets the q-axis current command value iqs* so that the secondary magnetic flux estimated value φdr estimated by the secondary magnetic flux estimation unit 36 ​​approaches the corrected secondary magnetic flux command value φdrcr*.

[0104] The current command value calculation unit 40 performs PI control based on the deviation between the corrected secondary magnetic flux command value φdrcr* and the secondary magnetic flux estimated value φdr to calculate a d-axis current command value ids*', imposes an upper limit on the d-axis current command value ids*', and calculates the d-axis current command value ids*. The current command value calculation unit 40 calculates a q-axis current command value iqs*' based on the torque command value T* and the corrected secondary magnetic flux command value φdrcr*, imposes an upper limit on the q-axis current command value iqs*', and calculates the q-axis current command value iqs*. The second embodiment is similar to the first embodiment except that the method of calculating the corrected secondary magnetic flux command value φdrcr* is different from the first embodiment, and therefore description thereof will be omitted.

[0105] The voltage command value calculation unit 41 changes the d-axis voltage command value vds* and the q-axis voltage command value vqs* so that the d-axis current detection value ids approaches the d-axis current command value idscr* corrected by the d-axis current correction value Δids* and the q-axis current detection value iqs approaches the q-axis current command value iqs*, and converts the d-axis voltage command value vds* and the q-axis voltage command value vqs* into three-phase voltage command values ​​vus*, vvs*, and vws* based on the secondary magnetic flux position θe. Except for the fact that the d-axis current command value ids* is changed to the corrected d-axis current command value idscr*, this embodiment is the same as the first embodiment, and therefore a description thereof will be omitted.

[0106] 2-1. Voltage saturation compensator 38 2-1-1. Configuration of voltage saturation compensator 38 As in the first embodiment, the voltage saturation compensator 38 calculates a voltage saturation degree Δvovr, which is the degree to which the oscillation amplitude vas of the three-phase voltage command values ​​with respect to the power supply voltage vdc of inverter 20 exceeds, based on the d-axis voltage command value vds* and the q-axis voltage command value vqs* or the three-phase voltage command values ​​vus*, vvs*, vws*.

[0107] Unlike the first embodiment, the voltage saturation compensator 38 integrates the voltage saturation difference Δvovr multiplied by the integral gain Kvi to calculate the d-axis current correction value Δids*. The voltage saturation compensator 38 then corrects the d-axis current command value ids* with the d-axis current correction value Δids* to calculate the corrected d-axis current command value idscr*. FIG. 10 shows a block diagram of the voltage saturation compensator 38 according to this embodiment. Control that directly changes the d-axis current command value ids* in accordance with the voltage saturation difference Δvovr is also referred to as automatic flux-weakening control. The method of calculating the voltage saturation difference Δvovr is the same as in the first embodiment, and therefore will not be described here.

[0108] The voltage saturation compensator 38 calculates the d-axis current correction value Δids* by integrating a value obtained by multiplying the voltage saturation degree Δvovr by the integral gain Kvi and then performing a filter process with a frequency characteristic Gflt (described later) on the result, as shown in the following equation. If no filter process is performed, Gflt is set to a fixed value. Here, s is the Laplace operator, and 1 / s represents an integrator.

[0109] In this embodiment, as shown in the following equation, the voltage saturation compensator 38 subtracts the d-axis current correction value Δids* from the d-axis current command value ids* to calculate the corrected d-axis current command value idscr*.

[0110] In addition, when the voltage saturation degree Δvovr is calculated by subtracting the amplitude vas of vibration from the power supply voltage vdc, the d-axis current correction value Δids* may be added to the d-axis current command value ids* to calculate the corrected d-axis current command value idscr*. In addition, addition and subtraction may be arbitrarily changed, and the positive and negative sign of the integral gain Kvi may be arbitrarily changed.

[0111] The voltage saturation compensation unit 38 does not perform an upward correction by increasing the d-axis current command value ids* using the d-axis current correction value Δids*, but performs a downward correction by decreasing the d-axis current command value ids* using the d-axis current correction value Δids*.

[0112] In this embodiment, the d-axis current correction value Δids* is subtracted from the d-axis current command value ids*, and therefore the voltage saturation compensator 38 uses a limiter to limit the lower limit of the d-axis current correction value Δids*' calculated by the integrator to 0, and calculates a d-axis current correction value Δids* that is equal to or greater than 0. Note that when the d-axis current correction value Δids* is added to the d-axis current command value ids*, the voltage saturation compensator 38 may use a limiter to limit the upper limit of the d-axis current correction value Δids*' calculated by the integrator to 0, and calculate a d-axis current command value ids* that is equal to or less than 0.

[0113] The integrator has an anti-windup function that holds the integral value Δids*′ when the limiter limits the integral value Δids*′ to a lower limit (or an upper limit) of 0.

[0114] <Setting of Integral Gain Kvi> The voltage saturation compensator 38 decreases the absolute value of the integral gain Kvi as the rotational speed ωe of the position of the secondary magnetic flux increases.

[0115] According to this configuration, as shown in equation (52) described later, the integral gain Kvi can be changed so as to cancel out the component that changes in proportion to the rotational speed ωe and is included in the frequency characteristics of the open-loop transfer function of the feedback control system of the oscillation amplitude vas of the voltage command value. Therefore, even if the rotational speed ωe changes, the stability and responsiveness of the feedback control system of the oscillation amplitude vas of the voltage command value can be maintained.

[0116] In this embodiment, as shown in the following equation, the voltage saturation compensator 38 changes the absolute value of the integral gain Kvi in ​​inverse proportion to the rotational speed ωe of the position of the secondary magnetic flux. Here, ωvc is the target response angular frequency of the feedback control system (automatic flux-weakening control system) of the voltage saturation degree Δvovr, and is set to a desired value. Note that the following equation is based on the premise that the integral gain Kvi is a positive value, but if the integral gain Kvi is a negative value, it is sufficient to further multiply it by -1.

[0117] In addition, taking into consideration various adaptation factors, the absolute value of the integral gain Kvi does not have to be completely inversely proportional to the rotational speed ωe of the secondary magnetic flux position, but rather it is sufficient if it decreases as the rotational speed ωe of the secondary magnetic flux position increases.

[0118] The voltage saturation compensator 38 may vary the integral gain Kvi based on the torque and rotational speed of the induction motor. For example, the voltage saturation compensator 38 varies the target response angular frequency ωvc based on the torque and rotational speed of the induction motor. The responsiveness of the control system can be appropriately adjusted according to the operating point of the torque and rotational speed of the induction motor. Here, a torque command value T* is used as the torque of the induction motor. Furthermore, the mechanical rotational angular velocity ωre of the rotor or the rotational speed ωe of the secondary magnetic flux position is used as the rotational speed of the induction motor. For example, the voltage saturation compensator 38 calculates the target response angular frequency ωvc corresponding to the current torque and rotational speed of the induction motor by referring to a characteristic function (e.g., map data or an approximate expression) that defines a predetermined relationship between the torque and rotational speed of the induction motor and the target response angular frequency ωvc.

[0119] In this embodiment, the voltage saturation compensator 38 calculates the d-axis current correction value Δids* by integrating the value obtained by multiplying the voltage saturation degree Δvovr by a gain and filtering it. The frequency characteristics of the filtering are set based on the electrical constants of the induction motor.

[0120] According to this configuration, as shown in equation (52) described later, filtering can be performed in consideration of the frequency characteristics of the transfer function corresponding to the electrical constants of the induction motor, which are included in the frequency characteristics of the open-loop transfer function of the feedback control system of the oscillation amplitude vas of the voltage command value, and the stability and responsiveness of the feedback control system of the oscillation amplitude vas of the voltage command value can be improved.

[0121] As the electrical constants of an induction motor, one or both of the inductance and resistance of the induction motor are used.

[0122] The frequency characteristics of the filter processing are set to the inverse function of the frequency characteristics simulating the frequency characteristics from the corrected d-axis current command value idscr* to the voltage saturation level Δvovr, excluding the components that change according to the rotational speed ωe of the secondary magnetic flux position.

[0123] According to this configuration, as shown in equation (52) described later, it is possible to cancel out the frequency characteristics of the transfer function corresponding to the electrical constants of the induction motor, which are included in the frequency characteristics of the open-loop transfer function of the feedback control system of the oscillation amplitude vas of the voltage command value, and to improve the stability and responsiveness of the feedback control system of the oscillation amplitude vas of the voltage command value.

[0124] In this embodiment, the filter processing is set to the frequency characteristics of a transfer function that combines a first-order lag and a first-order lead. The time constant of the first-order lag (cutoff angular frequency), the time constant of the first-order lead (break-point angular frequency), and the gain are set based on the electrical constants of the induction motor.

[0125] For example, the filtering process has a frequency characteristic Gflt expressed by the transfer function of the following equation.

[0126] where Ls is the inductance of the primary winding, Lr is the inductance of the secondary winding, Rr is the resistance value of the secondary winding, M is the mutual inductance between the primary winding and the secondary winding, and s is the Laplace operator. Note that the first-order lag and first-order lead constants calculated using the respective electrical constants may be set to values ​​calculated in advance using the respective electrical constants.

[0127] This configuration can cancel out the frequency characteristic of the transfer function according to the electrical constants of the induction motor in equation (52). As a result, as shown in the following equation, the frequency characteristic of the closed loop from the power supply voltage vdc to the oscillation amplitude vas of the three-phase voltage command values ​​becomes a first-order lag having the target response angular frequency ωvc, and stable response characteristics can be obtained.

[0128] The voltage saturation compensator 38 may change the frequency characteristics of the filtering process based on the torque and rotational speed of the induction motor. For example, the voltage saturation compensator 38 changes the electrical constants or the first-order lag and first-order lead constants used in the filtering process based on the torque and rotational speed of the induction motor. The frequency characteristics of the filtering process can be appropriately changed to match the electrical constants that change depending on the operating point of the torque and rotational speed of the induction motor. Here, a torque command value T* is used as the torque of the induction motor. Furthermore, the mechanical rotational angular velocity ωre of the rotor or the rotational speed ωe of the secondary magnetic flux position is used as the rotational speed of the induction motor. For example, the voltage saturation compensator 38 calculates the electrical constants or the first-order lag and first-order lead constants corresponding to the current torque and rotational speed of the induction motor by referring to a characteristic function (e.g., map data or an approximate expression) that defines a predetermined relationship between the torque and rotational speed of the induction motor and the electrical constants or the first-order lag and first-order lead constants.

[0129] Each electrical constant of an induction motor changes depending on the temperature of each part of the induction motor. Therefore, the voltage saturation compensator 38 may change the frequency characteristics of the filter processing depending on the temperature of the induction motor. For example, the resistance value Rr of the secondary winding changes depending on the temperature of the secondary winding. In an induction motor, current flows through the secondary winding, which generates heat and causes temperature changes. The voltage saturation compensator 38 may change the resistance value Rr of the secondary winding depending on the temperature of the secondary winding. Alternatively, the voltage saturation compensator 38 may directly change the first-order lag and first-order lead constants depending on the temperature of the secondary winding. For example, the voltage saturation compensator 38 refers to a characteristic function (e.g., map data or an approximate formula) that defines a predetermined relationship between the temperature of the secondary winding and the resistance value Rr of the secondary winding or the first-order lag and first-order lead constants, and calculates the resistance value Rr of the secondary winding or the first-order lag and first-order lead constants corresponding to the current temperature of the secondary winding, thereby setting the constants of the filter processing. The temperature of each part, such as the temperature of the secondary winding, is measured using a temperature sensor or estimated using a known estimation method.

[0130] In the first embodiment, the automatic flux-weakening control system is configured as an outer loop of the flux control system, which is an outer loop of the current control system. However, in the present embodiment, the automatic flux-weakening control system is configured as an outer loop of the current control system. Therefore, in the present embodiment, the target response angular frequency ωvc of the automatic flux-weakening control system can be set higher than in the first embodiment because the flux control system is not involved, thereby achieving high response of the automatic flux-weakening control system. For example, in the first embodiment, the target response angular frequency ωvc of the automatic flux-weakening control system is set to 1 / 25 of the target response angular frequency of the current control system. In the present embodiment, for example, the target response angular frequency ωvc of the automatic flux-weakening control system is set to 1 / 5 of the target response angular frequency of the current control system. Therefore, the target response angular frequency ωvc of the automatic flux-weakening control system in the present embodiment is set to 5 times the target response angular frequency ωvc of the first embodiment.

[0131] Fig. 11 shows the frequency characteristics of the closed loop according to this embodiment. Similar to Fig. 8 of the first embodiment, the frequency characteristics are first-order lag, but the cutoff frequency (target response angular frequency ωvc) is higher than in the first embodiment. Therefore, it can be seen that this embodiment can perform a good follow-up operation up to a higher frequency than in the first embodiment.

[0132] 2-1-2. Theoretical Explanation The following provides a theoretical explanation of the configuration of the voltage saturation compensator 38. Explanations of the contents already explained in the first embodiment will be omitted.

[0133] <Automatic Flux-Weakening Control According to Comparative Example> The automatic flux-weakening control according to the comparative example is shown in equation (48). Proportional control is not performed, and Kvp is set to 0. Unlike this embodiment, in the comparative example, the integral gain Kvi is not changed according to the rotation speed ωe, but is set to the target response angular frequency ωvc, and no filtering is performed on the voltage deviation. Here, vas* is a command value for the amplitude of vibration, and in this embodiment, it is set to the power supply voltage vdc.

[0134] When the control is performed as in equation (48), the secondary magnetic flux φdr cannot be well controlled in accordance with the command value φdr* of the secondary magnetic flux. Substituting equation (12) into equation (34), equation (49) is obtained.

[0135] Since the response of the current control is sufficiently high (usually set to 5 times or more) compared to the response of the automatic flux-weakening control, the response of the current control system is ignored, and the d-axis current ids in equation (49) is replaced with the d-axis current command value ids* as shown in equation (50).

[0136] By replacing equation (50) with equation (48) that takes into account the automatic flux-weakening control, equation (51) is obtained.

[0137] Equation (51) is feedback control of the amplitude vas of the voltage command value, and the open-loop transfer function Gop is expressed as in equation (52).

[0138] As shown in equation (52), the automatic flux-weakening control according to the comparative example includes in its open-loop transfer function a term for ωe that changes according to the rotational speed ωe, and a proportional term and an integral term that are affected by the electrical constants of the induction motor. Therefore, the feedback characteristics of the automatic flux-weakening control according to the comparative example are affected by the rotational speed ωe and also change depending on the electrical constants of the induction motor.

[0139] <Automatic Flux-Weakening Control According to This Embodiment> Next, the automatic flux-weakening control according to this embodiment is shown in equation (53). Proportional control is not performed, and Kvp=0 is set.

[0140] The open-loop transfer function of the automatic flux-weakening control system according to this embodiment is calculated as shown in Equation (54). In Equation (52) of the comparative example, the terms multiplied by ωvc / s and corresponding to the rotational speed ωe and the electrical constants of the induction motor are cancelled out.

[0141] From equation (54), the closed-loop transfer function of the automatic flux-weakening control system is expressed by equation (55), which shows that it is not affected by the rotational speed ωe or the electrical constants of the induction motor. Furthermore, since the closed-loop transfer function of the automatic flux-weakening control system is a first-order lag, it is possible to obtain stable response characteristics without overshoot or oscillation. In this embodiment, the command value vas* of the amplitude of oscillation is set to the power supply voltage vdc.

[0142] As described above, the automatic flux-weakening control system of this embodiment is configured in the outer loop of the current control system, while the automatic flux-weakening control system of embodiment 1 is configured in the outer loop of the flux control system. Therefore, in this embodiment, the flux control system is not used, so the target response angular frequency ωvc of the automatic flux-weakening control system can be set higher than in embodiment 1, and high response of the automatic flux-weakening control system can be achieved.

[0143] 3. Embodiment 3 An induction motor 2 and a control device 1 according to embodiment 3 will be described. Description of components similar to those of embodiment 1 or 2 above will be omitted. The basic configuration of the induction motor 2 and the control device 1 according to this embodiment is similar to that of embodiment 1, but this embodiment differs from embodiment 1 or 2 in that the use of the induction motor 2 is specified as a driving force source for the wheels of an electric vehicle 100. Figure 12 is a schematic configuration diagram of an electric vehicle 100 according to this embodiment.

[0144] The electric vehicle 100 includes an induction motor 2, an inverter 20, an induction motor control device 1, and a power transmission mechanism 101 that transmits the driving force of the induction motor 2 to wheels 102. The electric vehicle 100 also includes a power storage device 103 that supplies a power supply voltage VDC to the inverter 20.

[0145] The power transmission mechanism 101 includes a transmission 101a and a differential gear 101b. The input shaft of the transmission 101a is connected to the rotating shaft of the induction motor 2, and the output shaft of the transmission 101a is connected to two wheels 102 (left and right) via the differential gear 101b. Note that various types of power transmission mechanisms 101 may be used, such as one in which the rotating shaft of the induction motor 2 is directly connected to the differential gear 101b without the transmission 101a. Furthermore, the induction motor 2, inverter 20, and control device 1 for driving the front wheels may be provided separately from the induction motor 2, inverter 20, and control device 1 for driving the rear wheels.

[0146] The electric vehicle 100 also includes a vehicle control device 104 that controls the running of the vehicle. The vehicle control device 104 calculates a torque command value T* and transmits it to the control device 1. The vehicle control device 104 may also detect the battery voltage output from the power storage device 103 using a voltage sensor and transmit the battery voltage as a power supply voltage vdc to the control device 1. The battery voltage of the power storage device 103 changes depending on the state of charge.

[0147] The torque of the induction motor 2 at high vehicle speeds is related to acceleration performance on expressways, for example. In particular, when torque becomes insufficient at high vehicle speeds on uphill roads, vehicle speed cannot be increased, making it necessary to improve torque at high vehicle speeds. Furthermore, there is a demand for a smaller and lighter induction motor 2 to improve fuel economy. However, reducing the size and weight of the induction motor 2 reduces the torque of the induction motor 2 and makes it more susceptible to voltage saturation. Therefore, it is important to have an induction motor control device 1 that can achieve the required maximum torque. Furthermore, from the perspectives of quietness and driving feel, the electric vehicle 100 requires stable torque response without overshoot or vibration. However, at high vehicle speeds and high torques, voltage saturation of the induction motor 2 occurs, degrading torque controllability and increasing torque vibration. While automatic flux-weakening control can reduce voltage saturation, the responsiveness of the automatic flux-weakening control system becomes an issue. Therefore, by using the control device 1 for an induction motor described in the first and second embodiments and performing automatic flux-weakening control with the responsiveness of the target response angular frequency ωvc so that the amplitude of oscillation of the voltage command value does not exceed the battery voltage and voltage saturation does not occur, a stable torque response can be achieved without overshooting or oscillation in the high vehicle speed range and high torque range, allowing the vehicle to accelerate smoothly in the high vehicle speed range.

[0148] Other Embodiments In the above embodiments, the case where a three-phase primary winding is provided has been described as an example. However, the number of phases of the primary winding may be set to any number, such as two or four, as long as there are multiple phases.

[0149] In the above-described embodiments, the case where one set of three-phase primary windings and inverters is provided has been described as an example. However, two or more sets of three-phase primary windings and inverters may be provided, and the same control as in the embodiments may be performed on each set of three-phase primary windings and inverters.

[0150] In the above third embodiment, the induction motor 2 is described as a driving force source for the wheels of an electric vehicle. However, the induction motor 2 may be connected to various driving targets other than the wheels of an electric vehicle.

[0151] In the above embodiments, as shown in equations (2), (5), and (45), the oscillation amplitude command value vas* is set to the power supply voltage vdc, and the voltage saturation difference Δvovr is calculated based on the power supply voltage vdc. However, as shown in the following equation, the magnitudes of the voltage vectors of the d-axis voltage command value vds* and the q-axis voltage command value vqs*, or the three-phase voltage command values ​​vus*, vvs*, and vws* may be calculated, and the voltage saturation difference Δvovr may be calculated based on the magnitude of the vector of the voltage command values. That is, the oscillation amplitude command value vas* may be set to a value obtained by dividing the power supply voltage vdc by the conversion factor Kovr, and division by the conversion factor Kovr may not be performed in calculating the integral gain Kvi.

[0152] In the above embodiments, the voltage saturation compensator 38 performs only integral control. However, the voltage saturation compensator 38 may perform proportional control in addition to integral control. For example, the voltage saturation compensator 38 multiplies a value obtained by multiplying the integral gain Kvi and filtering the result by a proportional gain to calculate a proportional value, and then adds the proportional value to the integral value to calculate the secondary magnetic flux correction value Δφdr*' or the d-axis current correction value Δids*'. Even if proportional control is added in this way, stability and responsiveness are not degraded because the multiplication by the integral gain Kvi and filtering are performed before the proportional control.

[0153] In the above-described embodiments, the filtering process is performed after the multiplication process by the integral gain Kvi. However, the filtering process may be performed after the multiplication process by the integral gain Kvi.

[0154] Although various exemplary embodiments and examples are described in this disclosure, the various features, aspects, and functions described in one or more embodiments are not limited to the application of a particular embodiment, but may be applied to the embodiments alone or in various combinations. Therefore, countless variations not illustrated are anticipated within the scope of the technology disclosed in the specification of this disclosure. For example, this includes cases where at least one component is modified, added, or omitted, or where at least one component is extracted and combined with components of another embodiment.

[0155] 1: Induction motor control device, 2: Induction motor, 20: Inverter, 31: Rotation detection unit, 33: Current detection unit, 34: Current coordinate conversion unit, 35: Magnetic flux position estimation unit, 36: Secondary magnetic flux estimation unit, 37: Magnetic flux command value calculation unit, 38: Voltage saturation compensation unit, 39: Torque command value setting unit, 40: Current command value calculation unit, 41: Voltage command value calculation unit, 42: Switching control unit, 100: Electric vehicle, 101: Power transmission mechanism, 102: Wheel, 103: Power storage device, Cu: Primary winding, Kvi: Integral gain, T*: Torque command value, ids*: d-axis current command value, id s: d-axis current detection value, idscr*: d-axis current command value after correction, iqs: q-axis current detection value, iqs*: q-axis current command value, vdc: power supply voltage, vds*: d-axis voltage command value, vqs*: q-axis voltage command value, vus*: U-phase voltage command value, vvs*: V-phase voltage command value, vws*: W-phase voltage command value, Δφdr*: secondary magnetic flux correction value, φdr: secondary magnetic flux estimate, φdr*: secondary magnetic flux command value, φdrcr*: secondary magnetic flux command value after correction, ωe: rotational speed (rotational angular frequency) of secondary magnetic flux position, Δvorv: voltage saturation sum

Claims

1. A control device for an induction motor that controls, via an inverter, an induction motor having a stator with primary windings of multiple phases and a rotor with secondary windings, comprising: a current detection unit that detects currents flowing through the primary windings of multiple phases; a current coordinate conversion unit that converts the current detection values ​​of the primary windings of multiple phases into d-axis current detection values ​​set in the direction of the secondary magnetic flux and q-axis current detection values ​​set in a direction 90 electrical degrees ahead of the d-axis, based on the position of secondary magnetic flux generated by the secondary winding; a voltage command value calculation unit that changes the d-axis voltage command value and the q-axis voltage command value so that the d-axis current detection value approaches the d-axis current command value and the q-axis current detection value approaches the q-axis current command value, and converts the d-axis voltage command value and the q-axis voltage command value into multi-phase voltage command values ​​based on the position of the secondary magnetic flux; and a switching control unit that turns on and off multiple switching elements of the inverter based on the multi-phase voltage command values ​​to apply voltages to the primary windings of multiple phases. a voltage saturation compensation unit that calculates a voltage saturation degree, which is the degree of excess of an oscillation width of the voltage command values ​​for the multiple phases relative to a power supply voltage of the inverter, based on the d-axis voltage command value and the q-axis voltage command value, or the voltage command values ​​for the multiple phases, integrates a value obtained by multiplying the voltage saturation degree by an integral gain, calculates a secondary flux correction value, and corrects the secondary flux command value with the secondary flux correction value to calculate a corrected secondary flux command value; a secondary flux estimation unit that estimates a secondary flux generated by the secondary winding based on the d-axis current detection value; and a current command value calculation unit that changes the d-axis current command value and sets the q-axis current command value so that the estimated value of the secondary flux approaches the corrected secondary flux command value, wherein the voltage saturation compensation unit reduces an absolute value of the integral gain as the rotational speed of the position of the secondary flux increases.

2. The control device for an induction motor according to claim 1, wherein the voltage saturation compensation unit integrates a value obtained by multiplying the voltage saturation amount by the integral gain and filtering the result to calculate a correction value for the secondary magnetic flux, and the frequency characteristics of the filtering process are set based on the electrical constants of the induction motor.

3. The control device for an induction motor according to claim 2, wherein one or both of the inductance and the resistance value of the induction motor are used as the electrical constant of the induction motor.

4. The control device for an induction motor according to claim 2, wherein the voltage saturation compensation section changes the frequency characteristic of the filtering process based on the temperature of the induction motor.

5. An induction motor control device as described in claim 2, wherein the frequency characteristics of the filter processing are set to an inverse function of a frequency characteristic simulating a frequency characteristic from the command value of the secondary magnetic flux after the correction to the voltage saturation, excluding a component that changes according to the rotational speed of the position of the secondary magnetic flux.

6. The control device for an induction motor according to claim 2, wherein the frequency characteristic of the filtering process has the frequency characteristic of a first-order lag transfer function.

7. The inductance of the primary winding is Ls, the inductance of the secondary winding is Lr, the resistance value of the secondary winding is Rr, the mutual inductance between the primary winding and the secondary winding is M, and a Laplace operator is s. The filtering process has a frequency characteristic represented by the transfer function of the following equation: The control device for an induction motor according to claim 2.

8. An induction motor control device according to any one of claims 1 to 7, wherein the voltage saturation compensation section varies the integral gain inversely proportional to the rotation speed of the position of the secondary magnetic flux.

9. The control device for an induction motor according to claim 2, wherein the voltage saturation compensation section changes the frequency characteristic of the filtering process based on the torque and rotation speed of the induction motor.

10. A control device for an induction motor which controls, via an inverter, an induction motor having a stator provided with primary windings of multiple phases and a rotor provided with secondary windings, comprising: a current detection unit which detects currents flowing through the primary windings of multiple phases; a current coordinate conversion unit which converts the current detection values ​​of the primary windings of multiple phases into d-axis current detection values ​​set in the direction of the secondary magnetic flux and q-axis current detection values ​​set in a direction 90 degrees ahead of the d-axis in electrical angle, based on the position of the secondary magnetic flux generated by the secondary winding; and a voltage command value calculation unit which changes the d-axis voltage command value and the q-axis voltage command value so that the d-axis current detection value approaches the corrected d-axis current command value and the q-axis current detection value approaches the q-axis current command value, and converts the d-axis voltage command value and the q-axis voltage command value into multi-phase voltage command values ​​based on the position of the secondary magnetic flux. a switching control unit that turns on and off multiple switching elements included in the inverter based on the multiple-phase voltage command values ​​to apply voltages to the multiple-phase primary windings; a current command value calculation unit that calculates a d-axis current command value and a q-axis current command value; and a voltage saturation compensation unit that calculates a voltage saturation degree that is an excess degree of an oscillation width of the multiple-phase voltage command values ​​relative to a power supply voltage of the inverter based on the d-axis voltage command value and the q-axis voltage command value, or the multiple-phase voltage command values, integrates a value obtained by multiplying the voltage saturation degree by an integral gain to calculate a d-axis current correction value, and corrects the d-axis current command value with the d-axis current correction value to calculate the corrected d-axis current command value, wherein the voltage saturation compensation unit reduces an absolute value of the integral gain as the rotational speed of the position of the secondary magnetic flux increases.

11. The control device for an induction motor according to claim 10, wherein the voltage saturation compensation unit integrates a value obtained by multiplying the voltage saturation amount by the integral gain and filtering the result to calculate the d-axis current correction value, and the frequency characteristics of the filtering process are set based on the electrical constants of the induction motor.

12. The control device for an induction motor according to claim 11, wherein one or both of the inductance and the resistance value of the induction motor are used as the electrical constants of the induction motor.

13. The control device for an induction motor according to claim 11, wherein the voltage saturation compensator changes the frequency characteristic of the filtering process based on the temperature of the induction motor.

14. An induction motor control device as described in claim 11, wherein the frequency characteristics of the filter processing are set to an inverse function of a frequency characteristic simulating a frequency characteristic from the corrected d-axis current command value to the voltage saturation, excluding a component that changes according to the rotational speed of the secondary magnetic flux position.

15. The control device for an induction motor according to claim 11, wherein the frequency characteristic of the filtering process has a frequency characteristic of a transfer function that combines a first-order lag and a first-order lead.

16. The inductance of the primary winding is Ls, the inductance of the secondary winding is Lr, the resistance value of the secondary winding is Rr, the mutual inductance between the primary winding and the secondary winding is M, and the Laplace operator is s. The filter processing has a frequency characteristic represented by the transfer function of the following equation: The control device for an induction motor according to claim 11.

17. The control device for an induction motor according to any one of claims 10 to 16, wherein the voltage saturation compensator changes the absolute value of the integral gain inversely proportional to the rotational speed of the position of the secondary magnetic flux.

18. The control device for an induction motor according to claim 11, wherein the voltage saturation compensation section changes the frequency characteristic of the filtering process based on the torque and rotation speed of the induction motor.

19. An electric vehicle comprising: the induction motor, the inverter, and a control device for the induction motor according to claim 1 or 10; and a power transmission mechanism for transmitting the driving force of the induction motor to wheels.

20. The electric vehicle according to claim 19, wherein the power supply voltage of the inverter is a DC voltage supplied to the inverter from a power storage device provided in the electric vehicle.

Images