Motor

Abstract

This motor includes: coils (3a to 3i) of a plurality of electromagnetic actuators among which the values of one of an electric resistance and an induced voltage constant are different; a movable element (1) being a driven body driven by the coils (3a to 3i) of the electromagnetic actuators; and a current command value determination unit (23) that determines a command value of a current caused to flow through the coil of each of the electromagnetic actuators in order to control an operation of the movable element (1). The current command value determination unit (23) determines the current command value for the coil of each of the electromagnetic actuators, so as to minimize a total sum of copper losses occurring in the coils (3a to 3i) of the plurality of electromagnetic actuators, on the basis of the values of the one of the electric resistance and the induced voltage constant of the coils (3a to 3i) of the electromagnetic actuators.

Description

Motor

[0001] The present disclosure relates to a motor.

[0002] Many rotary motors or linear motors with a classical structure are designed to only allow movement with one degree of freedom. When driving a motor with one degree of freedom, one power supply circuit is used for each motor. However, when multiple motors with one degree of freedom are combined to create a mechanical device capable of movement with multiple degrees of freedom, the mechanical structure tends to become complicated.

[0003] Recently, in order to simplify the mechanical structure, there has been an increasing number of cases where more power supply circuits than motors are used. For example, in the factory automation (FA) industry, attention has been focused on moving magnet linear synchronous motors (hereinafter abbreviated as MMLSMs), which have a primary stator winding located on the ground and a secondary magnet located on the mover. It is not uncommon for FA MMLSMs to have hundreds of power supply units installed along their travel paths. FA MMLSMs can independently control multiple movers using hundreds of power supply units. Patent Document 1 discloses an MMLSM with a branch path.

[0004] MMLSMs have a number of excellent features not found in moving coil linear synchronous motors, such as the fact that they do not require power supply to the mover, making it easy to eliminate power cables and allowing for the formation of circular travel paths, the ease of creating complex-shaped transport paths with branches, and the ability to drive multiple movers in close proximity.

[0005] Recently, multi-degree-of-freedom motors such as magnetically levitated linear motors, magnetically levitated planar motors, and rotary bearingless motors have entered the practical application stage. In these multi-degree-of-freedom motors, multiple power supply circuits are used to simultaneously perform translational / rotational and levitation movements of the mover. Because these multi-degree-of-freedom motors do not have mechanical sliding parts, they have advantages such as small mechanical losses during high-speed operation and no dust generated by wear, making them suitable for use in clean environments.

[0006] Such motors have a redundant system in terms of control, since the number of power supply circuits is greater than the number of movers or the degrees of freedom of movement allowed for the movers. In a redundant system, there are an infinite number of current command values ​​that can simultaneously achieve the desired force command and torque command.

[0007] Methods for determining the motor current command as described above are known from Patent Documents 2 to 4. In each of these documents, a simultaneous equation of a force command, a torque command, and a current command is formulated based on the positional relationship between the mover and the coil or the magnetic pole arrangement, and the current command is calculated by solving the simultaneous equation under some constraint condition. Methods that use Lagrange's undetermined constants method or a pseudoinverse matrix can be applied to solve simultaneous equations with constraint conditions, and Patent Documents 2 to 4 also use such methods.

[0008] Furthermore, as redundant motors such as those described above have recently moved from the research and development stage to the practical application stage, new issues have arisen, such as application to combinations of coils with different resistance values, which are different from those described above.

[0009] Patent No. 6633516 Patent No. 6704705 Patent No. 6938457 JP 2021-188701 JP 2000-78830

[0010] This disclosure discloses a technology for solving the above-mentioned problems and aims to provide a so-called "redundant motor" that can be applied to various motors regardless of motor structure. However, a common problem with such redundant motors is that there are an infinite number of combinations of current commands to achieve the desired thrust and torque, making it difficult to determine how to allocate the current commands to each coil. Furthermore, in order to simplify the mechanism of a moving or rotating body, there has been an increasing number of cases in which a moving or rotating body is driven more directly using multiple electromagnetic actuators (composed of coils, inverters, magnets, etc.).

[0011] To address the above-mentioned problem, there is a method of "distributing current commands so as to minimize energy loss," and this method is known from the above-mentioned Patent Document 2. Patent Document 2 explains a method for calculating a current command that minimizes energy loss when driving a moving magnet type linear synchronous motor using multiple coil units.

[0012] However, the contents of Patent Document 2 alone cannot accurately estimate energy loss for other types of redundant motors, and energy loss may increase as a result. For example, this may occur when coils with different resistance values ​​are used in combination, or when only some of the coils are at high temperatures. Furthermore, with redundant motors, other constraints may need to be considered in addition to the constraint of minimizing energy loss. For these reasons, it has become increasingly difficult to determine the current command value for electromagnetic actuators in recent mobile body drive devices or rotating body control devices.

[0013] The present disclosure discloses a technology for solving the above-mentioned problems, and introduces a new method for determining a current command value for an electromagnetic actuator used in a redundant motor. The new method makes it possible to more accurately estimate the energy loss of a redundant motor, and aims to provide an energy-saving motor.

[0014] The motor disclosed herein comprises a plurality of electromagnetic actuators each having a different value of either an electrical resistance or an induced voltage constant; a moving body or a rotating body driven by the electromagnetic actuators; and a current command value determination unit that determines a command value of a current to be passed through each electromagnetic actuator to control the operation of the moving body or the rotating body, wherein the current command value determination unit determines a current command value for each electromagnetic actuator based on the value of either the electrical resistance or the induced voltage constant of the electromagnetic actuator so as to minimize the sum of losses occurring in the plurality of electromagnetic actuators.

[0015] According to the motor of the present disclosure, a new method for determining the current command value of the electromagnetic actuator used in the redundant motor can be introduced, and this new method makes it possible to more accurately estimate the energy loss of the redundant motor used, thereby providing an energy-saving motor.

[0016] 10 is a diagram for explaining the definitions of coordinate axes introduced to explain the motor according to the first embodiment. FIG. 10 is a diagram illustrating an example of the configuration of the motor according to the first embodiment. FIG. 10 is a diagram illustrating another example of the configuration of the motor according to the first embodiment. FIG. 10 is a diagram illustrating an example of the configuration of coil current control means of the motor according to the first embodiment. FIG. 10 is a diagram illustrating an outline of the thrust coefficient of each coil constituting the motor according to the first embodiment. FIG. 10 is a diagram illustrating an example of the configuration of control arithmetic means of the motor according to the first embodiment. FIG. 10 is a diagram illustrating an example of the configuration of control arithmetic means of the motor according to the second embodiment. FIG. 10 is a diagram illustrating an example of the configuration of control arithmetic means of the motor according to the third embodiment. FIG. 10 is a diagram illustrating an example of the configuration of control arithmetic means of the motor according to the fourth embodiment. FIG. 10 is a diagram illustrating an example of the configuration of the motor according to the fifth embodiment. FIG. 10 is a diagram illustrating an example of the configuration of the control arithmetic means of the motor according to the fifth embodiment. FIG. 10 is a diagram illustrating an example of a virtual conductance calculation of the motor according to the fifth embodiment. FIG. 10 is a diagram illustrating a comparative example of the virtual conductance calculation of the motor according to the fifth embodiment. FIG. 10 is a diagram illustrating an example of virtual conductance calculated by an assumed resistance setting unit of the motor according to the sixth embodiment. FIG. 10 is a diagram illustrating an example of a calculation result of a current command of the motor according to the fifth embodiment. FIG. 10 is a diagram illustrating a comparative example of a current command calculation result of the motor according to the fifth embodiment. 12 is a diagram showing an example of a current command calculation result of the motor according to embodiment 6. FIG. 13 is a diagram showing another example of a current command calculation result of the motor according to embodiment 6. FIG. 14 is a diagram showing an example of the configuration of a motor according to embodiment 7. FIG. 15 is a diagram showing a modified configuration of the motor according to embodiment 7. FIG. 16 is a diagram showing an outline of the thrust coefficient and branch lateral force coefficient of each coil of the motor according to embodiment 7. FIG. 17 is a diagram showing an example of calculation of a current command when the motor according to embodiments 1 to 7 is not used. FIG. 18 is a diagram showing an example of a current command calculation result of the motor according to embodiment 7. FIG. 19 is a diagram showing an example of the configuration of control and calculation means of the motor according to embodiment 7. FIG. 20 is a diagram for explaining torque related to the motors according to embodiments 1 to 7. FIG. 21 is a diagram showing an example of the configuration of control and calculation means of the motor according to embodiment 8. FIG. 22 is a diagram showing an example of the configuration of a motor according to embodiment 9. FIG. 23 is a diagram showing an example of the configuration of the control and calculation means of the motor according to embodiment 9. FIG. 24 is a diagram showing an example of the configuration of a motor according to embodiment 10.FIG. 14 is a diagram showing an example of the configuration of a control and arithmetic means of a motor according to embodiment 10. FIG. 15 is a diagram showing a configuration diagram of a magnetically levitated planar motor of a motor according to embodiment 11. FIG. 16 is a diagram showing an example of the configuration of a motor according to embodiment 11. FIG. 17 is a diagram showing an example of the configuration of a control and arithmetic means of a motor according to embodiment 11. FIG. 18 is a diagram showing a cross-sectional shape of a bearingless motor of a motor according to embodiment 12. FIG. 19 is a diagram showing a case where a bearingless motor of a motor according to embodiment 12 is used in one stage. FIG. 20 is a diagram showing a case where a bearingless motor of a motor according to embodiment 12 is used in two stages. FIG. 21 is a diagram showing an example of the configuration of a motor according to embodiment 12. FIG. 22 is a diagram for explaining forces acting on a bearingless motor by breaking down the acting forces. FIG. 23 is a diagram showing an example of the configuration of a control and arithmetic means of a motor according to embodiment 12. FIG. 24 is a diagram showing an example of the hardware configuration for realizing the motors according to embodiments 1 to 12.

[0017] Embodiment 1. This disclosure relates to a motor according to embodiment 1. The motor according to embodiment 1 will be described below with reference to the drawings. FIG. 1 shows the definitions of coordinate axes introduced to explain the motor according to this disclosure. In three-dimensional space, x, y, and z coordinate axes are defined as being orthogonal to each other. Furthermore, a rotation angle θ is defined for each coordinate axis in the direction of a right-hand screw. x , θ y , θ z In the case of a motor, the x-axis is basically set in the main propulsion direction. In the case of a rotating machine, the main rotation direction is set in θ z However, in this disclosure, the path of the linear synchronous motor may be curved or circular, and therefore, when considered in an absolute coordinate system, the propulsion direction may not coincide with the x-axis direction. Therefore, here, the discussion will be based on a local coordinate system that is based on the position of the mover or rotor, rather than the absolute coordinate system.

[0018] FIG. 2 shows an example of the configuration of a motor according to the first embodiment. The motor in FIG. 2 is a mover magnet type linear synchronous motor. Hereinafter, this linear synchronous motor will be referred to simply as an LSM (Linear Synchronous Motor). This LSM has permanent magnets 11a to 11c in a mover 1. These permanent magnets 11a to 11c are magnetically coupled by a back yoke 10. Note that while FIG. 2 shows only one mover, in an actual product, multiple movers may be present on the track. Furthermore, the stator 2 has coils 3a to 3i. Due to space limitations, FIG. 2 shows only nine coils, but in an actual product, the stator 2 may have several hundred or more coils. These coils are magnetically coupled by a core back 4.

[0019] A path is formed by a plurality of stators 2, but this path is not necessarily linear, and the stators 2 may be gently curved. In other words, in actual use, for example, a path of a complex shape can be formed by combining linear stators 2 and curved stators 2. The path may also be used in a circular shape.

[0020] This LSM can move by controlling the current flowing through the coils 3a to 3i. To control the current flowing through the coils 3a to 3i, this LSM has coil current control means 5a to 5i. In FIG. 2, the LSM is configured so that multiple movers can move in close proximity to each other. To do this, it is necessary to individually control the current of each coil, so the coil current control means 5a to 5i individually control the current of each coil in accordance with the current command determined by the control calculation means 7.

[0021] Recent LSM features include "miniaturization of the mover" and "narrower pitch between the movers." To achieve this, a non-three-phase motor structure is used, which cannot be driven by 120-degree energization. If the premise of "driving by 120-degree energization" is not present, the degree of freedom in the magnetic structure increases, making it easier to increase thrust density.

[0022] The motor in Figure 2 is a type of motor that is driven by 135-degree energization, but this is merely an example. In the motor in Figure 2, the ratio of the horizontal length per mover to the horizontal length per coil is 4:1, and three permanent magnets are attached to the mover. Here, the motor structure shown in Figure 2 will be referred to as a "3-pole, 4-slot structure." If the horizontal length per magnetic pole is 180 degrees, the electrical angle per coil in a 3-pole, 4-slot structure is 135 degrees, as shown in formula (1) below.

[0023]

[0024] For this reason, to drive the motor shown in FIG. 2, the phases of the currents in the coils must be shifted by 135 degrees.

[0025] By changing the length of the mover or the arrangement of the magnets, various structures can be considered, such as a "5-pole, 6-slot structure" or a "6-pole, 8-slot structure." The phase difference of the current in each coil changes depending on the number of pole slots, but even if the structure changes, the electrical angle θ for one coil remains the same. c can be calculated using the same formula.

[0026] If thrust density or narrower pitch between the movers is not pursued, the same as in general rotating machines, θ c = 120 degrees (see FIG. 3). Here, FIG. 3 is a diagram showing another example of the motor configuration according to the first embodiment. In FIG. 3, the mover 1 has a two-pole, three-slot structure. That is, the ratio of the horizontal length per mover to the horizontal length per coil is 3:1, and the mover has two permanent magnets attached. While the mover 1 in FIG. 2 has three magnets arranged within a width corresponding to four coils, the mover 1 in FIG. 3 has two magnets arranged within a width corresponding to three coils. This is because θ c This is an example of a mover structure in which θ = 120 degrees. c = 120 degrees, it can be driven by a three-phase power supply, which reduces circuit costs.

[0027] In the device shown in Figure 2 or 3, the current in each coil can be controlled individually, so the device shown in Figure 2 or 3 can generate thrust regardless of the combination of pole slot numbers. However, to control the thrust of an LSM, it is necessary to know the magnetic pole position of the mover, just as in the case of a rotating machine. Therefore, the magnetic pole position of the mover is measured using a mover position detection means and a mover speed detection means 6. At this time, the moving speed of the mover may also be calculated at the same time. Any means can be used to detect the mover position and mover speed, but known methods include using an optical encoder, a magnetic encoder, or a video camera, for example.

[0028] FIG. 4 shows an example of the configuration of the coil current control means. Here, an example of the configuration of the coil current control means 5a is described, but the coil current control means 5b to 5i may also have a similar configuration. In FIG. 4, the coil 3a is represented by an inductor symbol. A variable voltage source 9a is connected to the coil 3a. The variable voltage source 9a may be any circuit as long as it can output an arbitrary voltage and independently control the current of each coil. The variable voltage source 9a may be configured, for example, as a single-phase inverter circuit. Alternatively, multiple variable voltage sources 9a may be created in a single circuit using a polyphase inverter circuit.

[0029] The current detection means 8a detects the current flowing through the coil 3a. The current control calculation means 12a controls the current flowing through the coil so that the current command and the current command coincide with each other, and determines the voltage command.

[0030] A well-known current control calculation method is, for example, PID control (proportional-integral-derivative control). The variable voltage source 9a operates in accordance with the voltage command determined by the current control calculation means 12a, causing the desired current to flow through the coil. When multiple variable voltage sources 9a are created in a single circuit using a multiphase inverter circuit, it is possible to perform coordinate transformation such as three-phase-to-two-phase transformation or rotational coordinate transformation (also known as dq transformation), and then control the current in the coordinate system after the coordinate transformation.

[0031] In a device like the one shown in Figure 2, the question is what current commands should be given to the coil current control means 5a to 5i. In the device shown in Figure 2, the mover moves only in the x-axis direction, so the degree of freedom of movement is one. In contrast, the number of coils that can generate thrust in the mover (i.e., coils within the range of the magnetic force of the permanent magnet) is five to six coils near the mover.

[0032] For this reason, the device in Figure 2 is a redundant control system. In a redundant control system, some constraints must be imposed when determining the current command. To reduce copper loss in the motor, a known method is to determine the current command so that the sum of the squared values ​​of the currents in each coil (hereinafter simply referred to as the current sum of squares) is minimized.

[0033] Figure 5 shows the general shape of the thrust coefficient of each coil. Here, the horizontal axis represents the mover position, and the vertical axis represents the thrust coefficient. Here, zero degrees is considered to be when the center of coil 3e and the center of the mover are aligned. Three waveforms are shown here, which represent the thrust coefficients of coils 3d to 3f, respectively. The thrust generated by each coil is expressed as the product of these thrust coefficients and the current value of each coil.

[0034] In the motor of the first embodiment, the thrust coefficient of each coil in the x-axis direction can be found by differentiating the magnetic flux linkage of each coil with respect to the mover position in the x-direction. The magnetic flux linkage of each coil is a numerical value representing the amount of magnetic flux of the permanent magnets 11a to 11c that link with each coil.

[0035] Incidentally, the thrust coefficient is also known as the induced voltage coefficient or induced voltage constant. When the magnetic flux linking each coil changes over time due to the movement of the mover, a speed electromotive force is induced in each coil. The speed electromotive force is a voltage generated by the change in the number of magnetic flux linkages over time, and can be calculated by the time derivative of the number of magnetic flux linkages. And because this formula includes the thrust coefficient, the thrust coefficient and the induced voltage coefficient are generally treated as the same in permanent magnet machines.

[0036] In addition, since the moment of force is important in rotational motion, the thrust coefficient multiplied by the length of the moment arm is sometimes called the torque coefficient or torque constant (the terms torque coefficient or torque constant are more common in rotating machines). In addition, in this disclosure, the thrust coefficient in the y direction is sometimes called the branch lateral force coefficient, and the thrust coefficient in the z direction is sometimes called the repulsive force coefficient.

[0037] This disclosure deals with translational motion in the y or z direction as well as x direction, and rotational motion around the x, y, or z axis. Although the calculation formula differs depending on the motor structure, the thrust coefficient or torque coefficient in each direction can be expressed as a function of magnetic flux linkage, so in this disclosure these thrust coefficients or torque coefficients will be collectively referred to as the induced voltage constant.

[0038] The waveform V3e shown by the solid line is basically a sinusoidal wave, although some distortion can be seen in the range of ±180 degrees. The waveform distortion seen in Figure 5 is caused by non-uniformity in the magnetic properties of the mover ends, i.e., the end effect, and is not particularly uncommon in LSMs with a small number of magnetic poles.

[0039] Beyond the range of ±180 degrees, waveform V3e, shown by the solid line, gradually attenuates and finally reaches zero. The other two waveforms have the same shape, but their phases are shifted by 135 degrees. Due to space limitations, the waveforms of coils 3a to 3c or coils 3g to 3i are not shown here, but the thrust coefficients of these coils also have waveforms that are shifted by 135 degrees.

[0040] For example, consider the current command for six coils near the mover (coils 3b to 3g in Figure 2). Here, the thrust coefficient of each coil in the x-axis direction is k x1 ~k x6 , the current of each coil is i1 to i6, and the thrust in the x-axis direction is F x The thrust force F x can be expressed as the sum of the products of the thrust coefficient and the current, as in the following equation (2).

[0041]

[0042] 2 is a redundant control system, there are an infinite number of current combinations that satisfy equation (1). Therefore, when the current command distribution that minimizes the current sum of squares is found using Lagrange's undetermined multiplier method or pseudoinverse matrix, the following equation (3) is obtained.

[0043]

[0044] However, the superscript symbol * (asterisk) means the command value, and i1 * ~i6 * : current command for each coil, F x * : represents the thrust command in the x-axis propulsion direction.

[0045] Equation (3) is formulated under the implicit assumption that the resistance of each coil is equal. When the resistance of each coil is equal, the minimum sum of squares of the current minimizes the motor's copper loss, resulting in good energy efficiency.

[0046] However, when a straight-type stator 2 and a curved-type stator 2 are prepared, the resistance value of the coil of the straight-type stator 2 is not necessarily the same as the resistance value of the coil of the curved-type stator 2. Furthermore, the thrust coefficient is not necessarily the same for straight and curved sections. Therefore, at the boundary between the straight section and the curved section, the current command distribution of equation (3) does not necessarily correspond to the case where copper loss is minimum.

[0047] In this disclosure, we propose a current command calculation formula that takes into account the resistance value of each coil in order to further improve the energy efficiency of this motor. When the resistance values ​​of each coil are represented by symbols R1 to R6, the copper loss P of the motor is R can be written as follows:

[0048]

[0049] Using Lagrange's undetermined constant method, the copper loss P of the motor in equation (4) is calculated. R When the current command distribution that minimizes is found, the following equation (5) is obtained.

[0050]

[0051] Comparing equation (3) and equation (5), we can see that the contents of the equations differ by the amount of the resistance values ​​R1 to R6 of each coil. x1 ~k x6 is a square, while the resistance values ​​R1 to R6 of each coil are a linear. To our knowledge, such a formula has not been disclosed in any other prior art document. Note that, although formulas for six arbitrary coils were considered here, the current command can be determined using a similar formula even if the number of coils is changed.

[0052] Incidentally, the current command may be determined taking into account other constraints. For example, θ c For a linear synchronous motor such that π / 3, consider the constraints shown in the following equation (6): This is a constraint when driving a linear synchronous motor with a three-phase power supply.

[0053]

[0054] When current command distribution is calculated with this constraint condition added, the following formula is obtained: where A, B, etc. are as shown in the following formula (8).

[0055]

[0056]

[0057] Here, the superscript −1 means the inverse matrix, and the subscripts m, l, and o are integers within the range of values ​​shown in the following equation (9).

[0058]

[0059] In order to generate thrust in a mover magnet type linear synchronous motor as shown in Figure 2, it is necessary to select the coils to be energized appropriately. In this case, if the coils to be energized are selected appropriately, the inverse matrix B -1 Thus, even when other constraints are added, it is possible to calculate the current command distribution taking into account the resistance values ​​R1 to R6 of each coil.

[0060] Note that this calculation of current command distribution may be performed in other coordinate systems. In the field of motor control, coordinate transformations such as three-phase / two-phase transformation or rotational coordinate transformation (also known as dq transformation) are standard techniques. Of course, the calculation of current command distribution may be performed in a coordinate system after performing these coordinate transformations.

[0061] 6 shows an example of the configuration of the control and calculation means of the motor according to embodiment 1. The control and calculation means 7 has a position and speed control means 21, a thrust coefficient reference means 22, and a current command calculation means 23 (hereinafter also referred to as a current command value determiner 23). The position and speed control means 21 operates based on the position command and determines the thrust command. The position and speed control means 21 is configured as a feedback controller or a feedforward controller.

[0062] For example, a PID controller may be used as the feedback controller. Here, the mover position or mover speed is fed back to the position / speed control means 21 for control. The thrust coefficient reference means 22 calculates the thrust coefficient of each coil. As the thrust coefficient of each coil changes depending on the mover position as shown in FIG. 5, it is calculated based on the mover position. The current command calculation means 23 calculates a current command based on the thrust coefficient of each coil, the resistance value of each coil, and the thrust command. The current command calculation means 23 determines the distribution of the current commands using the above-mentioned equation (5).

[0063] By configuring the motor in this way, it is possible to calculate the current command that minimizes copper loss even when using electromagnetic actuators with different characteristics. In the case of a mover magnet type linear synchronous motor like the one shown in Figure 2, the characteristics of the electromagnetic actuator often differ at the boundary between the straight section and the curved section, but by using this calculation formula, it is possible to drive the motor more efficiently.

[0064] Although the constraint here is to minimize the copper loss of the motor, it is also possible to consider mathematical expressions for constraints that include other losses such as mechanical loss or iron loss. It is also possible to use the loss of the entire system as a constraint.

[0065] Embodiment 2 In a redundant control system device equipped with multiple electromagnetic actuators as shown in Figure 2, it may be necessary to continue operation even if a failure occurs in any of the coils 3a to 3i or the coil current control means 5a to 5i. In embodiment 2, a linear synchronous motor that can continue operation even in the event of a failure will be described.

[0066] 7 shows an example of the configuration of the control calculation means of embodiment 2. This is the linear synchronous motor described in embodiment 1, to which a fault detection means 24 and a parameter correction means 25 are added. The fault detection means 24 diagnoses the state of each coil and each coil current control means from the current value of each coil or data from a temperature sensor (not shown) or the like.

[0067] If it is determined that a fault has occurred in any of the coils, a fault alarm is sent to the parameter correction means 25. The parameter correction means 25 corrects the resistance value of the coil in which the fault has occurred to infinity.

[0068] As in the first embodiment, the current command calculation means 23 calculates a current command based on the thrust coefficient of each coil, the resistance value of each coil, and the thrust command, but if the resistance value of a coil in which an abnormality has occurred is set to infinity, the current command for that coil will always be 0. Then, thrust control of the linear synchronous motor is performed using only normal coils and coil current control means that are not in an abnormal state.

[0069] By configuring the control calculation means 7 in this way, even if a failure occurs in any of the coils 3a to 3i or the coil current control means 5a to 5i, operation can be easily continued.

[0070] Embodiment 3. When a linear synchronous motor is used as the motor in embodiment 3, the temperature of the coils changes due to losses generated inside the motor while the linear synchronous motor is running. This temperature change can be examined to calculate the resistance value of each coil online, and the result can be reflected in the current command calculation.

[0071] 8 shows an example of the configuration of the control and calculation means of a linear synchronous motor when a linear synchronous motor is used as the motor in embodiment 3. In this case, the control and calculation means 7 includes resistance calculation means 26. In embodiment 3, the temperature of each coil is calculated by a temperature sensor (not shown), and the resistance calculation means 26 calculates the resistance value of each coil based on the calculated temperature.

[0072] Then, the current command calculation means 23 calculates a current command based on the thrust coefficient of each coil, the resistance value of each coil, and the thrust command, as in embodiment 1. By configuring the control calculation means 7 in this way, it is possible to measure changes in resistance in real time, and to drive the linear synchronous motor with more energy savings.

[0073] Fourth Embodiment In the first to third embodiments, the current command is calculated using the actual resistance value, but the resistance value used in this calculation does not necessarily have to be the actual value.

[0074] Here, we consider a fictitious value called "virtual resistance value (hereinafter also referred to as assumed resistance value)" (here, this refers to a resistance value that is temporarily determined when calculating electrical loss, which differs from the actual electrical resistance value. This assumed resistance is also called assumed resistance. The same applies below) in addition to the true resistance value (here, true resistance value refers to the actual electrical resistance value, for example, the resistance value when the environmental conditions do not change (for example, the temperature is constant). The same applies below). Also, in addition to the true copper loss, we consider a fictitious value called "virtual copper loss". The "virtual copper loss" is calculated using the "assumed resistance value" and the motor current. Since the "assumed resistance value" is an fictitious value, it can be set freely. Here, the assumed resistance value of each coil is set as R ν1 ~R ν6 When expressed as the symbol, the virtual copper loss of the motor P RV can be written as shown in the following equation (10).

[0075]

[0076] Here, when the current command distribution that minimizes the motor copper loss in equation (10) is calculated using Lagrange's undetermined constant method, the following equation (11) is obtained. ν1 ~Rν6 It has been replaced with.

[0077]

[0078] If the current command is determined so as to minimize virtual copper loss, it is possible to change the current waveform by changing the value of the assumed resistance. Therefore, by setting the assumed resistance value of the coil you want to energize low and the assumed resistance value of the coil you do not want to energize high, and then solving the simultaneous equations using Lagrange's method of undetermined multipliers, you can determine the current command according to the ratio of the set assumed resistances. The current command determined by this calculation is the combination of current commands that will minimize virtual copper loss and can obtain the desired thrust.

[0079] The reciprocal of resistance is called "conductance," but here, the reciprocal of the above-mentioned assumed resistance will be called "virtual conductance" (the same applies below). When performing calculations of Equation (5) or Equation (11) on a computer, it is convenient to handle numerical data in the form of conductance or virtual conductance. Therefore, in the following explanation, the terms conductance or virtual conductance may be used instead of the terms resistance or assumed resistance.

[0080] Calculating the current command using the assumed resistance has various advantages. In the fourth embodiment, as an example, a method of preventing further temperature rise by reducing the current flowing through a coil that has reached a high temperature will be described.

[0081] 9 shows an example of the configuration of the control calculation means of the motor according to embodiment 4. In embodiment 4, an assumed resistance setting unit 27 is provided instead of the resistance calculation means 26 of embodiment 3. In embodiment 4, the temperature of each coil is measured by a temperature sensor (not shown), and based on this measured temperature data, the assumed resistance setting unit 27 calculates the assumed resistance value of each coil.

[0082] Then, the current command calculation means 23 calculates a current command based on the thrust coefficient of each coil, the assumed resistance value of each coil, and the thrust command. By configuring the control calculation means 7 in this way, it is possible to easily reduce the current flowing through coils that have become hot. This makes it possible to prevent further temperature increases in the coils that have become hot, distributing the heat load on the circuit or motor coil and enabling high-power operation to continue for a longer period of time compared to conventional linear synchronous motors that do not have an assumed resistance setting unit 27.

[0083] Fifth Embodiment In the fifth embodiment, a linear synchronous motor having a plurality of movers will be described. Fig. 10 shows an example of the configuration of a linear synchronous motor according to the fifth embodiment.

[0084] In a mover magnet type linear synchronous motor, multiple movers may run simultaneously (simultaneous running), so two movers, mover 1a and mover 1b, are shown in Fig. 10. Note that due to space limitations, only two movers are shown in Fig. 10, but three or more movers may run simultaneously.

[0085] If the coil current control means 5a-5i are configured to freely control the current in each coil, it is possible to drive two movers close to each other (to run simultaneously) without them touching each other. For example, since three magnets are arranged in each of the movers 1a and 1b in Figure 9, the mover length in the x-direction is 540 degrees (= 180 degrees × 3 poles). If the current in each coil can be freely controlled, the distance between the right end of mover 1a and the left end of mover 1b can be made smaller than the 540-degree length of each mover. The distance between the two movers can also be made smaller than the width of each coil (135 degrees). However, if the distance between the two movers is made smaller than 135 degrees (the width of each coil), there will be a coil where the magnetic flux generated by the permanent magnets of the two movers simultaneously interlinks. In Figure 10, coil 3e corresponds to this. Coil 3e is a coil close to permanent magnets 11c and 11d.

[0086] When a current is passed through this coil, a thrust is generated simultaneously in movers 1a and 1b. If permanent magnets 11c and 11d have the same polarity (both are N poles in Figure 10), the thrust generated in movers 1a and 1b will be in opposite directions. If you want to move two movers in the same direction while they are close to each other, it is not desirable to pass a current through such a coil.

[0087] To determine the current to flow through such a coil, it is sufficient to formulate simultaneous equations relating the thrust and current of multiple movers and solve those simultaneous equations, but as the number of adjacent movers increases, the calculations become more complicated. Also, since real-time calculations are important in this type of control device, complex calculations are not preferred. Therefore, we considered simplifying the calculations by using the concept of assumed resistance.

[0088] Fig. 11 shows an example of the configuration of the control and calculation means of the motor of embodiment 5. Fig. 11 shows a control system for one mover. Each mover is provided with the control and calculation means of Fig. 11. Therefore, when two movers are to be moved simultaneously, two control and calculation means of Fig. 11 are prepared.

[0089] The control calculation means of the fifth embodiment includes an assumed resistance setting unit 27a that calculates an assumed resistance based on the mover position. Then, the current command calculation means 23 calculates a current command based on the thrust coefficient of each coil, the assumed resistance value of each coil, and the thrust command.

[0090] Figures 12A, 12B, and 12C show examples of virtual conductance calculations. The horizontal axes in Figures 12A, 12B, and 12C represent the x-axis position of each mover. Here, zero degrees is defined as the position where the center of each coil coincides with the center of each mover. Hereinafter, when referring to the distance between a certain coil and a certain mover, this distance refers to the distance in the x-axis direction between the centers of those coils and the centers of the movers. For example, in Figure 10, the distance between mover 1a and coil 3d is approximately 1.5 coils, or 202.5 degrees. Here, Figure 12A shows an example of virtual conductance when multiple movers are close to each other. When multiple movers are close to each other, coils near both ends of each mover simultaneously affect the thrust of both movers. By setting the assumed resistance of such coils to infinity (virtual conductance to zero), it is possible to easily prohibit current flow to coils with infinite assumed resistance.

[0091] In the case of a motor with the structure of Figure 10, the coils described above are always at least ±202.5 degrees away from the center of each mover, so in Figure 12A, when the distance between the center of each mover and the center of each coil is at least ±202.5 degrees, the virtual conductance is set to zero.

[0092] 12A, the virtual conductance decreases linearly from the ±180° point, but this is merely an example. The point where the decrease begins does not necessarily have to be ±180°, and the decrease may be curved. Furthermore, the point where the virtual conductance becomes zero may be any other point as long as it is inside ±202.5°.

[0093] However, if the change in virtual conductance is discontinuous, the current command will also be discontinuous, so care must be taken. For example, if the virtual conductance is changed in steps, the current command will change suddenly, but the coil current will not be able to transiently follow the command value. Therefore, it is necessary to make the change in virtual conductance gradual enough to allow the coil current to follow the command value. This makes it possible to prevent discontinuities when there is a branch.

[0094] FIG. 12B is a graph for comparison, showing an example of setting the virtual conductance when there are no other movers near the mover 1a. In the case of a motor with the structure of FIG. 10, coils within a distance of approximately ±360 degrees from the mover can generate thrust. Therefore, in FIG. 12B, when the distance between the mover and each coil is within ±360 degrees, the virtual conductance is set to an arbitrary non-zero value. If the distance increases beyond this, the virtual conductance is gradually reduced and ultimately becomes zero. The curve in FIG. 12B is merely an example, and the detailed shape can be freely determined. FIG. 12C will be described in the sixth embodiment below.

[0095] Figures 13A and 13B are examples of current command calculation results for the motor of embodiment 5. Three waveforms are shown here, which are current commands for coil current control means 5d to 5f, respectively. These waveforms are calculated based on the thrust coefficients shown in Figure 4, and are therefore shifted by 135 degrees. The current commands for the other coils are omitted due to space limitations, but they have similar waveforms shifted by 135 degrees. Figure 13A shows the results of calculating a current command using the virtual conductance shown in Figure 12A. For comparison, Figure 13B shows the results of calculating a current command using the virtual conductance shown in Figure 12B. Note that the thrust command here is an arbitrary non-zero value.

[0096] 13A, only coils whose distance from the mover is within ±202.5 degrees are used, so the current command for each coil has a shape similar to a sinusoidal waveform for one period. Although distortion can be seen in the waveform of the current command due to the end effect of the mover, it is generally sinusoidal.

[0097] 13B, since coils that are at a distance of ±202.5 degrees or more from the mover are also used, the current command for each coil has a shape similar to a sine wave waveform for two periods. Due to the end effect of the mover, both ends of the current command are slightly attenuated, but this also remains roughly sinusoidal.

[0098] By changing the assumed resistance in this way, the current command for any coil can be easily set to zero. This can be used to easily determine the current command when driving multiple movers in close proximity. This is a great advantage when implementing a controller on a computer.

[0099] From the perspective of current command calculation, setting the assumed resistance of a certain coil to infinity (or setting the virtual conductance to zero) is equivalent to regarding the thrust coefficient of that coil as zero. If the assumed resistance is set to infinity, the current command of that coil will always be zero, but even if the thrust coefficient is set to zero instead, the current command of that coil will also be zero. Therefore, instead of the assumed resistance setting unit 27a, a calculation means may be provided that corrects the thrust constant to a value different from the true value.

[0100] Sixth Embodiment In a sixth embodiment, a case will be described in which the assumed resistance is changed according to the distance between the movers.

[0101] Since the copper loss of a motor is a function of the square of the current, if the current concentrates in one coil even if the thrust is the same, the copper loss will increase accordingly. Therefore, when comparing the current command waveforms in Figure 13A and 13B, if the thrust command is the same, the copper loss in Figure 13A will be greater because fewer coils are used.

[0102] The current command calculation method described in the fifth embodiment is suitable for driving multiple movers in close proximity, but has the drawback of increasing copper loss when the distance between the movers increases. In order to reduce copper loss, the number of coils used can be increased, but with conventional calculation methods, the current command changes suddenly when the number of coils used is changed. A sudden change in the current command is not desirable for position control or speed control. Therefore, in the sixth embodiment, a method is described in which the assumed resistance is changed according to the distance between the movers, and the number of coils used is continuously changed.

[0103] Fig. 14 shows an example of the configuration of the control calculation means of embodiment 6. Fig. 14 shows a control system for mover 1a. Although not shown in the figure, the control system for mover 1b has a similar configuration. Furthermore, when three or more movers are moved simultaneously, similar control systems are prepared for the number of movers.

[0104] The control calculation means of the sixth embodiment includes a subtractor 28 that calculates the distance between the movers, and an assumed resistance setting unit 27b that calculates an assumed resistance value (assumed resistance) based on the subtraction result. Then, a current command calculation means 23 calculates a current command based on the thrust coefficient of each coil, the assumed resistance value of each coil, and the thrust command.

[0105] Fig. 12C shows an example of virtual conductance calculated by the assumed resistance setting unit 27b of embodiment 6. The assumed resistance setting unit 27b of embodiment 6 changes the virtual conductance depending on the distance between the movers. Specifically, when the distance between the movers is short, the state shown in Fig. 12A is set, and when the distance between the movers is sufficiently far, the state shown in Fig. 12B is set. Depending on the distance between the movers, the virtual conductance may be set to a state between Fig. 12A and Fig. 12B. Note that the mover length, which is the length of the movers in the main direction of travel, is used here as an evaluation criterion for determining the degree of closeness of the distance between the movers.

[0106] It is not always the case that there are other movers close to both sides of a mover. For example, when two movers are driven close to each other, there is another mover close to one side, but no mover close to the other side. Therefore, it is perfectly acceptable to set the virtual conductance asymmetrically.

[0107] 15A and 15B show examples of current command calculation results in embodiment 6. Four waveforms are shown here, which are current commands for coil current control means 5c to 5f. These waveforms are calculated based on the thrust coefficients shown in FIG. 4, and are therefore shifted by 135 degrees.

[0108] For convenience, the case where the virtual conductance in Fig. 12A is used is referred to as the "three-coil current mode," and the case where the virtual conductance in Fig. 12B is used is referred to as the "five-coil current mode." The three-coil current mode is suitable for proximity driving, but has large copper loss. Conversely, the five-coil current mode has small copper loss, but does not allow proximity driving.

[0109] Fig. 15A shows an example of a waveform of a current command when switching from a 5-coil current-carrying mode to a 3-coil current-carrying mode. Conversely, Fig. 15B shows an example of a waveform of a current command when switching from a 3-coil current-carrying mode to a 5-coil current-carrying mode. Also, here, the section in which the distance between the movers changes and the assumed resistance value (also called assumed resistance) changes is referred to as a "mode switching section," but the change in the current command in the mode switching section is smooth and not discontinuous.

[0110] By using the concept of assumed resistance in this way, the number of coils used can be freely changed, and multiple movers can be freely controlled with less energy.

[0111] Seventh Embodiment It is not uncommon for recent mover magnet type linear synchronous motors to be capable of forming complex paths including branch paths. Therefore, in the following, when the motor according to the seventh embodiment is such a mover magnet type linear synchronous motor, a current command calculation method will be discussed in the case where this linear synchronous motor is capable of forming paths including branch paths.

[0112] Fig. 16 shows an example of the configuration of a motor according to embodiment 7. Fig. 17 shows a modified example of the motor according to embodiment 7. These devices control movers 1a2, 1b2 using coils 3a to 3r and coil current control means 5a to 5r. Permanent magnets 11a to 11c2 and 11d to 11f2 are attached to both the front and back sides of back yokes 10a, 10b of movers 1a2, 1b2.

[0113] Coils 3a to 3i are magnetically coupled to the core back 4, coils 3n to 3r in Fig. 16 are magnetically coupled to the core back 4b2, and coils 3j to 3r in Fig. 17 are magnetically coupled to the core back 4b. Here, the path formed by coils 3a to 3i is called the first path, and the path formed by the iron core module 8 and coils 3n to 3r (or coils 3j to 3r) is called the second path.

[0114] The difference between Figure 16 and Figure 17 is the iron core module 8 and coils 3j to 3m on the second track side. In Figure 16, to reduce the cost of the power circuit, an iron core module 8 made of magnetic steel plate or a magnetic material such as iron is placed. On the other hand, in Figure 17, coils 3j to 3m are placed in this area. In terms of copper loss in the motor, the more coils that can be used, the more advantageous it is, so Figure 17 is preferable to Figure 16. You can freely choose which configuration to use depending on your needs.

[0115] The movers 1a2 and 1b2 can select whether to run along the first or second path depending on the branching lateral force (force in the y direction) generated by the coils 3a to 3r and the coil current control means 5a to 5r. The movers 1a2 and 1b2 are attracted to one of the paths by the magnetic attractive force generated between the permanent magnet and the stator coil core (or iron core module). This magnetic attractive force allows the movers 1a2 and 1b2 to maintain the selected path, but if a magnetic branching lateral force (also called a magnetic lateral force) large enough to overcome this force is generated, the movers can change to the other path.

[0116] The movers 1a2 and 1b2 can also receive thrust in the x direction from the coils 3a to 3r, so the control and calculation means 7a and 7b (or 7b2) must simultaneously control the thrust in the x direction and the branch lateral force in the y direction of the movers 1a2 and 1b2.

[0117] Although the control and calculation means 7a and 7b (or 7b2) are arranged as two separate units due to space limitations, they do not necessarily have to be separated into two. They may be configured on a single CPU or on separate CPUs.

[0118] The control calculation means 7a and 7b (or 7b2) operate based on position commands and path selection commands for the movers 1a2 and 1b2, and transmit current commands to the coil current control means 5a to 5r. At this time, the question is what kind of current command should be transmitted.

[0119] Here, the formula for calculating the branch lateral force will be explained. For example, consider the current command for six coils near the mover 1a2 on the first track side (in FIG. 16, the coil (not shown) to the left of coil 3a and five coils 3a to 3e). Here, the branch lateral force coefficient in the y-axis direction of each coil is k y1 ~k y6 The force acting in the y-axis branch direction is represented by the symbol Fy. y can be expressed as the sum of the products of the branch lateral force coefficient and the current, as shown in the following equation (12).

[0120]

[0121] A desired thrust F is applied to the mover 1a2. x and branch lateral force F y In order to simultaneously generate the above equations, it is sufficient to solve simultaneous equations relating to the thrust, branch lateral force, and current. When the simultaneous equations are solved using a commonly known pseudo-inverse matrix and a current command is calculated, the following equation (13) is obtained.

[0122]

[0123] Here, F y * : is the branch lateral force command in the y direction. Calculations based on this formula can minimize the sum of squares of the currents i1 to i6 in each coil. In other words, formula (13) is the result of solving the simultaneous equations relating to thrust, branch lateral force, and current under constraints that minimize the sum of squares of currents i1 to i6. However, when the current command is calculated using formula (13), it is difficult to drive two or more movers in close proximity to each other.

[0124] The linear synchronous motor according to embodiment 7 is required to simultaneously control a plurality of movers in close proximity to each other, as in the linear synchronous motors according to embodiments 5 and 6. As described above, in order to simplify the current command calculation, it is necessary to prohibit the supply of current to the coils that generate thrust or branch lateral force to the two movers.

[0125] One possible method for prohibiting the supply of current to a specific coil (i.e., setting the current command for that coil to zero) is to overwrite the thrust coefficient or branch lateral force coefficient of that coil with zero. However, this method has the problem of creating a point where the current command changes abruptly.

[0126] Figure 18 is a graph showing the thrust coefficients and branch lateral force coefficients of each coil. This graph shows the thrust coefficients and branch lateral force coefficients of coil 3e. The coefficients of the other coils are not shown, but the waveforms are obtained by shifting the coefficients of coil 3e by 135 degrees. Ignoring attenuation due to end effect, the branch lateral force coefficient Y3e and thrust coefficient V3e resemble a sine wave and cosine wave that are 90 degrees out of phase.

[0127] In order to move the two movers close to each other, it is desired to set the current command for the coil (coil 3e in FIG. 16) that affects the thrust and branch lateral force of the two movers to zero. If the current command for that coil is i6 * If so, in the current command distribution formula (11), i * To make k = 0, x6 Tok y6 Just overwrite with zero. x6 Tok y6 Whatever the true value of k x6 Tok y6 If the current command is calculated assuming that is zero, then i6 * = 0. However, this method does not allow a current command with a smooth shape to be obtained.

[0128] FIG. 19 shows an example of a current command calculation when the linear synchronous motor of the present disclosure is not used. This graph shows three waveforms, all of which are current commands for coil 3e. The current commands for the other coils are these waveforms shifted by 135 degrees. Here, in order to move multiple movers in close proximity, the current commands for coils that are more than ±202.5 degrees away from the center of each mover are set to zero. Note that by setting the coil current command to zero, it is possible to suppress the current of a specific electromagnetic actuator (for example, one that includes a broken coil or a coil that is generating abnormal heat as a component).

[0129] The current command IRe is shown for comparison purposes and is the current command when no branch lateral force control is performed. Since the thrust coefficient graph becomes zero near ±180 degrees, a smooth current command can be obtained in this case without using the present disclosure.

[0130] Current commands IRe1 and IRe2 are examples of waveforms when thrust and branch lateral force are controlled simultaneously. When branch lateral force and thrust are controlled simultaneously, the current command has a shape with noticeable protrusions or steps, like IRe1 or IRe2. These protrusions or steps become more noticeable as the branch lateral force command becomes larger. Comparing IRe1 and IRe2, IRe2 has a larger branch lateral force command, but the protrusions or steps are larger.

[0131] This step or protrusion occurs because the current command for the coils whose center is more than ±202.5 degrees away from the center of each mover is forcibly set to zero (in other words, the current command is calculated assuming that the branch lateral force coefficient and thrust coefficient of that coil are zero). Figure 18 shows that the point where the branch lateral force coefficient and thrust coefficient are simultaneously zero is actually in the range of ±400 degrees or more. However, if the branch lateral force coefficient and thrust coefficient are assumed to be zero in that inner range, the protrusion or protrusion described above will occur when switching the coil to be energized.

[0132] Because the response speed of current control is limited, the coil current cannot transiently follow the command value. Therefore, the current command should have as smooth a waveform as possible. However, without utilizing the present disclosure, if a current command is calculated for each mover, a spike or step will occur in the current command. If a simultaneous equation is formulated that simultaneously considers the thrust and branch lateral forces of multiple movers, such current spikes or steps will be eliminated, but the amount of calculation required to solve the simultaneous equations will be extremely large. Therefore, without utilizing the present disclosure, it is extremely difficult to drive multiple movers in close proximity near the branch paths of a linear synchronous motor.

[0133] The present disclosure has been devised to solve such problems, and applies the concepts of the assumed resistance value and virtual copper loss described in the fourth to sixth embodiments. When a current command that satisfies the desired thrust command and branch side force command and minimizes the virtual copper loss is obtained, the following equation (14) is obtained.

[0134]

[0135] FIG. 20 shows an example of the current command calculation results of the seventh embodiment. This graph shows three waveforms, all of which are current commands for coil 3e. The current commands for the other coils are these waveforms shifted by 135 degrees. Here, in order to move multiple movers in close proximity, the current commands for coils that are more than ±202.5 degrees away from the center of each mover are set to zero. However, rather than regarding the thrust coefficient or branch lateral force coefficient as zero, the current commands are set to zero using equation (14) and the virtual conductance shown in FIG. 12A.

[0136] The waveform of IRe shown for comparison is the same as that in Figure 19, so a description thereof will be omitted. The waveforms of the current commands IRe1 and IRe2 when the branch lateral force and thrust are controlled simultaneously are gentler than those in Figure 19. The reason for the gentler waveforms is that the coils to be energized are smoothly switched using the concept of assumed resistance. Comparing Figure 19 and Figure 20, it is easier to control the current in Figure 20. Therefore, it can be said that the method disclosed herein is suitable for driving multiple movers in close proximity near the branch paths of a linear synchronous motor.

[0137] In addition, near a branching point, the track shapes on the first and second track sides are not necessarily symmetrical, so there are cases where the coil resistance values ​​differ between the first and second track sides. In such cases, the actual resistance value can be used instead of the assumed resistance value to calculate the current command.

[0138] The formula for calculating the current command based on the true resistance value is as follows: (15) When the current command is calculated using this formula (15), the copper loss of the motor can be reduced, resulting in energy savings.

[0139]

[0140] Figure 21 shows an example of the configuration of the control calculation means 7a and 7b2 of embodiment 7. Figure 21 shows the control system for the mover 1a2. Although not shown in the figure, the control system for the mover 1b2 has a similar configuration. Furthermore, when three or more movers are moved simultaneously, similar control systems are prepared for the number of movers.

[0141] The differences from the sixth embodiment are the thrust coefficient and branch lateral force coefficient reference means 22a, the branch control means 29, and the current command calculation means 23a. The thrust coefficient and branch lateral force coefficient reference means 22a is used to reference the thrust coefficient and branch lateral force coefficient based on the mover position. The branch control means 29 determines the branch lateral force command based on the path selection command for the mover 1a2. The current command calculation means 23a calculates the current command based on the above-mentioned equation (12) or equation (13).

[0142] The assumed resistance value used in the calculation of the current command may be set in the same way as explained in Figures 12A and 12B. In other words, when two or more movers are driven in close proximity, prohibiting the flow of current to coils that affect the thrust and branch lateral force of the two movers simplifies the calculation of the current command, so the virtual conductance of those coils is set to zero. On the other hand, when they are not driven in close proximity, the virtual conductance is adjusted to maximize the number of current-carrying coils in order to reduce copper loss. Note that the virtual conductance may be changed depending on the distance between the movers.

[0143] By configuring the control calculation means 7a and 7b2 in this way, it is possible to easily calculate the current command for the linear synchronous motor including the branch path. Moreover, it is also easy to drive a plurality of movers in close proximity to each other.

[0144] Embodiment 8. The linear synchronous motors described up to this point are mechanically constrained by mechanical guide rails or wheels (not shown). However, in a linear synchronous motor including a branch path as described in embodiment 7, the method of mechanically constraining the branch section becomes an issue. Naturally, it is necessary to design the mechanical structure so that the mover does not accidentally fall off or run off the wheel at the branch section.

[0145] If excessive torque is applied to the mover at the branch, there is a risk that the mover may come off or derail. However, even if it is a countermeasure, making the mechanical structure more complicated than necessary leads to increased costs, and is therefore undesirable. Therefore, in embodiment 8, in order to prevent the mechanical structure from becoming complicated, we consider a method of calculating a current command that does not apply excessive torque to the mover.

[0146] FIG. 22 is a diagram for explaining torque related to a linear synchronous motor. Here, we consider the torque that an arbitrary coil U1 applies to the mover. The U1 coil is assumed to be in the first quadrant when viewed from the center of rotation of the mover. The distance between the center of rotation of the mover and the U1 coil is defined as r xyu1 , the angle between the x-axis, the center of rotation, and the U1 coil is θ zu1 By placing these variables, the torque τ around the z-axis generated by the U1 coil is θzu1 can be written as the following equation (16).

[0147]

[0148] However, F xy1 : Thrust generated by U1 coil, F yu1 : The lateral force generated by the U1 coil. In other words, the torque around the z-axis generated by the U1 coil can be calculated from the position of the U1 coil seen from the center of gravity of the rotor and the thrust and lateral force generated by the U1 coil.

[0149] The thrust and lateral force generated by the U1 coil can be expressed by the thrust coefficient or the product of the branch lateral force coefficient and the current, respectively. The thrust coefficient of the U1 coil is k xu1 , the branch lateral force coefficient is k yu1 , current i u1 The torque τ around the z-axis generated by the U1 coil is θzu1 can be written as the following equation (17).

[0150]

[0151] However, k θzu1 is the torque coefficient around the z-axis, and is expressed by the following equation (18). Note that the torques generated by other coils can also be expressed by similar equations.

[0152]

[0153] Therefore, as in the seventh embodiment, we consider the current commands for six coils near the mover (in FIG. 16, the coil (not shown) to the left of coil 3a and five coils 3a to 3e). Here, the torque coefficient of each coil is k θz1 ~k θz6 , the torque around the z-axis is τ θz The torque around the z-axis is expressed as τ θz can be expressed as the following equation (19) using the sum of the products of the torque coefficient and the current.

[0154]

[0155] Since the equations for thrust, branch lateral force, and torque are known, the current command can be calculated by solving these three equations simultaneously. A pseudo-inverse matrix may be used to solve the simultaneous equations, but since the method using a pseudo-inverse matrix has the drawbacks described in the seventh embodiment, a method based on assumed resistance values ​​and virtual copper loss is used. The distribution equation for the current command that minimizes the virtual copper loss can be expressed as shown in the following equation (20). However, in the following equation (20), τ θz * is the torque command around the z-axis. Here, in order to prevent the mover from accidentally dropping off or running off, τ θz * It is desirable to set it to 0.

[0156]

[0157] FIG. 23 shows an example of the configuration of the control calculation means of embodiment 8. The difference from embodiment 7 lies in the thrust coefficient, branch lateral force coefficient, and torque coefficient reference means 22b and current command calculation means 23b. The thrust coefficient, branch lateral force coefficient, and torque coefficient reference means 22b calculates the thrust coefficient, branch lateral force coefficient, and torque coefficient of each coil based on the mover position. The current command calculation means 23b then calculates equation (18) based on these parameters and the assumed resistance value to determine the current command. Note that, although the formula is omitted, the calculation may be performed using the actual resistance value instead of the assumed resistance value.

[0158] By configuring a linear synchronous motor including a branch path in this way, it is possible to prevent the mover from falling off or running off without making the mechanical structure more complicated than necessary. Another advantage is that the calculation of the current command does not need to be more complicated than necessary.

[0159] Embodiment 9. There are various types of linear synchronous motors, and even linear synchronous motors without branch paths can have a structure that allows for motion with multiple degrees of freedom. The type of levitated mover coil linear synchronous motor proposed in Patent Document 5 is capable of translational motion in the x-axis direction as well as translational motion in the y-axis direction and rotational motion around the z-axis.

[0160] Although the motors described in the first to eighth embodiments and the levitated mover coil linear synchronous motor of the type proposed in Patent Document 5 are significantly different in structure, they share common challenges. Specifically, these common challenges include the need for fault tolerance, i.e., the ability to continue operation even if some of the coils fail (in extreme cases, the coils burn out) or if the power supply is broken; the need to dissipate heat generated by the motor; and the possibility of moving multiple movers in close proximity. Furthermore, if two or more types of coils with different resistance values ​​could be used in combination when designing this type of motor, the degree of freedom in designing the motor structure could be greatly increased.

[0161] It is believed that the current command calculation methods described in the first to eighth embodiments will be useful in solving these problems. In this regard, a control method will be described below using the levitated mover coil type linear synchronous motor of the ninth embodiment as an example. This control method makes it possible to deal with cases where the degree of freedom of movement is high in a moving body or a rotating body (both collectively referred to as a driven body).

[0162] 24 shows an example of the configuration of a motor according to embodiment 9. The mover 1 is levitated by a levitation device (not shown). The method of configuring the levitation device is not limited here. The levitation device may be a pneumatic type that uses air injection, or a magnetic type that levitates using magnetic force. Other methods of levitation are also acceptable.

[0163] A plurality of coils 3a to 3l are attached to the mover 1, and the currents in these coils are controlled by coil current control means 5a to 5l. A large number of permanent magnets 11a and 11b are arranged in the stator 2, and these permanent magnets are attached to a back yoke 10c. A large number of mover position detection means and mover speed detection means 6 are arranged to observe the position or attitude of the mover.

[0164] The position data or velocity data of the mover detected by the multiple mover position detection means and mover velocity detection means 6 is processed in control calculation means 7a and 7b and converted into values ​​suitable for control (for example, the position or angle based on the center of gravity of the mover is calculated). The control calculation means 7a and 7b operate based on the x-direction position command, y-direction position command, tilt angle command about the z-axis, and the mover position data and attitude data, and determine the current command for each coil.

[0165] Although the control and calculation means 7a and 7b are arranged as two separate units due to space limitations, they do not necessarily have to be separated into two units. They may be configured on a single CPU or on separate CPUs.

[0166] Here, we will explain an example of a motor in which the electrical angle per coil is 120 degrees. This is a magnetic pole arrangement that is suitable for driving the coils with a three-phase connection using a three-phase inverter power supply. However, this is just one example, and motors other than three-phase motors can also be considered.

[0167] The command value of the current flowing through the 12 coils installed in the mover 1 is i * ~i 12 * , the thrust coefficient of each coil in the x direction is k x1 ~k x12 , the thrust coefficient in the y direction is k y1 ~k y12 , the torque coefficient around the z-axis is k θz1 ~k θz12 , the assumed resistance value of each coil is R ν1 ~R ν12 , the thrust command in the x direction is F x * , the thrust command in the y direction is F y * , the torque command around the z-axis is τ θz * When this is done, the desired F x * , F y * , τ θz * The current command distribution that minimizes the virtual copper loss while satisfying the above can be written as the following equation (21).

[0168]

[0169] k x1 ~k x12 , k y1 ~k y12 and k θz1 ~k θz12 are all functions of the mover position. θz1 ~k θz12 The concept of this may be the same as that explained in FIG. 22. Although the calculation formula is omitted, the assumed resistance value R ν1 ~R ν12 The true resistance value may be used instead to calculate the current command.

[0170] Alternatively, the current command may be calculated in a coordinate system after three-phase to two-phase transformation or rotational coordinate transformation. For example, if 12 coils are divided into four groups and each group is three-phase connected, four groups of three-phase coils are created. Each of these four groups of coils may be subjected to three-phase to two-phase transformation or rotational coordinate transformation. For example, the following T d3 (θ e ) matrix is ​​well known (see equation (22)). e : Magnetic pole position.

[0171]

[0172] Using this, it is assumed that the dq axis current and the current of each coil can be expressed as in the following equation (23).

[0173]

[0174] Here, the relationship between the thrust in the x-axis direction, the thrust in the y-axis direction, the torque around the z-axis, and the current can be expressed as in the following equation (24). Here, by performing the above-mentioned coordinate transformation on the current, the equation can be transformed as shown in the following equation (25). However, for each proportionality coefficient in the last equation of equation (25), the following equation (26) holds.

[0175]

[0176]

[0177]

[0178] Here, k xd1 , ..., k xd4 : proportionality coefficient between the d-axis current of each coil group and the x-axis thrust, k xq1 , ..., k xq4 : proportionality coefficient between the q-axis current of each coil group and the x-axis thrust, k yd1 , ..., k yd4 : proportional coefficient between the q-axis current of each coil group and the y-axis thrust, k yq1 , ..., k yq4 : proportional coefficient between the q-axis current of each coil group and the y-axis thrust, k θzd1 , ..., k θzd4 : proportional coefficient between the d-axis current of each coil group and the torque around the z-axis, k θzq1 , ..., k θzq4 : The proportionality coefficient between the q-axis current of each coil group and the torque around the z-axis.

[0179] When the magnetization of a permanent magnet is sinusoidal, k xd1 , ..., k xd4 and k yq1 , ..., k yq4 are all zero. xq1 , ..., k xq4 and k yd1 , ..., k yd4 is a non-zero constant. θzd1 , ..., k θzd4 and k θzq1 , ..., k θzq4 is a function of the mover position.

[0180] When control is performed on the three-phase coordinate system, k x1 ~k x12 , k y1 ~k y12 , and k θz1 ~k θz12 were all functions of the mover position, but when transformation is performed on the dq coordinate, many parameters can be treated as constants. This simplifies control calculations, so there is an advantage to controlling such motors on the dq coordinate.

[0181] The true resistance of the four sets of three-phase connected coils is R dq1 ~R dq4 When expressed in terms of variables, the desired F x *, F y * , τ θz * The current command distribution that minimizes the sum of the copper losses generated in the four coils while satisfying the above can be written as the following equation (27). However, the matrix K RT and K. dq is expressed as in equation (28). RT The superscript T on the right side of the equation denotes a transposed matrix. Although the mathematical formula is omitted, even when controlling using dq coordinates, calculations may be performed using an assumed resistance value instead of the true resistance value.

[0182]

[0183]

[0184] 25 shows an example of the configuration of the control calculation means of embodiment 9. The control calculation means 7a and 7b are made up of a position / attitude control calculation means 21a, a thrust coefficient and branch lateral force coefficient and torque coefficient reference means 22b, and a current command calculation means 23b.

[0185] The position and attitude control calculation means 21a performs position control and attitude control so that the x-direction position of the mover, the y-direction position, and the tilt angle around the z-axis (attitude of the mover) are set to desired values, and outputs an x-direction thrust command F x * , y-direction thrust command F y * , torque command τ around the z-axis θz * The desired F x * , F y * , τ θz * If there is an actuator that can realize this, the x-direction position control, the y-direction position control, and the attitude control around the z-axis can be considered separately.

[0186] Therefore, the position / attitude control calculation means 21a performs an x-axis position control calculation, a y-axis position control calculation, and an attitude control calculation about the z-axis. Each control calculation may be performed by any method, but known methods include feedback control, feedforward control, and two-degree-of-freedom control, which is a combination of these two. A known feedback control method is a method using PID control.

[0187] The thrust coefficient and branch lateral force coefficient and torque coefficient reference means 22b calculates the thrust coefficient k in the x direction of each coil based on the x-direction position of the mover, the y-direction position, and the tilt angle around the z-axis (the attitude of the mover). x1 ~k x12 , y-direction thrust coefficient k y1 ~k y12 , torque coefficient k around the z axis θz1 ~k θz12 When a rotational coordinate transformation is performed, each coefficient after the dq coordinate transformation may be output.

[0188] The current command calculation means 23b calculates the current command based on the above-mentioned formula. It differs from the prior art in that it calculates the current command using the true resistance value or the assumed resistance value of each coil. Whether to use the true resistance value or the assumed resistance value may be changed depending on the purpose at the time.

[0189] Configuring a motor in this way has the following advantages: - It is easy to continue operation even if some of the coils fail. - The amount of current to coils that have become hot can be easily reduced, so heat generated by the motor can be dispersed. - When moving multiple movers that are close to each other, the current command for the coils that affect the movement of two movers can be easily set to zero, reducing the amount of control calculations. - When designing this type of motor, two or more types of coils with different resistance values ​​can be used together, greatly increasing the degree of freedom in designing the motor structure.

[0190] Embodiment 10. In embodiment 10, a magnetic levitation device is considered. In semiconductor manufacturing equipment and the like, dust generated by wear of sliding parts is undesirable, so levitation devices without sliding parts are preferred. There are various possible methods for configuring a levitation device, but here we consider a magnetic levitation device.

[0191] Figure 26 shows an example of the configuration of a motor according to embodiment 10. This controls the levitation height of levitation stage 30, as well as the tilt angle around the x-axis and the tilt angle around the y-axis. Four movers 31 of voice coil motors are attached to this levitation stage 30. Each voice coil motor is composed of a mover 31 and a stator 32. Each mover 31 contains coils 3a to 3d, and the currents therein are controlled by coil current control means 5a to 5d.

[0192] By changing the current in each of the coils described above, it is possible to change the magnetic attractive force and magnetic repulsive force of each voice coil motor, thereby manipulating the movement of the levitation stage 30. Note that although the number of voice coil motors is four here, five or more may also be used.

[0193] Incidentally, reducing the weight of the levitation-type stage 30 has various advantages, such as energy savings and improved positioning response speed, etc. However, reducing the weight of the levitation-type stage 30 can cause deformation and vibration of the levitation-type stage 30 when it is moved.

[0194] For this reason, recent reports have shown that the redundant degrees of freedom of the precision stage are used to perform torsion control (control to prevent twisting) of the floating stage 30. When torsion control is required or when fault-tolerant performance needs to be improved, five or more voice coil motors may be used.

[0195] Note that these voice coil motors do not necessarily have to be the same. In other words, different types of voice coil motors with different resistance values ​​or repulsive force coefficients (here, different types means different electrical properties such as electrical resistance or induced voltage constant) may be used in combination. For example, a large voice coil motor for levitation and a small voice coil motor for torsion control may be used in combination. Using different types of electromagnetic actuators makes it possible to respond to changing trends in linear motor structures.

[0196] The mover 31 of the voice coil motor is composed of coils 3a to 3d, a permanent magnet 33, and an iron core 35. When the current flowing through coil 3a is zero, the stator 32 is attracted to the mover 31, which includes the permanent magnet 33. Here, by manipulating the current flowing through coil 3a, it is possible to cancel out or strengthen the magnetic flux inside the mover 31, thereby changing the attractive force between the mover 31 and the stator 32. When the attractive force and gravity are well balanced, the levitation stage 30 can be magnetically levitated.

[0197] It is known that the force required for magnetic levitation is small if the stator is placed on the upper side and the mover is suspended. Therefore, suspended magnetic levitation is considered here, but this is merely one example. As will be explained in the eleventh embodiment and onwards, it is also possible to consider a magnetic levitation device in which the stator is placed on the lower side and the mover is pushed up.

[0198] A large number of mover position detection means and mover velocity detection means 6 are provided to observe the levitation height or attitude of the levitation stage 30. Position data or velocity data of the levitation stage 30 detected by the large number of mover position detection means and mover velocity detection means 6 is processed in control calculation means 7a and 7b and converted into values ​​suitable for control as appropriate (for example, the position or angle relative to the center of gravity of the levitation stage 30 is calculated).

[0199] Here, the above-mentioned control and calculation means 7a and 7b operate based on the z-direction position command, the tilt angle command around the x-axis, the tilt angle command around the y-axis, and the position data and attitude data of the levitation stage 30, and determine the current command for each coil.

[0200] Although the control and calculation means 7a and 7b are arranged in two parts due to space limitations, they do not necessarily have to be divided into two parts. They may be configured on a single CPU or on separate CPUs. Here, the repulsive force F of each voice coil motor is za1 ~F za4 and the current I of each coil a1 ~I a4 The relationship can be written as the following equation (29): Ra1 ~K Ra4 is the repulsion coefficient.

[0201]

[0202] When four voice coil motors are evenly arranged with reference to the center of gravity of the floating stage 30 as shown in FIG. 26, the repulsive force F generated by each voice coil motor is za1 ~F za4 and the lift force F at the center of gravity zc , torque τ around the x-axis θx , torque τ around the y-axis θy The relationship can be written as the following equation (30).

[0203]

[0204] However, L xa , L ya are the distances between the voice coil motors in the x-axis and y-axis directions, respectively, and K z1 ~K z4 : Lifting force at the center of gravity F zc and the current I of each coil a1 ~I a4 The proportionality coefficient of K θx1 ~K θx4 : Torque around the x-axis τ θx and the current I of each coil a1 ~I a4 The proportionality coefficient of K θy1 ~K θy4 : Torque around the y-axis τθy and the current I of each coil a1 ~I a4 Furthermore, the proportionality coefficient in the lowest equation of equation (30) satisfies equation (31).

[0205]

[0206] To control the floating stage 30, za1 ~F za4 It is necessary to solve the above simultaneous equations for the desired F zc , τ θx , τ θy To realize I a1 ~I a4 There are countless combinations of R. Therefore, some constraints are added, and in this disclosure, the constraints are considered to be minimizing the true copper loss or virtual copper loss. a1 ~R a4 When expressed as the symbol, the lift force command F zc * , torque command τ around the x-axis θx * , torque command τ around the y-axis θy * The current command distribution that satisfies the above and minimizes copper loss can be written as the following equation (32): Although the equation is omitted, calculation may be performed using an assumed resistance value instead of the true resistance value.

[0207]

[0208] Fig. 27 shows an example of the configuration of the control calculation means of embodiment 10. The control calculation means 7a and 7b are composed of a position / attitude control calculation means 21b, a repulsive force coefficient reference means 22c, and a current command calculation means 23b. Except for the difference in the direction of movement, Fig. 27 has almost the same configuration as Fig. 25, so a detailed description will be omitted.

[0209] Configuring a motor in this way has the following advantages: - It is easy to continue operation even if one of the coils fails. - The amount of current to a coil that has become hot can be easily reduced, so heat generated by the motor can be dispersed. - When designing this type of motor, two or more types of coils with different resistance values ​​can be used together, greatly increasing the degree of freedom in designing the motor structure. This advantage is particularly significant when performing torsion control.

[0210] Eleventh Embodiment In the ninth embodiment, a levitation-type linear synchronous motor was considered, and in the tenth embodiment, a levitation-type stage was considered, but in the eleventh embodiment, a magnetic levitation-type planar motor is considered.

[0211] Figure 28 shows the configuration of a magnetically levitated planar motor according to embodiment 11. Recently, magnetically levitated planar motors capable of six degrees of freedom of motion have been put to practical use. While a mover coil type planar motor is discussed here, a similar control method can also be considered for a mover magnet type planar motor.

[0212] The magnetically levitated planar motor in Figure 28 shows a mover 1 made up of multiple coils 3a-3l levitating and running on a stator 2 with permanent magnets arranged in a grid pattern (the mover in this case is also called a magnetically levitated stage). This motor is capable of translational motion in the x-axis direction, translational motion in the y-axis direction, levitation motion in the z-axis direction (motion in which the levitation direction is along the z-axis), and rotational motion around the x-, y-, and z-axes, allowing for six degrees of freedom of movement. However, because this motor has no sliding parts, it is beginning to be used in clean environments where dust is undesirable.

[0213] 29 shows an example of the configuration of a motor according to embodiment 11. The currents in coils 3a to 3l are controlled by coil current control means 5a to 5l. A large number of mover position detection means and mover speed detection means 6 are provided to observe the position or attitude of the mover 1. Position data or speed data of the mover 1 detected by the large number of mover position detection means and mover speed detection means 6 is processed in control calculation means 7a and 7b and converted into values ​​suitable for control as appropriate (for example, the position or angle relative to the center of gravity of the mover 1 is calculated).

[0214] The control and calculation means 7a and 7b operate based on x-, y-, and z-direction position commands, tilt angle commands about the x-, y-, and z-axes, and position data and attitude data of the mover 1, and determine the current command for each coil. Note that although a planar motor having 12 coils is considered here, the number of coils may be changed as appropriate.

[0215] In addition, although the twelve coils are divided into four groups and each group is connected in three phases, the three phase connection is not necessarily required, and each coil may be controlled by a single phase power supply.

[0216] Incidentally, reducing the weight of this type of planar motor has various advantages, such as energy savings and improved positioning response speed. However, reducing the weight of the mover 1 can cause the mover 1 to deform and vibrate when moved. For this reason, recent reports have shown that redundant degrees of freedom are used to perform torsion control (control to prevent twisting) of the mover 1. If torsion control or improved fault-tolerance performance is desired, 13 or more coils may be used.

[0217] The method of motion control of the planar motor will be explained. Let the current flowing through any coil U1 be i u1 Then, the forces in the x, y, and z directions generated by the coil can be written as the following equation (33).

[0218]

[0219] However, k xu1 , k yu1, k zu1 are the thrust coefficients in the x-axis, y-axis, and z-axis directions, respectively. xu1 , k yu1 , k zu1 is a function of the position or orientation of the mover, and its value changes depending on the position or orientation of the mover. In order to express the torque generated when current is passed through the U1 coil in a mathematical formula, it is convenient to express the position of the U1 coil as viewed from the center of gravity of the mover in polar coordinate format on each plane.

[0220] The position of the U1 coil as seen from the center of gravity of the mover is (x u1 , y u1 , z u1 ), the angle between this coil and the center of gravity of the mover in the xy plane, yz plane, and zx plane can be written as the following equation (34). Also, the distance between this coil and the center of gravity of the mover in the xy plane, yz plane, and zx plane can be written as the following equation (35). Here, r xyu1 , r yzu1 , r zxu1 correspond to the lengths of the moment arms for rotation around the x-axis, y-axis, and z-axis, respectively. zu1 , θ xu1 , θ yu1 is the angle between the U1 coil and the center of gravity of the mover in each plane. Furthermore, the torque τ θzu1 can be written as shown in the following equation (36), similarly to the explanation given in FIG.

[0221]

[0222]

[0223]

[0224] However, k θuz1 is the torque coefficient around the z-axis of the U1 coil, and k θuz1 =r xyu1 (-k xu1 sinθ zu1 +k yu1 cosθ zu1 ) k θuz1is a function of the position or orientation of the mover, and changes when the position or orientation of the mover changes. Similarly, when torque around the y-axis is considered, it can be written as the following equation (37).

[0225]

[0226] However, k θyu1 is the torque coefficient around the y-axis of the U1 coil, and k θyu1 =r zxu1 (k xu1 sinθ yu1 -k zu1 cosθ yu1 ) k θyu1 is a function of the position or orientation of the mover, and changes when the position or orientation of the mover changes. If we consider the other coils in a similar manner, the forces in the x, y, and z directions that each coil applies to the mover, as well as the torques around the x, y, and z axes, can be expressed mathematically. Therefore, the forces in the x, y, and z directions that the 12 coils apply to the mover, as well as the torques around the x, y, and z axes, can be expressed as in the following equation (38).

[0227]

[0228] where the thrust coefficient of each coil in the x direction is k x1 ~k x12 , y-direction thrust coefficient: k y1 ~k y12 : , thrust coefficient in the z direction: k z1 ~k z12 , torque coefficient around the x-axis: k θx1 ~k θx12 , torque coefficient around the y-axis: k θy1 ~k θy12 , torque coefficient around the z axis: k θz1 ~k θz12 , current of each coil: i1 to i 12 is.

[0229] The forces in the x, y, and z directions that the 12 coils apply to the mover, as well as the torques around the x, y, and z axes, may be expressed in a different coordinate system. For example, the following coordinate transformation matrix from the stationary two-phase coordinate system to the three-phase coordinate system may be used: 23The matrix (see equation (39)) is well known.

[0230]

[0231] Using this, it is assumed that the static two-phase currents (αβ axis currents) of each coil group and the currents of each coil can be expressed as shown in the following equation (40).

[0232]

[0233] Using the stationary two-phase currents (αβ-axis currents) of each coil group, the forces in the x, y, and z directions, and the torques around the x-axis, y-axis, and z-axis can be formulated as shown in the following equation (41).

[0234]

[0235] However, k xα1 ~k xα4 :Proportional coefficient between the α-axis current of each coil group and the x-direction thrust force, k xβ1 ~k xβ4 :Proportional coefficient between β-axis current and x-direction thrust of each coil group, k yα1 ~k yα4 :Proportional coefficient between the α-axis current of each coil group and the y-direction thrust force, k yβ1 ~k yβ4 : proportionality coefficient between the β-axis current of each coil group and the y-direction thrust, k zα1 ~k zα4 :Proportional coefficient between the α-axis current of each coil group and the z-direction thrust, k zβ1 ~k zβ4 : proportionality coefficient between the β-axis current of each coil group and the z-direction thrust, k θxα1 ~k θxα4 :Proportional coefficient between the α-axis current of each coil group and the torque around the x-axis, k θxβ1 ~k θxβ4 :Proportional coefficient between the β-axis current of each coil group and the torque around the x-axis, k θyα1 ~k θyα4 :Proportional coefficient of the α-axis current of each coil group and the torque around the y-axis, k θyβ1 ~k θyβ4 :Proportional coefficient of the β-axis current of each coil group and the torque around the y-axis, k θzα1 ~k θzα4 :Proportional coefficient of the α-axis current of each coil group and the torque around the z-axis, k θzβ1 ~k θzβ4: The proportionality coefficient between the β-axis current of each coil group and the torque around the z-axis. All of these proportionality coefficients can be expressed as a function of the position or posture of the mover.

[0236] In this way, the relationship between the force, torque, and current of a planar motor can be expressed as in Equation (38) or Equation (41). Alternatively, it can be expressed using other coordinate transformations. By treating these equations as simultaneous equations and solving for the current, the current command for the planar motor can be obtained. However, because there are more variables for the current than for the degrees of freedom of motion, there are an infinite number of current commands that satisfy the desired force and torque. Therefore, in this disclosure, the current command is determined so as to minimize the copper loss or virtual copper loss of the planar motor.

[0237] The command value of the current flowing through the 12 coils installed in the mover 1 is i * ~i 12 * , the resistance of each coil is R1 to R 12 , the thrust commands in the x, y, and z directions are respectively x * , F y * , F z * The torque commands around the x-axis, y-axis, and z-axis are respectively τ θx * , τ θy * , τ θz * Then, the current command distribution that minimizes copper loss while satisfying the desired thrust command and torque command can be written as the following equation (42). R , K. C etc. is as shown in the following equation (43).

[0238]

[0239]

[0240] Although the calculation formula is omitted, the true resistance values ​​R1 to R 12 Instead, the assumed resistance value R ν1 ~R ν12Alternatively, the current command may be calculated using the current in the stationary two-phase coordinate system as shown in the following equation (44). However, details of each equation in equation (44) are as shown in the following equations (45) to (47). Note that R αβ1 ~R αβ4 is the resistance value of each coil group in the stationary two-phase coordinate system.

[0241]

[0242]

[0243]

[0244]

[0245] Although the calculation formula is omitted, the true resistance values ​​R1 to R2 are also calculated in the static two-phase coordinate system. 12 Instead of the assumed resistance value R ν1 ~R ν12 Furthermore, the current commands may be calculated in other coordinate systems (for example, a dq rotating coordinate system) than the stationary two-phase coordinate system.

[0246] 30 shows an example of the configuration of the control calculation means of embodiment 11. The control calculation means 7a and 7b of embodiment 11 have a position / attitude control calculation means 21a, a torque coefficient reference means 22b, and a current command calculation means 23b.

[0247] The position / attitude control calculation means 21a controls the position and tilt angle of each axis so that the mover is at the desired position and attitude, and outputs thrust and torque commands for each axis. Any method of position and attitude control may be used, but a method using PID control is well known, for example. The torque coefficient reference means 22b calculates the thrust coefficient in each axial direction and the torque coefficient around each axis for each coil based on the position or attitude of the mover.

[0248] The current command calculation means 23b calculates a current command based on the resistance value of each coil, or the thrust command, torque command, and each proportionality coefficient of each axis. This differs from the prior art in that the current command is calculated using the true resistance value or the assumed resistance value of each coil. Whether the true resistance value or the assumed resistance value is used may be changed depending on the purpose at the time.

[0249] By configuring the control calculation means 7a and 7b in this way, it is possible to determine a current command that minimizes the copper loss or virtual copper loss of the planar motor.

[0250] Configuring a motor in this way has the following advantages: - It is easy to continue operation even if some of the coils fail. - The amount of current to coils that have become hot can be easily reduced, so heat generated by the motor can be dispersed. - When moving multiple movers that are close to each other, the current command for the coils that affect the movement of two movers can be easily set to zero, reducing the amount of control calculations. - When designing this type of motor, two or more types of coils with different resistance values ​​can be used together, greatly increasing the degree of freedom in designing the motor structure.

[0251] Twelfth Embodiment Up to the eleventh embodiment, we have considered the control of a linear motor or a planar motor, but the control theory described up to this point can of course also be applied to a rotary motor. Therefore, in the twelfth embodiment, we will describe a control method for a rotary bearingless motor.

[0252] Recently, rotating machines that use magnetic bearings have been put into practical use. Furthermore, there has been an increase in the number of practical examples of "bearingless motors" that integrate a magnetic bearing with a motor. Many bearingless motors have both a motor winding and a support winding, but there are also motors in which the motor winding and support winding are integrated (for example, the motor winding described in JP 2014-241725 A).

[0253] Here, we will describe a bearingless motor of a type in which the motor windings and support windings are integrated, but the technology disclosed herein can also be applied to motors of a type in which the motor windings and support windings are separate.

[0254] Figures 31A, 31B, and 31C are configuration diagrams of a bearingless motor according to embodiment 12. Figure 31A shows the cross-sectional shape of the motor. Magnets are arranged in the mover 1 (here, the mover is the rotor; the same applies below) so that six magnetic poles are formed. The stator 2 has nine coils 3a to 3i, and the movement of the mover 1 is controlled by passing current through these coils. The current flowing through the coils 3a to 3i is controlled by coil current control means 5a to 5i, which will be described later. The coils 3a to 3i may be divided into several groups and three-phase connected, but it is necessary to ensure that the degree of freedom of control is greater than the degree of freedom of movement.

[0255] Here we will discuss a 6-pole, 9-slot rotating machine, but the number of magnetic poles or the number of coil slots is by no means limited to this. Basically, the degree of freedom of control must be greater than the degree of freedom of movement, but the number of pole slots of the motor can be freely selected.

[0256] In a rotating machine using a normal mechanical bearing, θ z Although only rotational motion around the axis is possible, this motor has no mechanical constraints on the mover 1 (rotor 1), so the z Not only rotational movement around the axis but also translational movement along the x and y axes is possible.

[0257] 31B and 31C show the cross section of this bearingless motor when viewed from a different angle. This bearingless motor is supported in a horizontal position (also called a horizontal position) by (control by) a thrust magnetic bearing (not shown), and is configured to minimize movement in the z-axis direction (also called a vertical position).

[0258] When one stage of this bearingless motor is used, as in Figure 31B, three degrees of freedom of movement are possible. When two stages of this bearingless motor are used, as in Figure 31C, tilting around the x-axis or y-axis is also possible, making five degrees of freedom of movement possible. Regardless of which configuration is used, the method of calculating the current command does not change much, so here we will consider the configuration in Figure 31C.

[0259] In order to observe the rotation angle or position of the mover 1, a plurality of mover position detecting means and mover speed detecting means 6 are disposed inside the bearingless motor. Position data or speed data of the mover 1 detected by the many mover position detecting means and mover speed detecting means 6 is processed in control calculation means 7a and 7b and converted into values ​​suitable for control (for example, the position or angle of the center of gravity of the mover 1 as viewed from the position of each coil is calculated).

[0260] 32 shows an example of the configuration of a motor according to embodiment 12. Control calculation means 7 operates based on a speed command about the z-axis, a position command in the x and y directions, and a tilt angle command about the x and y axes, and determines a current command for each coil. The current flowing through each coil is controlled by coil current control means 5a to 5i.

[0261] Here, we consider a two-stage bearingless motor as shown in Figure 30(c), so there are two sets of coil current control means 5a to 5i. For the sake of observing the position and attitude of the mover, Figure 31 also shows multiple mover position detection means and mover speed detection means 6. In order to control a bearingless motor, it is necessary to understand the mechanics and electromagnetism of this motor, so these will be explained in detail.

[0262] 33 is a diagram for explaining the forces acting on a bearingless motor by breaking down the acting forces. Here, as an example, the breakdown of the force generated by the coil 3b is explained. The current flowing through the coil 3b causes F ν1 Let's say that a vector force of F ν1 can be decomposed into vectors in other directions. For example, the radial force (r-axis direction) F rν1 and the force F in the circumferential direction (c-axis) cν1 and the force in the x-axis direction F xν1 and the force F in the y-axis direction yν1 Here, the radial force (r-axis direction) F generated by the coil 3b is rν1 and the force F in the circumferential direction (c-axis) cν1 and the current i flowing through the coil 3b ν1The relationship can be expressed as the following equation (48).

[0263]

[0264] However, Φ aν1 : Number of interlinkage magnetic fluxes of coil 3b, l ν1 : distance from the center of the mover to the coil 3b, θ e : magnetic pole position (moving element angle), k rν1 : proportionality coefficient between the current in the coil 3b and the radial force, k cν1 : is the proportionality coefficient between the current in the coil 3b and the force in the circumferential direction. rν1、 k cν1 The relationship between the magnetic pole position and the like is shown in the following equation (49). Furthermore, the torque around the z-axis can be written as the following equation (50).

[0265]

[0266]

[0267] However, k θzν1 : k is the proportionality coefficient between the current in coil 3b and the torque around the z-axis rν1 , k cν1 , k θzν1 are all magnetic pole positions θ e It is a function of k rν1 , k cν1 , k θzν1 Is, l ν1、 or Φ aν1 It changes depending on

[0268] Coil 3b is located at a position offset by 2π / 9 from the x-axis. The radial force F rν1 and the force F in the circumferential direction (c-axis) cν1 is the force F in the x-axis direction. xν1 and the force F in the y-axis direction yν1 To convert this to the above, the rotation matrix shown in the following equation (51) can be used: where δ is an arbitrary angle.

[0269]

[0270] Using the rotation matrix, the force F in the x-axis direction exerted by coil 3b is xν1and the force F in the y-axis direction yν1 can be expressed as the following equation (52). However, in equation (52), equation (53) holds. Here, k xν1 : proportionality coefficient between the current in coil 3b and the force in the x-axis direction, k yν1 : The proportionality coefficient between the current in the coil 3b and the force in the y-axis direction. xν1 , k yν1 Also, the magnetic pole position θ e It is a function of k xν1 , k yν1 Also ν1 or Φ aν1 Therefore, the relationship between the current in coil 3b, the torque around the z-axis, the translational force in the x-direction, and the translational force in the y-direction can be written as in the following equation (54).

[0271]

[0272]

[0273]

[0274] By the way, when considering a two-stage bearingless motor as shown in FIG. 31C, F xν1 , or F yν1 When this occurs, rotational motion around the x-axis and rotational motion around the y-axis also occur simultaneously. In order to consider the rotational motion around the x-axis and rotational motion around the y-axis, we define the position of the coil 3b as seen from the center of gravity of the mover. The position of the coil 3b as seen from the center of gravity of the mover is defined as (x ν1 , y ν1 , z ν1 ), the angle between this coil and the center of gravity of the mover in the yz plane and the zx plane is expressed by the upper and lower equations of the following equation (55), respectively. Also, the distance between this coil and the center of gravity of the mover in the yz plane and the zx plane is expressed by the upper and lower equations of the following equation (56), respectively. Here, r yzν1、 r zxν1 corresponds to the length of the moment arm for rotation around the x and y axes. xν1 , θ yν1 is the angle formed by the coil 3b and the center of gravity of the mover in each plane.

[0275]

[0276]

[0277] The torque around the x-axis and the torque around the y-axis generated by the coil 3b can be expressed by the upper and lower equations of the following equation (57) using these parameters. However, in equation (57), the proportionality coefficient between torque and current is expressed by equation (58). Here, k θxν1 : proportionality coefficient between the current in coil 3b and the torque around the x-axis, k θyν1 : is the proportionality coefficient between the current in coil 3b and the torque around the y-axis. θxν1 , k θyν1 Also, the magnetic pole position θ e is a function of

[0278]

[0279]

[0280] Therefore, the current in coil 3b, the thrust force on each axis, and the torque around each axis can be expressed by the following equation (59). Note that similar equations can be derived for the other coils.

[0281]

[0282] Considering a two-stage bearingless motor as shown in Figure 31C, there are a total of 18 coils. When the forces and torques of all the coils are added up, the forces in the x and y directions and the torques around the x, y, and z axes can be expressed as in the following equation (60). However, the thrust coefficient of each coil in the x direction: k x1 ~k x18 , y-direction thrust coefficient: k y1 ~k y18 , torque coefficient around the x-axis: k θx1 ~k θx18 , torque coefficient around the y-axis: k θy1 ~k θy18 , torque coefficient around the z axis: k θz1 ~k θz18 , current of each coil: i1 to i 18 is.

[0283]

[0284] In the case of rotating machines, since there are many three-phase motors, here, six groups of three-phase connections are created by dividing the coils into groups of three. If a rotational dq transformation is performed on these three-phase connections, the forces in the x and y directions and the torques around the x, y, and z axes can be expressed as in the following equation (61).

[0285]

[0286] However, i d1 ~i d6 : d-axis current of each coil group, i q1 ~i q6 : q-axis current of each coil group, k xd1 ~k xd6 : proportionality coefficient between the d-axis current of each coil group and the x-direction thrust force, k xq1 ~k xq6 : proportionality coefficient between the q-axis current of each coil group and the x-direction thrust, k yd1 ~k yd6 : proportionality coefficient between the d-axis current of each coil group and the y-direction thrust force, k yq1 ~k yq6 : proportionality coefficient between the q-axis current of each coil group and the y-direction thrust, k θxd1 ~k θxd6 : proportionality coefficient between the d-axis current of each coil group and the torque around the x-axis, k θxq1 ~k θxq6 : proportionality coefficient between the q-axis current of each coil group and the torque around the x-axis, k θyd1 ~k θyd6 : proportionality coefficient between the d-axis current of each coil group and the torque around the y-axis, k θyq1 ~k θyq6 : proportionality coefficient between the q-axis current of each coil group and the torque around the y-axis, k θzd1 ~k θzd6 : proportionality coefficient between the d-axis current of each coil group and the torque around the z-axis, k θzq1 ~k θzq6 : proportionality coefficient between the q-axis current of each coil group and the torque around the z-axis,

[0287] Many of these proportionality coefficients can be expressed as a function of the position or posture of the mover. When the mover 1 has a surface magnet structure, k θzq1 ~k θzq4 From the magnetic pole position θ e Since the term k disappears, it can be considered as almost a constant.θzd1 ~k θzd4 From the magnetic pole position θ e The term disappears and can be approximated to approximately zero. On the other hand, the other proportional coefficients remain constant even after the rotational dq transformation. e The term remains.

[0288] The dq-axis currents of six coil groups can be manipulated (12 degrees of control freedom), while this bearingless motor has five degrees of freedom of movement, making it a redundant system. Therefore, there are an infinite number of combinations of current commands that satisfy the force commands in the x and y directions and the torque commands about the x, y, and z axes. In this disclosure, the current commands are determined so that the copper loss or virtual copper loss of the motor is minimized.

[0289] The six sets of dq axis current command values ​​are d1 * ~i d6 * , i q1 * ~i q6 * The resistance of each coil group in the dq coordinates is R dq1 ~R dq6 , x, and y directions thrust command F x * , F y * , the torque commands around the x-axis, y-axis, and z-axis are expressed as τ θx * , τ θy * , τ θz * Then, the current command distribution that minimizes copper loss while satisfying the desired thrust command and torque command can be written as in the following equation (62). However, details of each parameter are as in the following equations (63) and (64).

[0290]

[0291]

[0292]

[0293] Although the calculation formula is omitted, the true resistance values ​​R1 to R 12 Instead, the assumed resistance value R ν1 ~Rν12 Although the calculation formula is omitted here, the current command may be calculated by setting up simultaneous equations in the stationary two-phase coordinate system or another coordinate system.

[0294] 34 shows an example of the configuration of the control calculation means of embodiment 12. The control calculation means 7 of embodiment 12 has a position / attitude control calculation means 21a, a torque coefficient reference means 22b, and a current command calculation means 23b. The position / attitude control calculation means 21a performs position control and tilt angle control for each axis so that the mover is in the desired position and attitude, and outputs a thrust command and a torque command for each axis.

[0295] Any method for position control and attitude control may be used, but a method using PID control is well known. The proportionality coefficient reference means calculates the thrust coefficient in each axial direction and the torque coefficient around each axis for each coil based on the position or attitude of the mover. The current command calculation means 23b calculates a current command based on the resistance value of each coil, or the thrust command, torque command, and each proportionality coefficient for each axis. This differs from the prior art in that it calculates the current command using the true resistance value or assumed resistance value of each coil. Whether to use the true resistance value or assumed resistance value may be changed depending on the purpose at hand. By configuring the control calculation means 7 in this way, it is possible to determine a current command that minimizes the copper loss or virtual copper loss of the rotary bearingless motor.

[0296] Configuring a motor in this way has the following advantages: - It is easy to continue operation even if some of the coils fail. - The amount of current to coils that have become hot can be easily reduced, so the heat generated by the motor can be dispersed. - When designing this type of motor, two or more types of coils with different resistance values ​​can be used together, greatly increasing the degree of freedom in designing the motor structure. This means that, for example, if it is known in advance which coil groups will have a large load for supporting the shaft, it is possible to lower the resistance value of those coil groups.

[0297] 35 shows an example of a hardware configuration for realizing the present disclosure. The control and calculation means 7, 7a, and 7b according to embodiments 1 to 12 are realized by a processor 91, a memory 92, and a peripheral device 93. The control and calculation means 7, 7a, and 7b may be configured using one set of a processor 91, a memory 92, and a peripheral device 93, or the control and calculation means 7, 7a, and 7b may be configured using multiple sets of a processor 91, a memory 92, and a peripheral device 93. Furthermore, the coil current control means 5a, 5b, and 5c and the like may be configured inside the control and calculation means 7, 7a, and 7b.

[0298] The processor 91 is a CPU (Central Processing Unit, also called a central processing unit, processing unit, arithmetic unit, microprocessor, microcomputer, processor, or DSP (Digital Signal Processor)) or a system LSI (Large Scale Integration). Examples of the memory 92 include non-volatile or volatile semiconductor memories such as RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable Read Only Memory), and EEPROM (Electrically Erasable Programmable Read Only Memory). The memory 92 is not limited to these, and may also be a magnetic disk, optical disk, compact disk, minidisk, or DVD (Digital Versatile Disc). The peripheral device 93 may be, for example, a PWM pulse generating circuit, an analog-to-digital conversion circuit, or an encoder counter.

[0299] Incidentally, the PWM pulse generation circuit is used to drive the coil current control means 5a, 5b, 5c, 5d, 5e, 5f, 5g, 5h, etc. The analog-to-digital conversion circuit is used to detect the DC bus voltage or the motor phase current. The encoder counter is used to acquire mover position data.

[0300] As described above, this disclosure has proposed a method for calculating a copper loss minimizing current command and a virtual copper loss minimizing current command as a technology for solving all of the following problems related to motors of a type that combine a large number of electromagnetic actuators, such as linear motors, planar motors, and bearingless motors:

[0301] Although various exemplary embodiments and examples are described in this disclosure, the various features, aspects, and functions described in one or more embodiments are not limited to the application of a particular embodiment, but may be applied to the embodiments alone or in various combinations. Therefore, countless variations not illustrated are anticipated within the scope of the technology disclosed in this specification. For example, this includes cases where at least one component is modified, added, or omitted, or where at least one component is extracted and combined with components of another embodiment.

[0302] 1, 31 mover (rotor), 2, 32 stator, 3a to 3r coil, 4 core back, 5a to 5r coil current control means, 6 mover position detection means and mover speed detection means, 7, 7a, 7b control calculation means, 8 iron core module, 8a current detection means, 9a variable voltage source, 10, 10a, 10b, 10c back yoke, 11a to 11d, 33 permanent magnet, 12a current control calculation means, 21 position and speed control means, 21a, 21b position and attitude control calculation means, 22 thrust coefficient reference means, 23 current command value determination unit (current command calculation means), 24 fault detection means, 25 parameter correction means, 26 resistance calculation means, 27, 27a, 27b assumed resistance setting unit, 30 levitation stage, 35 iron core

Claims

1. A motor comprising: a plurality of electromagnetic actuators each having a different value of either an electrical resistance or an induced voltage constant; a moving body or a rotating body driven by said electromagnetic actuators; and a current command value determination unit that determines a command value of a current to be passed through each electromagnetic actuator to control the operation of said moving body or said rotating body, wherein said current command value determination unit determines a current command value for each electromagnetic actuator based on the value of either the electrical resistance or the induced voltage constant of said electromagnetic actuator so as to minimize the total loss generated in said plurality of electromagnetic actuators.

2. A motor comprising: a plurality of electromagnetic actuators; a moving body or a rotating body driven by said electromagnetic actuators; a current command value determination unit that determines a command value of a current to be passed through each electromagnetic actuator to control the operation of said moving body or said rotating body; and an assumed resistance setting unit that sets the electrical resistance of at least one of said plurality of electromagnetic actuators to an assumed value different from the actual electrical resistance value, wherein said current command value determination unit determines the current command value based on the resistance value set by said assumed resistance setting unit so as to minimize the total copper loss generated in said electromagnetic actuators.

3. The motor according to claim 2, characterized in that the assumed resistance setting unit sets the value of the electrical resistance of the electromagnetic actuator to a value different from the actual value of the electrical resistance depending on the position of the moving body or the rotation angle of the rotating body.

4. The motor according to claim 2, wherein the motor drives a plurality of the moving bodies or rotating bodies simultaneously, and the assumed resistance setting unit sets the value of the electrical resistance of the electromagnetic actuator in the assumed resistance setting unit to a value different from the actual value depending on the distance between the moving bodies or rotating bodies.

5. The motor according to claim 2, further comprising a temperature sensor for measuring the temperature of the electromagnetic actuator, and the assumed resistance setting unit sets the value of the electrical resistance of the electromagnetic actuator in the assumed resistance setting unit according to the temperature data measured by the temperature sensor.

6. A motor according to any one of claims 1 to 5, characterized in that it comprises a failure detection means for detecting a failure of the electromagnetic actuator and a parameter correction means, wherein the parameter correction means corrects either data relating to the induced voltage constant or data relating to the electrical resistance of the electromagnetic actuator in which a failure has been detected, and the current command value determination unit determines a current command value based on the corrected induced voltage constant or electrical resistance.

7. The motor according to any one of claims 1 to 6, wherein the moving body or the rotating body is a multi-degree-of-freedom motor having multiple degrees of freedom of movement.

8. The motor according to any one of claims 1 to 7, wherein the moving body is a linear motor.

9. The motor according to any one of claims 1 to 8, wherein the path along which the moving body moves is branched, and the path of the moving body is changed by the magnetic lateral force generated by the electromagnetic actuator.

10. A motor according to any one of claims 1 to 9, characterized in that it has multiple movers on the path of the moving body, and the multiple movers are moved simultaneously with the distance between the movers kept close to a distance shorter than the mover length, which is the length of the movers in the main direction of travel.

11. A motor as claimed in any one of claims 1 to 10, characterized in that it comprises a mover that is not mechanically constrained, and a levitation device that levitates the mover, and in that, while the mover is levitated, the movement of the mover, which has degrees of freedom in at least two directions different from the levitation direction, is controlled simultaneously by multiple electromagnetic actuators.

12. A motor according to any one of claims 1 to 11, characterized in that the moving body is a magnetic levitation stage that levitates a levitation stage using multiple electromagnetic actuators, and the levitation height and attitude of the magnetic levitation stage are controlled simultaneously.

13. The motor according to claim 12, wherein the electromagnetic actuator is driven so as to reduce mechanical deflection of the magnetic levitation stage.

14. A motor according to any one of claims 1 to 13, characterized in that the rotating body is a bearingless motor, and the rotation angle of the rotating body and the horizontal and vertical positions of the rotating body are simultaneously controlled by a plurality of the electromagnetic actuators.

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