Transient thermal resistance measurement device

Abstract

A transient thermal resistance measurement device 100 comprises: a heating power supply device 1 for supplying a heating current Ih to a semiconductor device 8 to be measured; a measurement power supply device 2 for supplying a measurement current Im to the semiconductor device 8 to be measured; a voltage measurement device 3 for measuring a voltage Vm across the semiconductor device 8 to be measured; and a constant current circuit 30 provided in a flow path of the measurement current Im. The constant current circuit 30 comprises: a transistor; a shunt resistor; a prescribed operational amplifier; and a prescribed voltage source. The prescribed operational amplifier controls, via a control terminal of the transistor, the current value of the measurement current Im flowing between main terminals of the transistor such that the voltage across the shunt resistor becomes equal to the output voltage of the prescribed voltage source. The prescribed operational amplifier has a unity gain frequency of 4 MHz or higher and a slew rate of 2 V / μs or higher.

Description

Transient thermal resistance measuring device

[0001] The present invention relates to a transient thermal resistance measuring device.

[0002] This application claims the benefit of foreign priority to Japanese Patent Application No. JP2024-113290, field July 16, 2024, which is incorporated by reference in its entirety. When performing the thermal design of a device incorporating a semiconductor device, it is important to understand the heat generation state of the semiconductor device. Power semiconductor devices in particular tend to be hot during use, potentially exceeding the product's guaranteed operating temperature range, so accurate temperature control is required. It is extremely important to understand the temperature rise process of power semiconductor elements during very short-term operation using pulse drive. In other words, it is necessary to accurately grasp the thermal characteristics that change over time during switching, and this can be achieved by transient thermal resistance measurement.

[0003] An example of an apparatus for measuring the thermal resistance of such a semiconductor device is the apparatus for measuring the thermal resistance of a semiconductor device described in Patent Document 1. In this measuring apparatus, a predetermined minute current is first supplied to the semiconductor device to be measured, and the voltage across the semiconductor device before heat generation is measured. Then, a predetermined current for heating is supplied to the semiconductor device for a predetermined period of time, causing the semiconductor device to generate heat and its temperature to rise. Then, a predetermined minute current is supplied and the voltage across the semiconductor device is measured. The amount of voltage change due to heat generation is then converted into an amount of temperature rise using the temperature dependency of the voltage of the semiconductor device, and the thermal resistance value of the semiconductor device is calculated from the amount of temperature rise and the power required to generate heat.

[0004] Japanese Patent Application Laid-Open No. 2013-113649 (see particularly FIGS. 11 and 15 and paragraphs

[0066] -

[0068] of the specification)

[0005] However, the conventional thermal resistance measuring device described above still has room for improvement in the following respects. Specifically, the transient thermal resistance characteristics of a semiconductor device can be measured during both the heat absorption process (temperature rise process) and the heat dissipation process (temperature fall process). However, the change in thermal resistance in the initial temperature change of a semiconductor device can be measured more accurately during the heat dissipation process than during the heat absorption process. However, the conventional thermal resistance measuring device described above has a problem in that it measures the thermal resistance of a semiconductor device during the heat absorption process. On the other hand, when measuring the heat dissipation process of a semiconductor device, there is a problem in that the transient thermal resistance cannot be accurately measured for a very short period of time immediately after the supply of a predetermined heating current is stopped (hereinafter simply referred to as "stop"), as described below.

[0006] The present invention has been made to solve the above-mentioned problems, and aims to provide a transient thermal resistance measuring device that can improve the measurement accuracy of transient thermal resistance during the very short period immediately after the heating current is stopped.

[0007] In order to achieve the above object, a transient thermal resistance measuring apparatus according to an aspect of the present disclosure includes a heating power supply that supplies a heating current to a semiconductor device under test for heating the device under test, a measuring power supply that supplies a minute DC measuring current to the semiconductor device under test, a voltage measuring apparatus that measures a voltage across the semiconductor device under test, and a constant current circuit provided in a flow path of the measuring current, the constant current circuit having a first main terminal, a second main terminal, and a control terminal, the measuring current flows between the first main terminal and the second main terminal, and a transistor whose current value changes in accordance with a potential difference of the control terminal with respect to a potential of the second main terminal, and one end of the transistor and a predetermined operational amplifier having an inverting input terminal to which the voltage at one end of the shunt resistor is input and a non-inverting input terminal to which the voltage at the positive terminal of a predetermined voltage source that outputs a current value setting voltage is input, and whose output is input to a control terminal of the transistor, wherein the first main terminal of the transistor and the other end of the shunt resistor are electrically connected to the flow path of the measurement current so that the constant current circuit is inserted in the flow path of the measurement current, and the negative terminal of the predetermined voltage source is electrically connected to the other end of the shunt resistor, and the predetermined operational amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

[0008] Furthermore, a transient thermal resistance measuring apparatus according to another aspect of the present disclosure includes a heating power supply that supplies a heating current to a semiconductor device under test to heat the device under test, a measuring power supply that is a constant current source that supplies a minute DC measuring current to the semiconductor device under test, a voltage measuring apparatus that measures the voltage across the semiconductor device under test, and a resistive element provided in the flow path of the measuring current.

[0009] The present invention has an effect of providing a transient thermal resistance measuring device that can improve the accuracy of measuring the transient thermal resistance in the very short period immediately after the heating current is stopped.

[0010] FIG. 1 is a functional block diagram showing an example of the configuration of a transient thermal resistance measuring apparatus according to a first embodiment of the present disclosure. FIG. 2 is a circuit diagram showing an example of the configuration of the constant current circuit of FIG. 1. FIG. 3 is a circuit diagram showing an example of the circuit configuration when the transistor of the constant current circuit of FIG. 2 is an N-channel MOSFET. FIG. 4 is a circuit diagram showing an example of the circuit configuration when the transistor of the constant current circuit of FIG. 2 is a P-channel MOSFET. FIG. 5A is a circuit diagram showing a method for measuring Vsd when a semiconductor device under test includes a MOSFET as a semiconductor element whose heat generation state is to be determined. FIG. 5B is a circuit diagram showing a method for measuring Vds when a semiconductor device under test includes a MOSFET as a semiconductor element whose heat generation state is to be determined. FIG. 5C is a circuit diagram showing a method for measuring Vth when a semiconductor device under test includes a MOSFET as a semiconductor element whose heat generation state is to be determined. FIG. 5D is a circuit diagram showing a method for measuring Vce when a semiconductor device under test includes an IGBT as a semiconductor element whose heat generation state is to be determined. FIG. 5E is a circuit diagram showing a method for measuring Vth when the semiconductor device under test includes an IGBT as a semiconductor element whose heat generation state is to be monitored. FIG. 5F is a circuit diagram showing a method for measuring the voltage across the semiconductor device under test when the semiconductor device under test is a TEG including a semiconductor element whose heat generation state is to be monitored. FIG. 6 is a flowchart showing the control content of the control device and voltage measurement device of FIG. 1. FIG. 7 is a graph schematically showing the effect of the comparative example. FIG. 8 is a graph schematically showing the effect of the transient thermal resistance measurement device of embodiment 1. FIG. 9 is a waveform diagram showing an actual measurement example of the effect of the comparative example. FIG. 10 is a graph showing the transition of junction temperature obtained by the comparative example. FIG. 11 is a waveform diagram showing an actual measurement example of the effect of the transient thermal resistance measurement device of embodiment 1. FIG. 12 is a graph showing the transition of junction temperature obtained by the transient thermal resistance measurement device of embodiment 1. FIG. 13 is a graph showing transient thermal resistance characteristics obtained by the comparative example and the transient thermal resistance measurement device of embodiment 1. FIG. 14 is a circuit diagram showing an example of the configuration of an improved constant current circuit together with a circuit for simulation. FIG. 15 is a waveform diagram showing the results of a simulation of the operation of a transient thermal resistance measuring device using an improved constant current circuit.Fig. 16 is a graph showing the results of a study on the response speed required for a specific operational amplifier. Fig. 17A is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 1 in Fig. 16. Fig. 17B is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 2 in Fig. 16. Fig. 17C is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 3 in Fig. 16. Fig. 17D is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 4 in Fig. 16. Fig. 17E is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 5 in Fig. 16. Fig. 18 is a diagram showing a resistance element in a transient thermal resistance measurement device according to a second embodiment of the present disclosure.

[0011] (Findings that led to the present disclosure) The present disclosure was made based on the following findings.

[0012] <First Finding> The specifications of semiconductor devices are generally defined when the temperature rise due to heat generation is in equilibrium. On the other hand, in a transient state of temperature rise due to heat generation, the semiconductor device itself absorbs the heat generated by the heat generation. Therefore, when a semiconductor device is used in a transient state of temperature rise due to heat generation, a large current can flow through the semiconductor device. This is the first finding.

[0013] <Second Finding> Therefore, the transient thermal resistance of a semiconductor device that is used to pass a large current by utilizing such a transient state cannot be accurately measured unless the semiconductor device is heated with a large current (e.g., 20 A to 1000 A) that is within the allowable limit of the semiconductor device. Therefore, the inventors attempted to measure the transient thermal resistance of a semiconductor device with such a large heating current. Measurement of transient thermal resistance is regulated, for example, by the standard JEDEC51-14 (JESD51-14). However, the following problems arose.

[0014] In applications that utilize a transient state to pass a large current as described above (hereinafter referred to as "transient state applications"), the voltage across the semiconductor device (junction temperature) must be accurately measured during a very short period of time immediately after the heating current is stopped (hereinafter referred to as the "specific transient thermal resistance measurement period"). This specific transient thermal resistance measurement period is, for example, within 10 μs. However, during this specific transient thermal resistance measurement period, the measurement current oscillates due to the sudden decrease in the measurement current caused by the cessation of the heating current, which causes oscillations in the voltage across the semiconductor device, making it difficult to accurately measure the transient thermal resistance. This oscillation in the measurement current was negligible when the heating current was small. This is the second finding.

[0015] <Third Finding> The inventors estimated that this oscillation in the measurement current was caused by the parasitic impedance of the wiring supplying the heating current to the semiconductor device, and came up with the idea of ​​inserting a specified resistor in the measurement current supply path to dampen the oscillation in the measurement current. Furthermore, they estimated that this oscillation in the measurement current was also caused by the constant current control of the current source supplying the measurement current being unable to keep up with (respond) to a sudden decrease in the measurement current, and as a countermeasure, they configured the specified resistor in a constant current circuit. As a result, they were able to accurately measure transient thermal resistance during the specified transient thermal resistance measurement period. This is their third finding.

[0016] The present disclosure has been made based on the above first to third findings. Specific contents of the present disclosure will be described in detail below.

[0017] A transient thermal resistance measuring apparatus according to an aspect of the present disclosure includes a heating power supply that supplies a heating current to a semiconductor device under test for heating the device under test, a measuring power supply that supplies a minute DC measuring current to the semiconductor device under test, a voltage measuring apparatus that measures a voltage between both ends of the semiconductor device under test, and a constant current circuit provided in a flow path of the measuring current, the constant current circuit having a first main terminal, a second main terminal, and a control terminal, the measuring current flows between the first main terminal and the second main terminal, and the current value of the measuring current changes depending on the potential difference of the control terminal with respect to the potential of the second main terminal, and a transistor having one end connected to the second terminal of the transistor. a predetermined operational amplifier having an inverting input terminal to which the voltage at one end of the shunt resistor is input and a non-inverting input terminal to which the voltage at the positive terminal of a predetermined voltage source that outputs a current value setting voltage is input, and whose output is input to a control terminal of the transistor, wherein the first main terminal of the transistor and the other end of the shunt resistor are electrically connected to the flow path of the measurement current so that the constant current circuit is inserted in the flow path of the measurement current, and the negative terminal of the predetermined voltage source is electrically connected to the other end of the shunt resistor, and the predetermined operational amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

[0018] Here, if the transistor is a MOSFET, the first main terminal, the second main terminal, and the control terminal are the drain, the source, and the gate, respectively. If the transistor is a bipolar transistor, the first main terminal, the second main terminal, and the control terminal are the collector, the emitter, and the base, respectively. If the transistor is an IGBT, the first main terminal, the second main terminal, and the control terminal are the collector, the emitter, and the gate, respectively.

[0019] With this configuration, depending on the type of transistor (NPN type, PNP type, N-channel type, or P-channel type), the measurement current flows sequentially through the first main terminal of the transistor, the second main terminal of the transistor, and the shunt resistor, or through the shunt resistor, the second main terminal of the transistor, and the first main terminal of the transistor. If the transistor is an NPN or N-channel type, an increase in the measurement current increases the absolute value of the voltage (positive voltage) across the shunt resistor. As a result, when the voltage across one end of the shunt resistor exceeds the voltage at the positive terminal of the predetermined voltage source, the predetermined operational amplifier lowers the potential at the control terminal of the transistor. This decreases the measurement current, decreasing the absolute value of the voltage across the shunt resistor. When the voltage across one end of the shunt resistor equals the voltage at the positive terminal of the predetermined voltage source, an equilibrium state is reached. On the other hand, if the transistor is a PNP or P-channel type, an increase in the measurement current increases the absolute value of the voltage (negative voltage) across the shunt resistor. As a result, when the voltage at one end of the shunt resistor falls below the voltage at the positive terminal of the specified voltage source, the specified operational amplifier increases the potential at the control terminal of the transistor. This decreases the measurement current, decreasing the absolute value of the voltage across the shunt resistor. When the voltage at one end of the shunt resistor equals the voltage at the positive terminal of the specified voltage source, an equilibrium state is reached. In this equilibrium state, the measurement current equals the absolute value of the current setting voltage divided by the resistance of the shunt resistor (hereinafter referred to as the set current value). This ensures that the measurement current is always controlled to the set current value. Furthermore, because the specified operational amplifier has a unity gain frequency of 4 MHz or higher and a slew rate of 2 V / μs or higher, the constant current circuit effectively follows the sudden decrease in the measurement current due to the cessation of the heating current, effectively suppressing oscillations in the measurement current associated with the sudden decrease. As a result, the accuracy of transient thermal resistance measurements can be improved for the very short period immediately after the heating current is stopped.

[0020] The predetermined operational amplifier may have a unity gain frequency of 50 MHz or more and a slew rate of 20 V / μs or more.

[0021] With this configuration, the constant current circuit more effectively follows the sudden decrease in the measurement current caused by the cessation of the heating current, and oscillation of the measurement current caused by the sudden decrease in the measurement current is more effectively suppressed.

[0022] The constant current circuit may include a main operational amplifier that is the specified operational amplifier, and a differential amplifier circuit that includes a sub-operational amplifier, wherein the differential amplifier circuit amplifies the difference between the voltage at one end of the shunt resistor and the voltage at the other end of the shunt resistor using the sub-operational amplifier, and the output of the sub-op amplifier is input to the inverting input terminal of the main operational amplifier as the voltage at one end of the shunt resistor relative to the voltage at the other end of the shunt resistor, and the sub-op amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

[0023] According to this configuration, the voltage across the shunt resistor is accurately detected, and the measurement current is therefore controlled to an accurate current value.

[0024] The main operational amplifier and the sub operational amplifier may each have a unity gain frequency of 50 MHz or more and a slew rate of 20 V / μs or more.

[0025] According to this configuration, the measurement current is controlled to an accurate current value, and oscillation of the measurement current due to a sudden decrease in the measurement current is more suitably suppressed.

[0026] Furthermore, a transient thermal resistance measuring apparatus according to another aspect of the present disclosure includes a heating power supply that supplies a heating current to a semiconductor device under test to heat the device under test, a measuring power supply that is a constant current source that supplies a minute DC measuring current to the semiconductor device under test, a voltage measuring apparatus that measures the voltage across the semiconductor device under test, and a resistive element provided in the flow path of the measuring current.

[0027] With this configuration, the resistance of the resistive element present in the measurement current flow path functions as a damper against sudden fluctuations in the measurement current, mitigating the sudden changes in the measurement current, thereby improving the accuracy of measuring the transient thermal resistance in the very short period immediately after the heating current is stopped.

[0028] The resistive element may have a resistance value of 2 Ω or more and 200 Ω or less.

[0029] This configuration effectively alleviates sudden changes in the measurement current, thereby effectively improving the accuracy of measuring the transient thermal resistance in the very short period immediately after the heating current is stopped.

[0030] Specific embodiments of the present disclosure will be described below with reference to the drawings. Note that, hereinafter, identical or corresponding elements throughout the drawings will be designated by the same reference numerals, and redundant description thereof will be omitted. Furthermore, since the following drawings are for explaining the present disclosure, elements unrelated to the present disclosure may be omitted, dimensions may be inaccurate due to exaggeration or the like, or may be simplified, and the shapes of corresponding elements may not match in multiple drawings. In the specification and claims of this application, "connecting" electrically or mechanically includes both "directly connecting" and "indirectly connecting." Furthermore, the present disclosure is not limited to the following embodiments.

[0031] <Definition> In this specification and claims, "measuring transient thermal resistance characteristics" includes both "directly measuring transient thermal resistance characteristics" and "indirectly measuring transient thermal resistance characteristics." Therefore, measuring the voltage Vm across a semiconductor device under test while supplying a minute measurement current Im to the semiconductor device under test is nothing other than "measuring transient thermal resistance characteristics."

[0032] For convenience, the voltage across the semiconductor device under test (DUT) when only the measurement current Im is supplied is referred to as the "voltage across Vm," and the voltage across the semiconductor device under test (DUT) when the heating current Ih (and the measurement current Im) are supplied is referred to as the "forward voltage VF."

[0033] Ignoring the minute measurement current Im, the "heating current Ih" is essentially the forward current IF for heating the semiconductor device under test (DUT). Since the term "forward current IF" is usually used when calculating transient thermal resistance characteristics, the "heating current Ih" may also be referred to as the "forward current IF" when calculating transient thermal resistance characteristics.

[0034] (Embodiment 1)

[0035] 1 is a functional block diagram showing an example of the configuration of a transient thermal resistance measuring device 100 according to the first embodiment of the present disclosure. Referring to Fig. 1, the transient thermal resistance measuring device 100 includes a heating power supply device 1, a measurement power supply device 2, a voltage measuring device 3, a constant current circuit 30 as a resistor, and a control device 5.

[0036] The heating power supply 1 supplies a heating current Ih to a semiconductor device under test (hereinafter sometimes referred to as DUT) 8 to heat the DUT 8. The measurement power supply 2 supplies a measurement current Im of a predetermined minute DC current value to the DUT 8. The voltage measurement device 3 measures the voltage Vm across the DUT 8. The control device 5 controls the operation of the heating power supply 1, measurement power supply 2, and voltage measurement device 3, and calculates the junction temperature transition and transient thermal resistance characteristics. The constant current circuit 30 is provided in the flow path of the measurement current Im.

[0037] The positive terminal of the heating power supply device 1 is connected to the positive terminal of the DUT 8 by a wire W1, and the negative terminal of the heating power supply device 1 is connected to the negative terminal of the DUT 8 by a wire W2. A backflow blocking diode 6 is provided in the middle of the wire W1. The heating power supply device 1 supplies a heating current Ih to the DUT 8 through the wires W1 and W2.

[0038] The positive terminal of the measurement power supply device 2 is connected to the positive terminal of the DUT 8 by a wire W3, and the negative terminal of the measurement power supply device 2 is connected to the negative terminal of the DUT 8 by a wire W4. A constant current circuit 30 is provided midway along the wire W3. A reverse current blocking diode 7 is also provided midway along the wire W3.

[0039] The measurement power supply device 2 passes a minute measurement current Im through the wires W3 and W4 to the DUT 8. Note that the negative terminal of the wire W3 and the positive terminal of the wire W4 may be connected to the wires W1 and W2 for the heating current Ih, respectively. However, this is not preferable because it increases the influence of the wiring impedance of the wires W1 and W2, thereby increasing the oscillation of the measurement current Im when the heating current Ih is stopped.

[0040] The positive input terminal of the voltage measuring device 3 is connected to the positive terminal of the DUT 8 by a wire W5, and the negative input terminal of the voltage measuring device 3 is connected to the negative terminal of the DUT 8 by a wire W6. The voltage measuring device 3 measures the voltage Vm across the DUT 8 (hereinafter may be simply referred to as the "voltage across the DUT") through the wires W5 and W6, creates a data file D of the voltage across the DUT 8 measurement data, and sends this to the control device 5. The voltage measuring device 3 also controls the on / off of the switch unit 1b of the heating power supply device 1 by a control signal Sc4.

[0041] The control device 5 uses control signals Sc1 to Sc3 to control the operations of the heating power supply device 1, the measurement power supply device 2, and the voltage measurement device 3. The control device 5 also calculates the junction temperature transition and transient thermal resistance characteristics using a data file D of the end-to-end voltage measurement data from the voltage measurement device 3.

[0042] Each of these components will be described in detail below.

[0043] <Heating Power Supply Device 1> The heating power supply device 1 includes a heating power supply unit 1a and a switch unit 1b.

[0044] The heating power supply unit 1a is not particularly limited as long as it can output a predetermined large current (e.g., 20 A to 1000 A) as the heating current Ih. The heating current Ih is a direct current. An example of the heating power supply unit 1a is a constant current source. The constant current source is, for example, configured as a switching power supply, but may also be configured as a linear power supply.

[0045] The switch unit 1b includes, for example, a switching element 11 connected to the positive output terminal of the heating power supply unit 1a and a diode 12 having an anode connected to the positive output terminal and a cathode connected to the negative output terminal of the heating power supply unit 1a. The switching element 11 is turned on and off by a control signal Sc4 from the voltage measurement device 3. When the switching element 11 is turned off while the switching element 11 and the heating power supply unit 1a are on and a heating current Ih is being output from the heating power supply device 1, the output current of the heating power supply unit 1a flows through the diode 12, thereby instantly stopping the output of the heating current Ih. Since the transient thermal resistance measurement device 100 measures the heat dissipation characteristics of the DUT 8, it is important that the output of the heating current Ih can be instantly stopped at the desired timing. Note that the switch unit 1b may be omitted from the heating power supply device 1. In this case, the supply of the heating current Ih is started or stopped by starting or stopping the heating power supply unit 1a, respectively.

[0046] <Measurement power supply device 2> The measurement power supply device 2 is not particularly limited as long as it can output a minute DC measurement current Im. The measurement power supply device 2 is configured, for example, as a constant voltage source or a constant current source, and in this case, the measurement power supply device 2 is configured as a constant voltage source that outputs a predetermined voltage Vo.

[0047] <Voltage Measuring Device 3> The voltage measuring device 3 measures the "end-to-end voltage" Vm, generates a data file D of end-to-end voltage measurement data, and turns on and off the switching element 11 of the switch unit 1b of the heating power supply device 1 using a control signal Sc4. The voltage measuring device 3 is configured, for example, by a computing unit having a processor and memory. A predetermined program for executing the functions of the voltage measuring device 3 is stored in the memory of this computing unit, and the voltage measuring device 3 is realized as a functional block by the processor reading and executing this predetermined program. In this case, the computing unit operates as the voltage measuring device 3. This computing unit can be configured, for example, by a computer, a personal computer, a microcontroller, an MPU, an FPGA (Field Programmable Gate Array), a PLC (Programmable Logic Controller), or the like.

[0048] <Control device 5> The control device 5 controls the operation of the heating power supply device 1, the measurement power supply device 2, and the voltage measurement device 3 using control signals Sc1 to Sc3, respectively, and calculates the junction temperature transition and transient thermal resistance characteristics. The control device 5 is configured, for example, by a computing unit having a processor and memory. A predetermined program for executing the functions of the control device 5 is stored in the memory of this computing unit, and the control device 5 is realized as a functional block by the processor reading and executing this predetermined program. In this case, the computing unit operates as the control device 5. This computing unit can be configured, for example, by a computer, a personal computer, a microcontroller, an MPU, an FPGA (Field Programmable Gate Array), a PLC (Programmable Logic Controller), etc.

[0049] It should be noted that the functions of the elements disclosed herein can be performed using circuits or processing circuits, including general-purpose processors, special-purpose processors, integrated circuits, ASICs (Application Specific Integrated Circuits), conventional circuits, and / or combinations thereof, configured or programmed to perform the disclosed functions. A processor is considered a processing circuit or circuit because it includes transistors and other circuitry. In this disclosure, a "circuit" or "unit" is hardware that performs the recited functions or hardware that is programmed to perform the recited functions. The hardware may be hardware disclosed herein or other known hardware that is programmed or configured to perform the recited functions. Where hardware is a processor, which is considered a type of circuit, the "circuit" or "unit" is a combination of hardware and software, and software is used to configure the hardware and / or processor.

[0050] <Constant Current Circuit 30> FIG. 2 is a circuit diagram showing an example of the configuration of the constant current circuit 30 of FIG. 1. Referring to FIG. 2, the basic constant current circuit 30 includes a transistor Q as a variable resistance element, a shunt resistor SH, a predetermined operational amplifier OA, and a predetermined voltage source 10. The transistor Q includes a first main terminal Tf, a second main terminal Ts, and a control terminal Tc. A measurement current Im flows between the first main terminal Tf and the second main terminal Ts, and the magnitude of the measurement current Im varies depending on the potential difference between the second main terminal Ts and the control terminal Tc. The predetermined operational amplifier OA has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more. The basis for these numerical limitations will be explained in detail later. The predetermined voltage source 10 outputs a current value setting voltage Vsig.

[0051] One end of the shunt resistor SH is connected to the second main terminal Ts of the transistor Q. One end of the shunt resistor SH is connected to the inverting input terminal of the predetermined operational amplifier OA. As a result, the voltage Vsh at one end of the shunt resistor SH is input to the inverting input terminal of the predetermined operational amplifier OA. One end of the shunt resistor SH may be directly connected to the inverting input terminal of the predetermined operational amplifier OA, or may be indirectly connected to the inverting input terminal of the predetermined operational amplifier OA (e.g., via a preceding circuit). The positive terminal of the predetermined voltage source 10 is connected to the non-inverting input terminal of the predetermined operational amplifier OA. The positive terminal of the predetermined voltage source 10 may be directly connected to the non-inverting input terminal of the predetermined operational amplifier OA, or may be indirectly connected to the non-inverting input terminal of the predetermined operational amplifier OA (e.g., via an input resistor).

[0052] The first main terminal Tf of the transistor Q constitutes the first terminal T1 of the constant current circuit 30, the other end of the shunt resistor SH constitutes the second terminal T2 of the constant current circuit 30, and the negative terminal of the predetermined voltage source 10 constitutes the third terminal T3 of the constant current circuit 30. The constant current circuit 30 is inserted into the wiring W3 (the flow path of the measurement current Im) so that the first terminal T1 and the second terminal T2 are connected to the wiring W3. The third terminal T3 of the constant current circuit 30 is connected to a reference potential point (ground as a negative reference potential point or Vcc as a positive reference potential point). The negative terminal of the predetermined voltage source 10 (the third terminal T3 of the constant current circuit 30) is electrically connected to the other end of the shunt resistor SH (the second terminal T2 of the constant current circuit 30). Specifically, for example, as shown in FIG. 14 , the negative terminal of the predetermined voltage source 10 is connected to ground, and the other end of the shunt resistor SH is substantially connected to ground by a differential amplifier circuit (measurement circuit 31). In the predetermined operational amplifier OA, the non-inverting input terminal and the inverting input terminal are imaginarily shorted, so that the shunt resistor SH and the predetermined voltage source 10 are substantially connected in parallel with each other.

[0053] Here, the transistor Q is a type of variable resistance element, an active resistance element whose resistance value can be controlled. If the transistor Q is a MOSFET, the first main terminal Tf, the second main terminal Ts, and the control terminal Tc are the drain, source, and gate, respectively. If the transistor Q is a bipolar transistor, the first main terminal Tf, the second main terminal Ts, and the control terminal Tc are the collector, emitter, and base, respectively. If the transistor Q is an IGBT, the first main terminal Tf, the second main terminal Ts, and the control terminal Tc are the collector, emitter, and gate, respectively.

[0054] 3 is a circuit diagram showing an example of a circuit configuration in which the transistor Q of the constant current circuit 30 of FIG. 2 is an N-channel MOSFET. Referring to FIGS. 1 and 3, the first terminal T1 of the constant current circuit 30 is connected to the downstream end of the measurement current Im at the measurement power supply 2 side of the wiring W3, and the second terminal T2 of the constant current circuit 30 is connected to the upstream end of the measurement current Im at the DUT 8 side of the wiring W3. The negative terminal of the predetermined voltage source 10 is electrically connected to the other end of the shunt resistor SH as described above. With this constant current circuit 30, the measurement current Im flows sequentially through the drain D (first main terminal Tf) of the transistor Q, the source S (second main terminal Ts) of the transistor Q, and the shunt resistor SH. If the measurement current Im increases, the absolute value of the voltage (positive voltage) across the shunt resistor SH increases. As a result, when the voltage at one end of the shunt resistor SH exceeds the voltage at the positive terminal of the predetermined voltage source 10, the predetermined operational amplifier OA lowers the potential at the gate G (control terminal Tc) of the transistor Q. This causes the measurement current Im to decrease, decreasing the absolute value of the voltage across the shunt resistor SH. When the voltage at one end of the shunt resistor SH becomes equal to the voltage at the positive terminal of the predetermined voltage source 10, an equilibrium state is reached. In this equilibrium state, the measurement current Im becomes equal to the absolute value of the current value setting voltage Vsig divided by the resistance value Rsh of the shunt resistor SH (hereinafter referred to as the set current value). This ensures that the current value of the measurement current Im is always controlled to the set current value. The current value setting voltage Vsig is selected so that the current value obtained by dividing the current value setting voltage Vsig by the resistance value Rsh of the shunt resistor SH becomes the desired set current value.

[0055] The response speed of the constant current circuit 30 is essentially determined by the response speed of the specified operational amplifier OA. Regarding the unity gain frequency and slew rate, which are indicators of the response speed of an operational amplifier, this specified operational amplifier OA has a unity gain frequency of 4 MHz or higher and a slew rate of 2 V / μs or higher. Therefore, the constant current circuit 30 can effectively follow the sudden decrease in the measurement current Im caused by the cessation of the heating current Ih, and oscillation of the measurement current Im due to the sudden decrease is effectively suppressed. As a result, the accuracy of transient thermal resistance measurement can be improved for the very short period immediately after the cessation of the heating current Ih. The effects of NPN-type bipolar transistors and IGBTs are similar to those described above.

[0056] 4 is a circuit diagram showing an example of a circuit configuration in which the transistor Q of the constant current circuit 30 of FIG. 2 is a P-channel MOSFET. Referring to FIGS. 1 and 4, the second terminal T2 of the constant current circuit 30 is connected to the downstream end of the measurement current Im at the measurement power supply 2 side of the wiring W3, and the first terminal T1 of the constant current circuit 30 is connected to the upstream end of the measurement current Im at the DUT 8 side of the wiring W3. The negative terminal of the predetermined voltage source 10 is electrically connected to the other end of the shunt resistor SH as described above. With this constant current circuit 30, the measurement current Im flows sequentially through the shunt resistor SH, the source D (second main terminal Ts) of the transistor Q, and the drain D (first main terminal Tf) of the transistor Q. If the measurement current Im increases, the absolute value of the voltage (negative voltage) across the shunt resistor SH increases. As a result, when the voltage at one end of the shunt resistor SH falls below the voltage at the positive terminal of the predetermined voltage source 10, the predetermined operational amplifier OA increases the potential at the gate G (control terminal Tc) of the transistor Q. This causes the measurement current Im to decrease, decreasing the absolute value of the voltage across the shunt resistor SH. When the voltage across the shunt resistor SH becomes equal to the voltage at the positive terminal of the predetermined voltage source 10, an equilibrium state is reached. In this equilibrium state, the measurement current Im reaches a current value (hereinafter referred to as the set current value) obtained by dividing the absolute value of the current value setting voltage Vsig by the resistance value Rsh of the shunt resistor SH. This ensures that the current value of the measurement current Im is always controlled to the set current value. The current value setting voltage Vsig is selected so that the current value obtained by dividing the current value setting voltage Vsig by the resistance value Rsh of the shunt resistor SH becomes the desired set current value. Furthermore, because the operational amplifier OA has a unity gain frequency of 4 MHz or higher and a slew rate of 2 V / μs or higher, the constant current circuit 30 can effectively follow the sudden decrease in the measurement current Im caused by the cessation of the heating current Ih, effectively suppressing oscillation of the measurement current Im due to the sudden decrease. As a result, the accuracy of transient thermal resistance measurement can be improved for the very short period immediately after the cessation of the heating current Ih. The effects of PNP-type bipolar transistors and IGBTs are similar to those described above.

[0057] <Semiconductor Device Under Test (DUT) 8> The DUT 8 includes one or more semiconductor elements whose heat generation state is to be monitored. Examples of such semiconductor elements include diodes, MOSFETs, IGBTs, and bipolar transistors. In this embodiment, the transient thermal resistance characteristics of the DUT 8 during its heat dissipation process are measured. To monitor the heat generation state of these semiconductor elements, the DUT 8 is heated by a heating current Ih (forward current IF), and then the heating current Ih is stopped. Then, a minute measurement current Im is supplied to the DUT 8 during its heat dissipation process, while measuring the voltage Vm across the DUT 8. This voltage Vm is converted to a junction temperature Tj based on its dependence on the junction temperature, and then converted to a transient thermal resistance θth(j-c). Furthermore, the DUT 8 is placed on, for example, a water-cooled heat sink for heat dissipation.

[0058] {Method of measuring various end-to-end voltages Vm} The end-to-end voltage Vm of the DUT 8 varies depending on the type and location of the semiconductor element whose heat generation state is to be grasped, and these will be explained in order below with reference to Figures 1 and 5A-5F.

[0059] <<Forward Voltage of Diode>> Referring to FIG. 1, when the DUT 8 includes a diode as a semiconductor element whose heat generation state is to be grasped, a forward heating current Ih is supplied to the diode to heat it up, and a forward measurement current Im is supplied to the diode to measure the forward voltage of the diode as a voltage Vm across both ends.

[0060] <MOSFET Vsd> Figure 5A is a circuit diagram showing a method for measuring Vsd when the DUT 8 includes a MOSFET as a semiconductor element whose heat generation state is to be monitored. In Figure 5A, an N-channel MOSFET is illustrated as an example of the MOSFET. Referring to Figure 5A, the N-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the source and a negative voltage is applied to the drain. A gate voltage negative with respect to the source is then applied to the gate, turning the N-channel MOSFET off. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the body diode 8a of the N-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the source-drain voltage Vsd of the N-channel MOSFET, which is the forward voltage of the body diode 8a due to the measurement current Im, is measured as the voltage Vm across both ends.

[0061] On the other hand, if the MOSFET is a P-channel MOSFET (not shown), the P-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the drain and a negative voltage is applied to the source. A gate voltage that is positive with respect to the source is applied to the gate, turning the P-channel MOSFET off. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the body diode of the P-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the source-drain voltage Vsd of the P-channel MOSFET, which is the forward voltage of the body diode due to the measurement current Im, is measured as the voltage Vm across both ends.

[0062] <MOSFET Vds> Figure 5B is a circuit diagram showing a method for measuring Vds when the DUT 8 includes a MOSFET as a semiconductor element whose heat generation state is to be monitored. Figure 5B illustrates an N-channel MOSFET as an example of the MOSFET. Referring to Figure 5B, the N-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the drain and a negative voltage is applied to the source. A gate voltage that is positive relative to the source is then applied to the gate, turning the N-channel MOSFET on. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the N-channel (inversion layer) of the N-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the drain-source voltage Vds of the N-channel MOSFET due to the measurement current Im is measured as the voltage Vm across both ends.

[0063] On the other hand, if the MOSFET is a P-channel MOSFET (not shown), the P-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the source and a negative voltage is applied to the drain. A gate voltage that is negative with respect to the source is then applied to the gate, turning the P-channel MOSFET on. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the P-channel (inversion layer) of the P-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the drain-source voltage Vds of the P-channel MOSFET due to the measurement current Im is measured as the voltage Vm across both ends.

[0064] <MOSFET Vth> Figure 5C is a circuit diagram showing a method for measuring Vth when the DUT 8 includes a MOSFET as a semiconductor element whose heat generation state is to be monitored. Figure 5C illustrates an N-channel MOSFET as an example of the MOSFET. Referring to Figure 5C, the N-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the drain and a negative voltage is applied to the source. The gate and drain are short-circuited, placing the N-channel MOSFET in a so-called "diode-connected" state. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the N-channel (inversion layer) of the N-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the threshold voltage Vth of the N-channel MOSFET due to the measurement current Im is measured as the voltage Vm across both ends.

[0065] On the other hand, if the MOSFET is a P-channel MOSFET (not shown), the P-channel MOSFET is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the source and a negative voltage is applied to the drain. The gate and drain are short-circuited, placing the P-channel MOSFET in a so-called "diode-connected" state. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the P-channel (inversion layer) of the P-channel MOSFET, heating the DUT 8. Next, the heating current Ih is stopped, and the threshold voltage Vth of the P-channel MOSFET due to the measurement current Im is measured as the voltage Vm across both ends.

[0066] <<IGBT Vce>> Figure 5D is a circuit diagram showing a method for measuring Vce when the DUT 8 includes an IGBT as the semiconductor element whose heat generation state is to be monitored. In Figure 5D, an N-channel IGBT is shown as an example of the IGBT. Referring to Figure 5D, the N-channel IGBT is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the collector and a negative voltage is applied to the emitter. A gate voltage that is positive with respect to the emitter is then applied to the gate, turning the N-channel IGBT on. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the N-channel IGBT, heating the DUT 8. Next, the heating current Ih is stopped, and the collector-emitter voltage Vce of the N-channel IGBT due to the measurement current Im is measured as the voltage Vm across both ends.

[0067] On the other hand, if the IGBT is a P-channel IGBT (not shown), the P-channel IGBT is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the emitter and a negative voltage is applied to the collector. A gate voltage that is negative with respect to the emitter is then applied to the gate, turning the P-channel IGBT on. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the P-channel IGBT, heating the DUT 8. Next, the heating current Ih is stopped, and the collector-emitter voltage Vce of the P-channel IGBT due to the measurement current Im is measured as the voltage Vm across both ends.

[0068] <<IGBT Vth>> Figure 5E is a circuit diagram showing a method for measuring Vth when the semiconductor device under test includes an IGBT as a semiconductor element whose heat generation state is to be monitored. Figure 5E illustrates an N-channel IGBT as an example of the IGBT. Referring to Figure 5E, the N-channel IGBT is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the collector and a negative voltage is applied to the emitter. The gate and collector are short-circuited, placing the N-channel IGBT in a so-called "diode-connected" state. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the N-channel (inversion layer) of the N-channel IGBT, heating the DUT 8. Next, the heating current Ih is stopped, and the threshold voltage Vth of the N-channel IGBT due to the measurement current Im is measured as the voltage Vm across both ends.

[0069] On the other hand, if the IGBT is a P-channel IGBT (not shown), the P-channel IGBT is connected to the heating power supply 1 and the measurement power supply 2 so that a positive voltage is applied to the emitter and a negative voltage is applied to the collector. The gate and collector are short-circuited, placing the P-channel IGBT in a so-called "diode-connected" state. In this state, a heating current Ih and a measurement current Im are first supplied. This causes the heating current Ih to flow through the P-channel (inversion layer) of the P-channel IGBT, heating the DUT 8. Next, the heating current Ih is stopped, and the threshold voltage Vth of the P-channel IGBT due to the measurement current Im is measured as the voltage Vm across both ends.

[0070] <<Voltage across TEG>> Figure 5F is a circuit diagram showing a method for measuring the voltage across the DUT 8 when the DUT 8 is a TEG (Test Elementary Group) including a semiconductor element whose heat generation state is to be monitored. Referring to Figure 5F, the TEG constituting the DUT 8 includes an IGBT 8c, which is the semiconductor element whose heat generation state is to be monitored, and a diode 8d is located near the IGBT. In this case, a heating current Ih is supplied to the IGBT 8c and a measurement current Im is supplied to the diode 8d to heat the DUT 8. After the heating current Ih is stopped, the forward voltage of the diode 8d due to the measurement current Im is measured as the voltage across the DUT 8. The same applies when the semiconductor element whose heat generation state is to be monitored is a MOSFET.

[0071] In the first embodiment, the heat radiation characteristics (temperature drop characteristics) of the DUT 8 are measured to measure the transient thermal resistance, so the DUT 8 is placed on a heat sink.

[0072] {Junction Temperature Tj of DUT 8} As is clear from the above explanation, the voltage Vm across DUT 8, which is the voltage drop due to the measurement current Im of the semiconductor element whose heat generation state is to be grasped, includes Vsd, which is the forward voltage of the diode and the forward voltage of the body diode of the MOSFET, as well as forward drop voltages of the pn junction such as the threshold voltage Vth of the MOSFET and the IGBT, and drop voltages of the channel (inversion layer) such as Vds, which is the drain-source voltage of the MOSFET, and Vce, which is the collector-emitter voltage of the IGBT. Therefore, hereinafter, for convenience, both the temperature at which the forward drop voltage of the pn junction is converted into the temperature of the pn junction and the temperature at which the drop voltage of the channel (inversion layer) is converted into the temperature of the channel (inversion layer) will be referred to as the "junction temperature Tj of DUT 8."

[0073] {Relationship with the Present Disclosure} As is clear from the above explanation, the heating power supply 1 and measurement power supply 2 of the transient thermal resistance measuring device 100 of the present disclosure only serve to supply the heating current Ih and measurement current Im to the DUT 8, respectively, and therefore the present disclosure is not limited in any way by the configuration of the DUT 8.

[0074] [Operation] Next, the operation of the transient thermal resistance measuring device 100 configured as above will be described with reference to Figures 1 and 6. This operation is performed by the control device 5 and voltage measuring device 3 working together to control the transient thermal resistance measuring device 100. Figure 6 is a flowchart showing the control details of the control device 5 and voltage measuring device 3 in Figure 1.

[0075] 1 and 6, first, the control device 5 starts operation of the heating power supply unit 1a of the heating power supply device 1 and starts supplying the measurement current Im to the measurement power supply device 2 (step S1). Specifically, the control device 5 outputs a control signal Sc1, which is an ON signal, to the heating power supply unit 1a of the heating power supply device 1. The heating power supply unit 1a then outputs a heating current to the switch unit 1b. This heating current flows through the diode 12 in the switch unit 1b and returns to the heating power supply unit 1a. The control device 5 also outputs a control signal Sc2, which is an ON signal, to the measurement power supply device 2. This causes the measurement power supply device 2 to start supplying the measurement current Im to the DUT 8. At this time, the constant current circuit 30 controls the measurement current Im to a desired set current value.

[0076] Next, the voltage measuring device 3 starts supplying a heating current Ih (forward current IF) to the switch unit 1b of the heating power supply device 1 and starts measuring the forward voltage VF of the DUT 8 (step S2). Specifically, the voltage measuring device 3 outputs an ON control signal Sc4 to the switch unit 1b of the heating power supply device 1. This turns on the switching element 11 of the switch unit 1b, and the heating current from the heating power supply unit 1a flows split between the diode 12 and the switching element 11. The heating current Ih output from the switching element 11 is supplied to the DUT 8 via the wiring W1 and wiring W2. This heating current Ih is detected by a heating current sensor (not shown) and feedback-controlled to a predetermined current value by the heating power supply unit 1a. The control device 5 also outputs an ON control signal Sc3 to the voltage measuring device 3. This causes the voltage measuring device 3 to start measuring the forward voltage VF of the DUT 8.

[0077] Next, after a predetermined heating period has elapsed, the voltage measuring device 3 causes the switch unit 1b of the heating power supply device 1 to stop supplying the heating current Ih, and causes the control device 5 to stop the heating power supply unit 1a of the heating power supply device 1 (step S3). The "predetermined heating period" is determined appropriately taking into consideration the structure of the DUT 8, the measurement conditions for the transient thermal resistance characteristics, and the like. The "predetermined heating period" is also determined to be a time sufficient for the temperature of the DUT 8 to saturate. Specifically, after the predetermined heating period has elapsed, the voltage measuring device 3 outputs a control signal Sc4, which is an off signal, to the switch unit 1b of the heating power supply device 1 and notifies the control device 5 of this. This turns off the switching element 11 of the switch unit 1b, and stops the output of the heating current Ih. As a result, all of the heating current from the heating power supply unit 1a flows through the diode 12. Meanwhile, upon receiving the above notification from the voltage measuring device 3, the control device 5 outputs a control signal Sc1, which is an off signal, to the heating power supply unit 1a of the heating power supply device 1. This causes the heating power supply unit 1a to stop outputting the heating current.

[0078] When the supply of the heating current Ih is stopped, a voltage drop occurs across the DUT 8 due to only the measurement current Im, and the voltage measuring device 3 begins to measure this voltage drop as the voltage Vm across the DUT 8 (step S4).

[0079] Next, when a predetermined measurement time has elapsed since the heating current Ih was stopped, the control device 5 causes the voltage measuring device 3 to acquire measurement data (step S5). Specifically, the control device 5 outputs a control signal Sc3, which is an off signal, to the voltage measuring device 3. This causes the voltage measuring device 3 to stop measuring the voltage Vm across the DUT 8 and create a data file D of the forward voltage VF, the voltage Vm across the DUT 8, the heating current Ih (forward current IF), and the measurement current Im, and transmits this data file D to the control device 5. The control device 5 also outputs a control signal Sc2, which is an off signal, to the measurement power supply device 2. This causes the measurement power supply device 2 to stop supplying the measurement current Im. Note that the order in which the measurement of the voltage Vm across the DUT 8 and the supply of the measurement current Im are stopped is not particularly limited. The control device 5 then calculates the transient thermal resistance characteristics of the DUT 8 based on this data file D, in accordance with the user's operation of a user interface (not shown). Specific procedures for calculating the transient thermal resistance characteristics will be described later.

[0080] [Operational Effects] The operational effects of the transient thermal resistance measuring device 100 of the first embodiment will be described below.

[0081] <Procedure for measuring transient thermal resistance characteristics> First, the procedure for measuring transient thermal resistance characteristics will be briefly described. Note that the measurement of transient thermal resistance θth(j−c) is specified in the standard: JEDEC51-14 (JESD51-14) as mentioned above.

[0082] First, the change characteristic of the voltage Vm across the DUT 8 with respect to the junction temperature Tj of the DUT 8 (hereinafter, sometimes referred to as the dependency of the voltage Vm across the DUT 8 on the junction temperature Tj) is measured. This measurement is performed, for example, by placing the DUT 8 in a thermostatic chamber and changing the temperature while supplying a measurement current Im to the DUT 8 to change the voltage Vm across the DUT 8. As is widely known, the voltage Vm across the DUT 8 decreases almost linearly as the junction temperature Tj increases.

[0083] Next, the voltage Vm across the DUT 8 is measured while changing the junction temperature Tj of the DUT 8. In this embodiment, the voltage Vm across the DUT 8 is measured while lowering the junction temperature Tj of the DUT 8 by heat dissipation through cooling using a water-cooled heat sink. To achieve this, the DUT 8 is first heated by a forward current IF (heating current Ih) for a "predetermined heating time." This heats the DUT 8 so that its junction temperature Tj reaches a saturated state. The forward voltage VF across the DUT 8 is measured during the "predetermined heating time." After the "predetermined heating time" has elapsed, heating of the DUT 8 is stopped. Then, from the point at which the heating is stopped, the voltage Vm across the DUT 8 is measured while the DUT 8 is cooled by the heat sink.

[0084] Next, using the above-mentioned dependency of the voltage Vm across the DUT 8 on the junction temperature Tj, the measured time-series data of the voltage Vm across the DUT 8 is converted into the junction temperature Tj of the time-series data of the DUT 8. In this case, since the voltage Vm across the DUT 8 immediately after the forward current IF (heating current Ih) is stopped cannot be accurately measured due to the recovery characteristics of the DUT 8, the junction temperature Tj(0) at the time the forward current IF (heating current Ih) is stopped is estimated.

[0085] Next, the transient thermal resistance characteristics of the time-series data are calculated by dividing the junction temperature change (Tj(0) - Tj(t)) obtained by subtracting the junction temperature Tj of the time-series data of DUT 8 from this junction temperature Tj(0) by the power P required to heat DUT 8. The power P required to heat DUT 8 is calculated by integrating the heating current Ih (set current value) x forward voltage VF at every moment during the "predetermined heating time."

[0086] Through the above measurements and calculations, the transient thermal resistance characteristics of the DUT 8 are measured.

[0087] <Example> An example was carried out to confirm the effect of the transient thermal resistance measurement device 100 of the first embodiment. The example is the transient thermal resistance measurement device 100 described above. A comparative example was also carried out to compare with the example. The configuration of the comparative example is the same as that of the transient thermal resistance measurement device 100, except that the constant current circuit 30 is not present.

[0088] {Overview} FIG. 7 is a graph schematically illustrating the effects of the comparative example. FIG. 8 is a graph schematically illustrating the effects of the transient thermal resistance measurement device 100 of the first embodiment. In FIGS. 7 and 8 , the top graph shows a waveform diagram of the heating current Ih (forward current IF), the second graph from the top shows a waveform diagram of the measurement current Im, the third graph from the top shows waveform diagrams of the forward voltage VF and the voltage Vm across the DUT 8, the fourth graph from the top shows the transition of the junction temperature Tj, and the fifth graph from the top shows the transient thermal resistance characteristic (θth(j-c)). Note that Tj(0) in the fourth graph from the top represents the estimated junction temperature immediately after the heating current is turned off. In each graph, the horizontal axis represents time. Furthermore, the vertical dashed line indicates the time when the heating current Ih is stopped.

[0089] 7, in the comparative example, when the heating current Ih is stopped, as shown in the area surrounded by the dotted circle, the measurement current Im oscillates and vibrates significantly immediately after the heating current Ih is stopped, which causes the voltage Vm across the DUT 8 to vibrate significantly. As a result, the portion of the transition of the junction temperature Tj immediately after the start of measurement fluctuates, making it impossible to measure this portion accurately, and making it impossible to measure the transient thermal resistance θth(j−c) during the very short period immediately after the heating current Ih is stopped, which is the "specific transient thermal resistance measurement period."

[0090] 8 , in the transient thermal resistance measurement apparatus 100 of the first embodiment, when the heating current Ih is stopped, the oscillation of the measurement current Im is suppressed immediately after the heating current Ih is stopped, as shown in the area surrounded by the dotted circle, and therefore the oscillation of the voltage Vm across the DUT 8 is also suppressed. This is because the oscillation of the measurement current Im is suppressed by the constant current circuit 30. As a result, the fluctuation of the junction temperature Tj immediately after the start of measurement is improved, improving the measurement accuracy of that portion. Furthermore, during the "specific transient thermal resistance measurement period," the transient thermal resistance θth(j−c) can be measured up to a time region very close to the start of measurement.

[0091] {Measurement Example} In this measurement example, a fast response electronic load was used as the constant current circuit 30. A commercially available multi-function electronic load device in constant current mode was used as this fast response electronic load.

[0092] FIG. 9 is a waveform diagram showing an example of measurements of the effects of the comparative example. FIG. 10 is a graph showing the transition of the junction temperature Tj obtained by the comparative example. FIG. 11 is a waveform diagram showing an example of measurements of the effects of the transient thermal resistance measurement device 100 of the first embodiment. FIG. 12 is a graph showing the transition of the junction temperature Tj obtained by the transient thermal resistance measurement device 100 of the first embodiment. FIG. 13 is a graph showing the transient thermal resistance characteristics obtained by the comparative example and the transient thermal resistance measurement device 100 of the first embodiment. This transient thermal resistance characteristic includes contact thermal resistance. FIGS. 9 and 11 were created by tracing images on the display screen of an oscilloscope, which displays the measurement results. FIGS. 10, 12, and 13 were created by tracing image data of graphs showing the measurement results.

[0093] 9 and 11, the upper waveform shows the waveform of the voltage Vm across the DUT 8, the middle waveform shows the waveform of the heating current Ih, and the lower waveform shows the waveform of the measurement current Im. Also, "T" in the figures indicates the time when the heating current Ih is stopped.

[0094] Referring to Figure 9, in the comparative example, as shown in the area surrounded by the dotted circle, immediately after the heating current Ih was stopped, the measurement current Im jumped out by about 160 mA, and this change did not converge for about 30 μs.

[0095] On the other hand, referring to Figure 11, in the transient thermal resistance measuring device 100 of this embodiment 1, as shown in the area surrounded by the dotted circle, immediately after the heating current Ih was stopped, the measurement current Im jumped out by about 60 mA, but this change converged within about 10 μs.

[0096] Referring to FIG. 10, in the comparative example, as shown in the area surrounded by the dotted circle, the transition of the junction temperature Tj fluctuated in the portion up to 1 ms after the start of measurement, and this portion could not be measured accurately.

[0097] On the other hand, referring to FIG. 12, in the transient thermal resistance measuring apparatus 100 of the present embodiment 1, as shown in the area surrounded by the dotted circle, the transition of the junction temperature Tj hardly fluctuated even up to 1 ms after the start of measurement, and this part could be accurately measured.

[0098] 13, the dashed line indicates the measurement results of the comparative example, and the solid line indicates the measurement results of the transient thermal resistance measurement device 100 of this embodiment 1. Referring to Fig. 13, as shown in the area surrounded by the dotted ellipse, in the comparative example, the measurement results were inaccurate during the "specific transient thermal resistance measurement period," and the transient thermal resistance θth(j−c) could not be measured accurately, whereas in the transient thermal resistance measurement device 100 of this embodiment 1, the transient thermal resistance θth(j−c) could be measured accurately during the "specific transient thermal resistance measurement period."

[0099] The above measurement examples have demonstrated that the transient thermal resistance measuring device 100 of this embodiment 1 can improve the measurement accuracy of the transient thermal resistance θth(j-c) during a specific transient thermal resistance measurement period (a very short period immediately after the heating current Ih is stopped).

[0100] [Improved Constant Current Circuit] FIG. 14 is a circuit diagram showing an example of the configuration of an improved constant current circuit 30A obtained by improving the constant current circuit 30 of FIGS. 2 and 3, together with a circuit for simulation.

[0101] 14, an improved constant current circuit 30A includes a transistor Q, a measurement circuit 31, and a control circuit 32. The transistor Q is the same as the transistor Q in the constant current circuit 30 of FIGS. 2 and 3, and therefore a description thereof will be omitted.

[0102] The measurement circuit 31 is composed of a differential amplifier circuit including an auxiliary operational amplifier OA2. Specifically, in the measurement circuit 31, a shunt resistor SH has a resistance value of, for example, 2.5 Ω. One end of the shunt resistor SH is connected to the non-inverting input terminal of the auxiliary operational amplifier OA2 via resistor element R8. Resistor elements R8 and R9 form a resistor voltage divider circuit, with the negative terminal of resistor element R9 connected to ground. The other end of the shunt resistor SH is connected to the inverting input terminal of the auxiliary operational amplifier OA2 via resistor element R7, which is an input resistor. Resistor element R7 is connected to resistor element R10, which is a negative feedback resistor. Resistor elements R7 to R10 each have a resistance value of, for example, 1 kΩ. Therefore, the amplification factor of this differential amplifier circuit is 1. In other words, this differential amplifier circuit serves to accurately measure the voltage across the shunt resistor SH relative to the ground potential, and therefore to accurately measure the measurement current Im flowing through the shunt resistor SH. Resistor element R10 and resistor element R9 are connected in parallel with capacitors C2 and C3, respectively. Therefore, the parallel circuit of resistor element R10 and capacitor C2, and the parallel circuit of resistor element R9 and capacitor C3, each form a low-pass filter. This differential amplifier circuit also serves to remove unwanted signals from the voltage across shunt resistor SH. This differential amplifier circuit also removes common-mode noise.

[0103] The portion of the measurement current Im flow path (wires W3 and W4) from the other end of shunt resistor SH to one output terminal (here, the negative terminal) of measurement power supply device 2 is connected to a reference potential point (here, ground (see FIG. 14)). Therefore, this differential amplifier circuit cancels the voltage drop due to backflow blocking diode 7 and DUT 8 in the measurement current Im flow path, and as a result, as described above, this differential amplifier circuit plays a role in accurately measuring the voltage across shunt resistor SH with respect to the ground potential.

[0104] The control circuit 32 includes a main operational amplifier OA1, which is the predetermined operational amplifier OA in FIGS. 2 and 3. The control circuit 32 performs the same function as the predetermined operational amplifier OA in FIGS. 2 and 3. The positive terminal of a predetermined voltage source 10, which outputs a current value setting voltage Vsig, is connected to the non-inverting input terminal of the main operational amplifier OA1 via a resistor element R5, which serves as an input resistor. The negative terminal of the predetermined voltage source 10 is connected to ground. The output terminal of the auxiliary operational amplifier OA2 is connected to the inverting input terminal of the main operational amplifier OA1 via a resistor element R6, which serves as an input resistor. The output terminal of the main operational amplifier OA1 is connected to the gate G of an N-channel MOSFET, which serves as a transistor Q, via a resistor element R2, which serves as a gate resistor. A snubber circuit consisting of a series circuit of a resistor element R3 and a capacitor C1 is provided between the inverting input terminal and output terminal of the main operational amplifier OA1. The power supply terminals of the main operational amplifier OA1 and the sub operational amplifier OA2 are connected to a pair of constant voltage sources 33 and 34, respectively.

[0105] The gate G of the N-channel MOSFET is connected to the source S via a resistive element R1, which is a gate-source resistor. The main operational amplifier OA1 and the sub operational amplifier OA2 are each composed of a high-speed operational amplifier. For example, a commercially available high-speed operational amplifier can be used as this high-speed operational amplifier.

[0106] In this improved constant current circuit 30A, the main operational amplifier OA1 receives the current setting voltage Vsig and the accurately measured voltage Vsh across the shunt resistor SH. The main operational amplifier OA1 operates to create an imaginary short circuit, resulting in Vsig - Vsh = 0. Then, a voltage obtained by multiplying Vsig - Vsh by the amplification factor of the main operational amplifier OA1 is output to the output terminal of the main operational amplifier OA1 and applied to the gate of the transistor Q. The transistor Q then generates a drain current (= measurement current Im) corresponding to the output characteristics of the transistor Q. This allows the voltage Vsh across the shunt resistor SH to be accurately detected, thereby controlling the measurement current Im to an accurate current value. Furthermore, because the main operational amplifier OA1 and the auxiliary operational amplifier OA2 are high-speed operational amplifiers, the response speed of the improved constant current circuit 30A is improved, and sudden changes in the measurement current Im are more effectively suppressed. As a result, transient thermal resistance characteristics can be measured more accurately. The reason why the response speed of the improved constant current circuit 30A is improved by using a high-speed operational amplifier will be described later.

[0107] 14 shows a circuit configuration for simulating a transient thermal resistance measurement device 100 using an improved constant current circuit 30A. Therefore, in this circuit configuration, the switch unit 1b of the heating power supply 1 is omitted. Lt1 and Lt2 are assumed as the wiring impedances of the wiring through which the heating current Ih flows. In this simulation, the heating current Ih was 200 A, and the time from when the heating current Ih started to stop until it was completely stopped was 5 μs. The measurement current Im was 1 A.

[0108] Figure 15 is a waveform diagram showing the results of a simulation of the operation of the transient thermal resistance measurement device 100 using the improved constant current circuit 30A. Figure 15 was created by tracing an image on the display screen of an oscilloscope, which is a display device for measuring results. In Figure 15, the horizontal axis represents time, and the vertical axis represents the current values ​​of the heating current Ih and the measurement current Im. In Figure 15, the upper waveform diagram represents the waveform of the heating current Ih, and the lower waveform diagram represents the waveform of the measurement current Im.

[0109] 15 , in this simulation, the measured current Im fluctuated as the heating current Ih was stopped. However, the fluctuations converged within 5 μs, from the start to the end of the heating current Ih stopping, and only an overshoot of approximately 1.04 A was observed. In other words, when the heating current Ih was stopped, the measured current Im changed into a single pulse with a peak value of approximately 0.04 A (40 mA) and did not oscillate. Therefore, the fluctuations in the measured current Im were drastically improved compared to when the improved constant current circuit 30A was not present.

[0110] Therefore, the above simulations confirmed that the transient thermal resistance measuring device 100 using the improved constant current circuit 30A can more effectively suppress sudden changes in the measurement current Im, and as a result, can measure the transient thermal resistance characteristics more accurately.

[0111] In the above, an N-channel MOSFET is used as the transistor Q in the improved constant current circuit 30A, but a P-channel MOSFET, a bipolar transistor, or an IGBT may also be used as the transistor Q. Since creating the improved constant current circuit 30A in this manner is self-evident from the above description and Figures 2 to 4, a description thereof will be omitted.

[0112] [Consideration of response speed required for a given operational amplifier OA] The response speed of the constant current circuit 30 is substantially determined by the response speed of the operational amplifier. This also applies to the improved constant current circuit 30A. Therefore, the inventors of the present invention have considered the response speed required for a given operational amplifier OA. Generally, the unity gain frequency (f Ｔ ) and slew rate (SR) are used. In the simulation circuit of FIG. 14, the inventors have set the main operational amplifier OA1 and the sub operational amplifier OA2 at various unity gain frequencies (f Ｔ The simulation was performed by replacing the operational amplifier with one having a unity gain frequency (f) and a slew rate (SR) to confirm the state of the measurement current Im when the heating current Ih is stopped. ＴThe results of the simulation for each sample operational amplifier are summarized in Fig. 16.

[0113] 16 is a graph showing the results of a study of the response speed required for a given operational amplifier OA. In the graph of FIG. 16, the horizontal axis represents the unity gain frequency (f Ｔ ) [MHz], and the vertical axis shows the slew rate (SR) [V / μs] on a logarithmic scale. On this graph, each sample operational amplifier is shown with its specified unity gain frequency (f Ｔ ) and slew rate (SR). Circles indicate plots of operational amplifiers in which the measurement current Im did not oscillate when the heating current Ih was stopped. Crosses indicate plots of operational amplifiers in which the measurement current Im oscillated when the heating current Ih was stopped. The area enclosed by the dashed dotted line represents the unity gain frequency (f Ｔ ) is the range in which oscillation of the measurement current Im can be prevented when the heating current Ih is stopped, and the range surrounded by the two-dot chain line is the range in which oscillation of the measurement current Im can be prevented when the heating current Ih is stopped, when viewed from the slew rate (SR). It goes without saying that if oscillation of the measurement current Im can be prevented when the heating current Ih is stopped, the accuracy of measuring the transient thermal resistance can be improved in the very short period immediately after the heating current Ih is stopped.

[0114] Referring to FIG. 16, the unity gain frequency (f Ｔ The higher the slew rate (SR), the better the results. To prevent the measurement current Im from oscillating when the heating current Ih is stopped, the operational amplifier must have a unity frequency (f) of approximately 4 MHz or higher. Ｔ) and a slew rate (SR) of 2 V / μs or more. The range that satisfies this requirement includes sample operational amplifiers in which the measurement current Im oscillates when the heating current Ih is stopped, but this is unavoidable since each sample operational amplifier has specifications specialized for its respective application. In order to obtain an operational amplifier suitable for the predetermined operational amplifier OA of the present disclosure, it is sufficient to select, through simulation, experiment, calculation, etc., from commercially available operational amplifiers that fall within this range, an operational amplifier that can prevent the measurement current Im from oscillating when the heating current Ih is stopped. Furthermore, operational amplifiers that can prevent the measurement current Im from oscillating when the heating current Ih is stopped include those with a unity frequency (f) of 4 MHz or more. Ｔ ) and a dedicated operational amplifier with a slew rate (SR) of 2 V / μs or greater.

[0115] Generally, the unity gain frequency (f Ｔ ) is said to be 50 MHz or higher, so the unity gain frequency (f Ｔ The slew rate (SR) of the sample operational amplifier with a frequency of 100 kHz is 20 V / μs or more. The area surrounded by the dotted line is the unity gain frequency (f Ｔ The operational amplifiers tested in this range are all capable of preventing oscillation of the measurement current Im when the heating current Ih is stopped. It is estimated that commercially available high-speed operational amplifiers fall within this range.

[0116] The plots numbered 1 to 5 are plots of representative sample operational amplifiers whose measurement current Im waveforms are shown in Figures 17A to 17F. Figure 17A is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 1 in Figure 16. Figure 17B is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 2 in Figure 16. Figure 17C is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 3 in Figure 16. Figure 17D is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 4 in Figure 16. Figure 17E is a waveform diagram showing the waveform of the measurement current Im in a simulation of the operation of a transient thermal resistance measurement device using the sample operational amplifier numbered 5 in Figure 16. In these figures, the horizontal axis represents time (μs) and the vertical axis represents current (A).

[0117] Referring to FIG. 16, the sample operational amplifier number 1 is included in the area surrounded by the dotted line and has the second highest unity gain frequency (f Ｔ 17A, the measurement current Im exhibited a single pulse-like change with a peak value of about 0.036 A (36 mA) when the heating current Ih was stopped, and did not oscillate.

[0118] Referring to FIG. 16, the sample operational amplifier No. 2 is included in the area surrounded by the dotted line and has the highest unity gain frequency (f Ｔ 17B, ​​the measurement current Im exhibited a single pulse-like change with a peak value of about 0.038 A (38 mA) when the heating current Ih was stopped, and did not oscillate.

[0119] Referring to FIG. 16, the sample operational amplifier No. 3 is located at the lower limit of both the range enclosed by the dashed line and the range enclosed by the dashed line, and has a unity gain frequency (f Ｔ17C, the measurement current Im exhibited a single pulse-like change with a peak value of about 0.049 A (49 mA) when the heating current Ih was stopped, and did not oscillate.

[0120] Referring to FIG. 16, the sample operational amplifier No. 4 is located at the lower limit of the range enclosed by the dashed line and at the bottom of the range enclosed by the dashed line, and has a unity gain frequency (f Ｔ 17D, the measurement current Im oscillated with an amplitude of approximately 0.030 A (30 mA) on one side when the heating current Ih was stopped.

[0121] Referring to FIG. 16, the sample operational amplifier No. 5 is located outside the range surrounded by the dashed line and the range surrounded by the two-dot chain line, and has the highest unity gain frequency (f Ｔ 17E, the measurement current Im oscillated with an amplitude of approximately 0.300 A (300 mA) on one side when the heating current Ih was stopped.

[0122] Thus, the state of the measured current Im when the heating current Ih is stopped in the representative plots numbered 1 to 5 supports the conclusions regarding the consideration of the response speed required for the specified operational amplifier OA described above. Since the response speed of the constant current circuit 30 is substantially determined by the response speed of the operational amplifier, it goes without saying that the results of the above consideration apply to the constant current circuit 30.

[0123] Second Embodiment FIG. 18 is a diagram showing a resistance element in a transient thermal resistance measuring device according to a second embodiment of the present disclosure.

[0124] 18, in the transient thermal resistance measuring device according to the second embodiment, a resistance element 40 is arranged in the flow path of the measurement current Im instead of the constant current circuit 30 of the first embodiment. Also, the measurement power supply device 2 is configured as a constant current source. Other than this, the configuration of the transient thermal resistance measuring device according to the second embodiment is the same as that of the transient thermal resistance measuring device 100 according to the first embodiment.

[0125] The resistance element 40 has a resistance value of, for example, 2 Ω or more and 200 Ω or less. The resistance value of the resistance element 40 is determined according to the current value of the measurement current Im and the forward voltage VF of the DUT 8. The resistance value of the resistance element 40 is determined, for example, as follows. First, the measurement current Im is set to a current value at which internal heat generation of the DUT 8 is negligible. The measurement current Im is typically set so that internal heat generation of the DUT 8 is within 1 W. For example, if the DUT 8 is a diode, and the measurement current Im is set to 0.1 A and the forward voltage VF of the diode (voltage Vm across the DUT 8) is, for example, 0.6 V, the internal heat generation of the DUT 8 is 0.1 A x 0.6 V = 0.06 W (≦1 W).

[0126] The output voltage Vo of the measurement power supply 2 (see FIG. 1) depends on the specifications of the measurement power supply 2. For example, if the output voltage Vo of the measurement power supply 2 is 20 V and the measurement current Im is 0.1 A, the resistance value R of the resistive element 40 is given by the following formula:

[0127] According to the second embodiment, the damping function of the resistance element 40 reduces sudden changes in the measurement current Im, thereby improving the measurement accuracy of the transient thermal resistance characteristics.

[0128] Other Embodiments In the first or second embodiment, the heating power supply device 1 may supply a small heating current Ih.

[0129] In the first or second embodiment, the transient thermal resistance measuring apparatus 100 may measure the heat absorption characteristics (temperature rise characteristics) of the DUT 8 .

[0130] In the first or second embodiment, the switch unit 1b may be omitted from the heating power supply device 1, and the heating power supply device 1 may output the heating current Ih as a rectangular wave pulse current.

[0131] Many modifications and alternative embodiments will be apparent to those skilled in the art in light of the above description, and therefore the above description should be construed as illustrative only.

[0132] The transient thermal resistance measuring device of the present invention is useful as a transient thermal resistance measuring device that can improve the accuracy of measuring the transient thermal resistance in the very short period immediately after the heating current is stopped.

[0133] REFERENCE SIGNS LIST 1 heating power supply device 1a heating power supply section 1b switch section 2 measurement power supply device 3 voltage measurement device 5 control device 6, 7 reverse current blocking diode 8 semiconductor device under test (DUT) 10 predetermined voltage source 30 constant current circuit 30A improved constant current circuit 40 resistance element OA predetermined operational amplifier OA1 main operational amplifier OA2 auxiliary operational amplifier

Claims

1. A device comprising: a heating power supply that supplies a heating current to a semiconductor device under test for heating the device under test; a measurement power supply that supplies a minute DC measurement current to the semiconductor device under test; a voltage measurement device that measures the voltage across the semiconductor device under test; and a constant current circuit provided in a flow path of the measurement current, wherein the constant current circuit comprises: a transistor having a first main terminal, a second main terminal, and a control terminal, through which the measurement current flows between the first main terminal and the second main terminal, and the current value of the measurement current changes in accordance with the potential difference of the control terminal relative to the potential of the second main terminal; a shunt resistor having one end electrically connected to the second terminal of the transistor; and a predetermined operational amplifier having an inverting input terminal to which the voltage of one end of the shunt resistor is input, and a non-inverting input terminal to which the voltage of the positive terminal of a predetermined voltage source that outputs a current value setting voltage is input, and whose output is input to the control terminal of the transistor, a first main terminal of the transistor and the other end of the shunt resistor are electrically connected to the flow path of the measurement current so that the constant current circuit is inserted in the flow path of the measurement current, and a negative terminal of the specified voltage source is electrically connected to the other end of the shunt resistor; and the specified operational amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

2. The transient thermal resistance measuring device according to claim 1, wherein the predetermined operational amplifier has a unity gain frequency of 50 MHz or more and a slew rate of 20 V / μs or more.

3. The transient thermal resistance measuring device according to claim 1, wherein the constant current circuit comprises a main operational amplifier which is the specified operational amplifier, and a differential amplifier circuit including a sub-operational amplifier, wherein the differential amplifier circuit amplifies the difference between the voltage at one end of the shunt resistor and the voltage at the other end of the shunt resistor using the sub-operational amplifier, wherein the output of the sub-operational amplifier is input to the inverting input terminal of the main operational amplifier as the voltage at one end of the shunt resistor relative to the voltage at the other end of the shunt resistor, and wherein the sub-operational amplifier has a unity gain frequency of 4 MHz or more and a slew rate of 2 V / μs or more.

4. The transient thermal resistance measuring device according to claim 3, wherein the main operational amplifier and the auxiliary operational amplifier each have a unity gain frequency of 50 MHz or more and a slew rate of 20 V / μs or more.

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