Method, device and optical arrangement for determining and controlling a phase difference between a measurement signal and a reference signal

Abstract

The invention relates to a method and a device for determining a phase difference (6) between a measurement signal (4) and a reference signal (5), at least comprising: converting the measurement signal (4) into a first square-wave signal (33) and the reference signal (5) into a second square-wave signal (35), which are each substantially symmetrical square-wave signals (33, 35), the pulse duration of which corresponds to half a period duration; determining the phase difference (6) by means of a phase comparison of the two square-wave signals (33, 35), wherein, by means of a logic module (36, 63), a pulse-duration-modulated signal is generated, from which the phase difference (6) is extracted. The invention also relates to an optical arrangement (200) for determining a phase difference (6) between a first optical signal (1) and a second optical signal (2).

Description

[0001] Description

[0002] title

[0003] Method, device and optical arrangement for determining and controlling a phase difference between a measurement signal and a reference signal

[0004] State of the art

[0005] The invention relates to a method, a device and an optical arrangement for determining a phase difference between a measurement signal and a reference signal, as well as an optical arrangement for determining and controlling a phase difference between a measurement signal and a reference signal.

[0006] To measure and control the phases of multiple optical beams within a beam, the phase shift between the beams must be measured. A well-known method for measuring the phase shift between two beam sources, for example in the near-infrared range, is based on heterodyne detection. Here, a beam source is superimposed with a frequency-shifted reference signal. The interference of the beams with the reference source generates a measurement signal for each beam, containing information about its phase and frequency. Unlike the original beams, the measurement signal lies within a frequency range that can be directly measured, for example, with photodiodes. Due to the acousto-optic modulator used to shift the reference signal, the beat frequencies typically lie between 10 MHz and 200 MHz.Of central importance here is the determination of the phase difference to a reference from these measurement signals. This allows the phase differences of several optical beams within the beam array to be determined. Furthermore, control variables can be calculated from the information about the beam phase differences, which are then transmitted, usually in analog form, to actuators. These actuators are then able to influence the phase of the individual beams.

[0007] There are various approaches to obtaining the phase shift between the measurement signal and the reference signal. One such approach, for example, uses an exclusive-OR (XOR) gate to determine the phase difference between two signals.

[0008] An integrated circuit that outputs a phase signal between two signals using an XOR gate and can also be used in the frequency ranges in question is, for example, the AD8302. Its phase signal is analog, which is why an analog-to-digital converter (ADC) is required to process the phase signal in a digital controller. In addition, a microcontroller and a digital-to-analog converter (DAC) are typically needed to calculate a new control value from the phase signal and output it in analog form. Alternatively, there are also analog applications where, for example, a proportional-integral differential controller (PID controller) is used. If wavefront measurements are to be performed at many measurement points using this technique, a relatively large number of components are required per channel, which corresponds to one measurement point.Not only are the costs per channel high with this arrangement, but also the large space requirement per channel on a circuit board and the requirements for signal integrity mean that this arrangement is only scalable to a limited extent with the number of channels.

[0009] Other approaches, for example, use so-called field-programmable gate arrays (FPGAs) to enable the parallelization of multiple channels in a single processing unit or component. However, these approaches require at least one dedicated multi-bit ADC per channel if an XOR gate is used for phase demodulation. If the so-called in-phase quadrature (I&Q) method is used for phase demodulation, two multi-bit ADC conversions are required per phase value. Furthermore, a dedicated DAC is always used for the analog control signal output. The additional components required for these conversions increase the footprint on the circuit board and its complexity, which hinders the scalability of the channels. Analog PID control would also be possible here, but this would require a correspondingly large number of components.

[0010] US6147755A discloses a detector for dynamically determining the optical phase state of light, which provides an output signal with a magnitude that changes linearly relative to the optical phase state of a first light beam. An analog phase signal is extracted for phase evaluation. US6229616B1 describes an optical heterodyne wavefront sensor that uses a radio frequency signal to measure an optical wavefront whose phase state varies across the entire aperture. Phase evaluation is performed using an XOR phase comparator and a subsequent integrator to generate an analog phase signal.

[0011] US6366356B1 describes a high average power fiber laser system with a parallel

[0012] High-speed wavefront sensor for phase compensation of a secondary beam. Digital dividers are used to divide electrical signals by a selected integer, and an exclusive-OR gate is used to generate a pulse train with a duty cycle that reflects the phase to be compensated.

[0013] US6243168B1 describes a dynamic optical micrometer for measuring the position of a moving object.

[0014] In the cited texts, the phase value of the optical rays is only available in analog form and must subsequently be digitized using a multi-bit ADC.

[0015] Disclosure of the invention

[0016] The object of the invention is to provide an efficient method for determining a phase difference between a measurement signal and a reference signal. A further object is to create a device for determining a phase difference between a measurement signal and a reference signal using such a method, which allows for scalability of a number of measurement channels.

[0017] Another task is to create an optical arrangement for determining a phase difference between a first optical signal and a second optical signal using such a method and such a device.

[0018] Another task is to create an optical arrangement for determining and controlling a phase difference between a first optical signal and a second optical signal using such a method and such a device.

[0019] The problems are solved by the features of the independent claims. Favorable embodiments and advantages of the invention become apparent from the further claims, the description, and the drawings.

[0020] According to one aspect of the invention, a method for determining a phase difference between a measurement signal and a reference signal is proposed, comprising at least converting the measurement signal into a first square wave signal and the reference signal into a second square wave signal, each being substantially symmetrical square wave signals whose pulse duration corresponds to half a period; and determining the phase difference by comparing the phases of the two square wave signals, wherein a pulse-duration modulated signal is generated by means of a logic module, from which the phase difference is extracted. The square wave signals at the input of the logic module can be analog or digital square wave signals. The processing of the signals after the logic module can be entirely digital. In an embodiment with an optical signal, the optical signal is modulated with the logic states.The further processing of the optical signal can involve analog signal processing, for example with a camera.

[0021] Both signals, the measurement signal and the reference signal, can be electrical signals and, for example, exhibit a sinusoidal waveform. These two signals are converted into signals consisting of logic states in order to be processed further. The shape of these signals corresponds to an essentially symmetrical square wave, whose pulse duration is half a period, thus each having a duty cycle of 50%. Furthermore, the frequency of these square waves corresponds to the frequency of the respective input signal.

[0022] For each channel, i.e., each beam of an optical beam, a circuit with a comparator as its core element can be used. The unaltered sinusoidal oscillation of the electrical signal is fed to the first input of the comparator. The second input also receives this sinusoidal oscillation, but it is first filtered, for example, by low-pass filtering. This ensures that the signal is converted into a substantially symmetrical square wave, regardless of its amplitude or DC component. The output of the comparator can then be directly connected electrically to a phase comparator. The proposed method can be advantageously used for targeted energy transfer. This enables power scaling of lasers with high beam quality, particularly through coherent laser coupling.Optical distortions can be compensated for to their advantage.

[0023] In an alternative embodiment, the proposed method can also be used to decode at least one phase-coded data transmission. The measurement signal can be generated, for example, using the so-called phase-shift keying method. A scalable number of phase-coded data signals can be fed as measurement signals via photodiodes, or directly as a voltage signal, into binarization units to generate square wave signals. The reference signal can be generated locally and is not permanently linked to a transmitting side of the measurement signal. Alternatively, the reference signal can be transmitted separately or extracted from the data signal.

[0024] The proposed method requires, for example, only one analog comparator and one filter per channel, specifically at least one low-pass filter, or more precisely, two low-pass filters if a control signal is to be output using a delta-sigma modulator. This allows the phase difference to be determined from a sinusoidal measurement signal to a reference signal, a setpoint to be calculated, and the setpoint to be output in analog form. Optionally, a high-pass or band-pass filter can be used instead of the low-pass filter. Therefore, compared to conventional arrangements, neither a multi-bit ADC nor a dedicated multi-bit DAC is required. Furthermore, no additional components such as RF mixers or dedicated logic ICs are used.

[0025] Since the low-pass filters can be implemented as RC circuits and thus each only have one resistor and one capacitor, the arrangement required for the proposed method, besides a logic element, needs only a few components per channel if this arrangement is not to be implemented as an integrated circuit. A typical FPGA is capable of processing the logic operations of several hundred channels simultaneously. Since the other components per channel have very small dimensions (less than or equal to 1 mm), the design is very efficient. 2 Since these components are available, the scaling problems described above are significantly reduced. It is also conceivable to implement the entire system, including the photodiodes and transimpedance amplifiers, as an integrated circuit, which further increases scalability.

[0026] Advantageously, the use of an analog low-pass filter after the exclusive-OR operation can be omitted. It is not necessary to output the signal after the exclusive-OR operation, integrate it analogously as in the prior art, and then digitize it again. The integration is advantageously implemented solely within the counter structure in the logic.

[0027] In contrast, in US6147755A and US6229616B1, the signal from the exclusive-OR operation is output as a voltage value. To obtain a phase signal, the signal is low-pass filtered in each case. Only the low-pass filtered signal then represents a phase signal. According to the invention, the signal remains digital. Only in the case of optional evaluation using a downstream camera can the camera itself act as a low-pass filter.

[0028] Advantageously, the square wave signals can be generated in such a way that their output signals are symmetrical regardless of the DC component of the output signal and their pulse duration corresponds to half a period.

[0029] Alternatively, a setup with a capacitor as a high-pass filter and a comparator that is compatible with negative voltages can be chosen.

[0030] With a favorable implementation of the method, the phase difference can be determined by means of an exclusive-OR operation (XOR operation) of the two rectangular signals.

[0031] In such a phase detector, the two signals can each be fed to an input of an XOR gate. The XOR gate outputs a one whenever only one of the two signals is true. In all other cases, it outputs zero. Accordingly, the output of the XOR gate outputs a one for each period of one signal, always for the time that one signal is shifted relative to the other. If a low-pass filter is applied to the resulting digital signal, the output signal represents the phase difference between the two input signals.

[0032] With a favorable implementation of the method, the output binary states of a logic operation of the logic module can be digitally evaluated over a defined time range. Advantageously, the states of the logic operation, which are typically read out at a clocked rate, can be summed and thus averaged over a defined period, for example, over one period of the reference signal. Advantageously, AND, OR, and XOR gates, as well as NAND, NOR, and NXOR gates and combinations thereof, and SR flip-flops, can generate a pulse-width modulated signal that depends on the phase difference of the inputs. Preferably, for example, XOR gates and an asynchronous, edge-triggered SR flip-flop can be used.

[0033] In a favorable embodiment of the method, the pulse duration of the pulse-duration modulated signal can be directly detected. In this alternative embodiment, unlike other embodiments where the average state of the logic module is determined, the pulse duration can be measured directly.

[0034] With a favorable implementation of the method, the result of the logic block's operation can be processed by a first counter, and the clock signal can be processed by a second counter. If the second counter overflows, the current value of the first counter can be used as the phase difference value.

[0035] Advantageously, a unit can evaluate the output states, for example, of an exclusive-OR operation, over a defined time period and calculate an average. This average can then be used as the phase difference. Specifically, the result of the exclusive-OR operation can be processed using a first counter. The clock signal can be processed using a second counter, and when the second counter overflows, the current value of the first counter can be used as the phase difference value.

[0036] The result of the XOR operation can then be fed into a counter. The counting range of this counter can advantageously correspond to the desired phase resolution N or a multiple thereof. In parallel to this counter, another counter can be operated, which counts the clock pulses of the clock generator. The signals are sampled at the same clock pulse from the clock generator. Both counters can conveniently have the same counting range. When the counter driven by the clock generator overflows, the current count of the counter after the XOR gate is adopted as the new phase value.

[0037] The phase difference value determined in this way, however, contains no information about the sign of the phase difference. Since the XOR operation is always performed at fixed phase values ​​due to the fixed relationship between the clock signal and the reference frequency, it is also possible to determine whether the XOR gate output positive values ​​in the first or second half of the oscillation period, or indeed at all. With this information, the sign can then be directly derived.

[0038] The resolution can also be increased without adjusting the sampling rate. This can be achieved by exploiting frequency fluctuations in the clock signal and the averaging properties of the counter's design. However, the necessary increase in the counter's counting range reduces the frequency at which new phase values ​​are available. This can be mitigated by multiplying the initially calculated sampling rate. Correspondingly to the multiplication of the sampling rate, the counting range can then also be multiplied.

[0039] With a favorable implementation of the method, the pulse-duration modulated signal generated by the logic module can be output as an optical signal. The signal can advantageously be digitally modulated. Furthermore, photoelectric integration of the signal is possible.

[0040] The advantages of optical transmission lie in the fact that the signal remains binary (pulse-width modulated). Therefore, the signal can be evaluated both digitally, using a photodetector and subsequent components as in other configurations, and analogously, for example, through photoelectric integration.

[0041] For photoelectric integration, for example, a CCD / CMOS camera that records pixels can be used with an exposure time that is greater than the modulation frequency of the phase signals.

[0042] Preferably, the camera's image acquisition is synchronized to the pulse-width modulated signal to achieve consistent results. With synchronization, even the recording of just one modulation is sufficient to capture a suitable signal.

[0043] A camera, for example, allows for the simultaneous acquisition of well over 100,000 signals. Alternatively, for photoelectric integration, a photodetector (e.g., photodiode, phototransistor, photomultiplier) with low-pass characteristics or a downstream low-pass filter can be used.

[0044] Signal transmission can be optical. This allows for good signal separation and shielding from interference. Crosstalk can be avoided. This can be achieved using separate optical fibers or via free-space transmission.

[0045] If the phase information originates from an optical system and, for example, encodes phase errors requiring correction, then in a free-jet transmission system, the phase information can be sent retrogradely through the optical system to the corresponding phase-correcting elements in order to implement the appropriate phase corrections using phase modulators installed there. This allows the phase correction to be assigned to the individual phase modulators simply and without additional cables. To minimize interference with the outgoing optical radiation, the retrograde optical signal can have a different wavelength, polarization, spatial distribution (beam profile), or a slightly different direction.

[0046] The phase information is encoded in the pulse width. Therefore, the signal is in principle resilient to unknown attenuations, especially in digital evaluation methods.

[0047] In a favorable embodiment of the method, the second square wave signal, converted from the electrical reference signal, can be combined with a clock signal that exhibits a frequency shift relative to the electrical reference signal. Advantageously, this allows for high phase resolution without requiring very high sampling rates. In a favorable embodiment of the method, the electrical measurement signal and the electrical reference signal can be sampled with the clock signal using bistable elements, particularly D flip-flops, bistable multivibrators, or cascaded bistable elements. Optionally, the exclusive-OR operation can be clocked with the clock signal. Alternatively, an asynchronous logic device without clocking can be used for this purpose.

[0048] A flip-flop is a bistable multivibrator that has two stable output states, with the output signal also available as an inverted signal. The simplest type of flip-flop is the so-called SR flip-flop. This basic element permanently holds any one of its two possible output states. It can be changed via its two inputs, usually labeled S (set input) and R (reset input). The clock-level-controlled D flip-flop contains, in addition to the data input D, a static input that reacts to a specific level. This multivibrator represents a typical latch as a one-bit data storage device.

[0049] To achieve sufficiently high phase resolution without requiring very high sampling rates, the reference signal and the measurement signal can be sampled using bistable elements and a clock generator slightly frequency-shifted relative to the reference signal. Since the reference signal, while not having the same frequency as the clock signal, is fixed relative to it, a chain of bistable elements can be used to reduce metastable states. A suitable sampling frequency for this clock signal can be determined as follows: = ref (1 -(N + 1 )' 1 Here, fret is the frequency of the reference signal and N is the desired phase resolution as an integer. The XOR operation between the previously sampled values ​​is performed at the same clock signal.

[0050] With a favorable implementation of the method, the phase difference can be determined from the two square wave signals using a logic component, in particular an SR flip-flop. An XOR gate, for example, can be used as the logic component. The advantage of the SR flip-flop is that the full phase can be resolved over a range of 2π. The output signal of the logic component can be advantageously processed using an optical emitter or digitally.

[0051] In a favorable embodiment of the method, at least one optical emitter can be operated with an output signal from the logic module, the average intensity of which is proportional to the phase. In particular, the intensity can be determined using a camera.

[0052] In another embodiment, a circuit almost identical to the previously described examples, using a bistable element as a logic comparator, can be used to modulate the duty cycle of an optical emitter such as a light-emitting diode (LED) or a laser diode, for example, a vertical cavity surface-emitting laser (VCSEL). In this case, the pulse width encodes the phase of the system. By duplicating the optical emitters in an alternating emission mode (on / off off / on), it can also be ensured that the sign is spatially encoded. A preferably synchronized camera can thus record an optical signal that encodes the phase of the element. This avoids scalability limitations due to signal lines or crosstalk between the lines.Furthermore, a clear assignment of the signal sources is possible, and with the correct selection of optical emitters, even technically sophisticated high-speed cameras can be used for laser radiation.

[0053] If the laser source emits at 1550 nm to generate the optical signals, for example, the signals can be detected by the photodiodes and the secondary signal can then be output with LEDs at, for example, 850 nm.

[0054] Alternatively, the radiation emitted by the optical emitter can be propagated back through an optical system to be controlled, in order to directly provide an analog signal at the radiation source to be controlled, which is suitable for control.

[0055] According to a favorable embodiment of the method, the electrical measurement signal and the electrical reference signal can be subjected to a reference DC voltage, whereby the reference DC voltage is used to define a common voltage threshold for the two square wave signals.

[0056] In this embodiment of input signal binarization, the input signals can be modified to include an externally determined DC component. This DC component corresponds to the common reference DC voltage. This reference DC voltage can also be fed into an FPGA of the phase comparator. Here, this reference DC voltage is used to define the voltage threshold that distinguishes between the high and low states of the square wave signals. One logic standard that supports this is, for example, high-speed transceiver logic (HSTL). As with the previously described binarization units, this arrangement ensures that the input signals correspond to an essentially symmetrical square wave, meaning that the low and high states of the square wave signal each have a duty cycle of 50%.

[0057] With a favorable embodiment of the method, a control variable for regulating the phase difference of the optical measurement signal can be determined from the phase difference, particularly by means of a control unit. Specifically, an analog control variable for regulating the phase difference of the optical measurement signal can be provided by means of a delta-sigma modulator. The delta-sigma modulator does not determine the control variable itself, but is merely part of the digital-to-analog converter and thus only outputs the control variable.

[0058] The determined phase value can then be fed as an actual value into any control algorithm. Since the phase value is in digital form, it can also be easily stored, further analyzed, or output via a digital interface.

[0059] The manipulated variable calculated in the control algorithm can be fed to a delta-sigma modulator if the manipulated variable is to be output analogously. In the delta-sigma modulator, a bitstream with a bit width of one can be generated from the multi-bit manipulated variable. If this bitstream is low-pass filtered, an analog manipulated variable signal is obtained. The low-pass filter can be implemented, for example, with an RC circuit. The analog manipulated variable signal can then be passed to an actuator, which can change the phase of a beam as desired.

[0060] With a favorable embodiment of the method, a plurality of phase differences of a plurality of optical measurement signals can be determined simultaneously. Advantageously, a large number of rays from an optical beam can thus be measured.

[0061] With a favorable implementation of the method, a plurality of phase differences of a plurality of optical measurement signals can be controlled simultaneously. Advantageously, phase differences of a large number of beams within an optical beam can thus be controlled.

[0062] In a favorable embodiment of the method, the reference signal can be an electrical reference signal.

[0063] With a favorable embodiment of the method, the phase difference between an optical measurement signal and the reference signal can be determined. The optical measurement signal can then be converted into an electrical measurement signal.

[0064] Advantageously, the optical measurement signal, which can be generated from two optical signals using a heterodyne optical system, can be converted into an electrical voltage signal using a photodetector and a transimpedance amplifier. The second input signal can be the electrical reference signal used to generate the optical measurement signal. This electrical reference signal can be generated, for example, with a function generator.

[0065] In a favorable embodiment, the method can further comprise generating a first optical signal with a first frequency and a first phase using a laser source; generating a second optical signal with a second frequency and a second phase from the first optical signal from the laser source using the electrical reference signal, wherein the second optical signal has a frequency shift relative to the first optical signal; superimposing the first optical signal with the second optical signal to form the optical measurement signal.

[0066] Advantageously, the optical measurement signal can be generated from the two optical signals using a heterodyne optical arrangement. The optical measurement signal is a beat signal generated by superimposing the two optical signals.

[0067] Interfering with the second optical beam generated by the reference source, the first optical beam creates a measurement signal containing information about the beam's phase and frequency. Unlike the output beam, the measurement signal lies within a frequency range that can be directly measured, for example, using photodiodes. Due to the acousto-optic modulator used to shift the reference signal, the beat frequency typically lies between 10 MHz and 200 MHz.According to a further aspect of the invention, a device for determining a phase difference between a measurement signal and a reference signal is proposed using a method described above, comprising at least a binarization unit which converts the measurement signal into a first square wave signal and the reference signal into a second square wave signal, each of which is essentially a symmetrical square wave signal whose pulse duration corresponds to half a period, a phase comparison unit which determines the phase difference by comparing the phases of the two square wave signals, and a logic module which generates a pulse-duration modulated signal from which the phase difference can be extracted.

[0068] Both signals, the measurement signal and the reference signal, can be electrical signals and, for example, exhibit a sinusoidal waveform. These two signals are converted in the binarization unit into signals consisting of logical states in order to be processed further.

[0069] The shape of these signals corresponds to an essentially symmetrical square wave, whose pulse duration is half a period, thus each having a duty cycle of 50%. Furthermore, the frequency of these square waves corresponds to the frequency of the respective output signal.

[0070] For each channel, i.e., each ray of an optical beam, a circuit with a comparator as its core element can be used. The unaltered sine wave of the electrical signal is fed to the first input of the comparator. The second input also receives this sine wave, but it is first low-pass filtered. This ensures that the signal is converted into a substantially symmetrical square wave, regardless of its amplitude or DC component. The comparator's output can then be directly connected to a phase comparator.

[0071] The proposed device can be advantageously used for targeted energy transfer. This enables power scaling of lasers with high beam quality, particularly through coherent laser coupling. Optical disturbances can also be advantageously compensated.

[0072] The proposed device requires, in addition to an element in which logic and other digital operations of the phase comparison unit can be implemented, such as an FPGA, only one analog comparator and one filter device per channel, for example a low-pass filter, in order to determine the phase difference to a reference signal from a sinusoidal measurement signal, to calculate a setpoint and to output this in analog form.

[0073] Optionally, a high-pass filter or a band-pass filter can be used instead of the low-pass filter.

[0074] Therefore, compared to conventional devices for a heterodyne process, neither a multi-bit ADC nor a dedicated multi-bit DAC is required. Furthermore, no additional components such as RF mixers or dedicated logic ICs are used. Since the low-pass filters can be implemented as RC circuits and thus each consist of only a resistor and a capacitor, the proposed device requires only a few components per channel, in addition to a logic element, if this arrangement is not to be implemented as an integrated circuit. A typical FPGA is capable of processing the logic operations of several hundred channels simultaneously. Since the additional components per channel are very small (less than or equal to 1 mm), the device is designed for very small dimensions. 2Since these components are available, the scaling problems described above are significantly reduced. It is also conceivable to implement the entire system, including the photodiodes and the transimpedance amplifier, as an integrated circuit, which further increases scalability.

[0075] Advantageously, the proposed device can be designed in a modular manner. For example, key components of the phase comparison unit, including a control unit, a clock generator, and / or a reference voltage source, can be integrated into an FPGA chip, enabling a very compact device design.

[0076] Such a device can advantageously be connected to an existing optical arrangement which provides a measurement signal and a reference signal in order to realize a determination and / or control of phase differences of a large number of beams of a beam.

[0077] In a favorable embodiment of the device, the binarization unit for the measurement signal and the reference signal can each comprise a comparator and a filter, in particular a low-pass filter. The measurement signal can be fed to the first input of one comparator, and the reference signal to the first input of the other comparator. In parallel, the measurement signal and the reference signal are filtered, in particular by low-pass filtering, and then fed to the corresponding second input of each comparator. This ensures that the signal is converted into the form of a substantially symmetrical square wave, regardless of its amplitude or DC component. Optionally, the two signals can also be high-pass or band-pass filtered. Instead of the device with a low-pass filter, devices using high-pass or band-pass filters can also be employed.

[0078] In a favorable design of the device, the phase comparison unit can include an exclusive-OR gate and a clock generator.

[0079] In such a phase detector, the two signals can each be fed to an input of an XOR gate. The XOR gate outputs a one whenever only one of the two signals is true. In all other cases, it outputs zero. Accordingly, the output of the XOR gate outputs a one for each period of one signal, always for the time that one signal is shifted relative to the other. If a low-pass filter is applied to the resulting digital signal, the output signal represents the phase difference between the two input signals. To achieve a sufficiently high phase resolution without requiring very high sampling rates, the reference signal and the measurement signal can be sampled using bistable elements with a clock frequency shifted relative to the reference signal. The frequency-shifted clock, in turn, generates a second beat frequency in the phase measurement.

[0080] In a favorable embodiment of the device, an output of the exclusive-OR gate can be electrically coupled to a first counter, and the clock generator to a second counter. The first and second counters can be coupled to a bistable element, in particular a D flip-flop, at whose output the phase difference is applied. In particular, the first counter and the second counter can comprise a plurality of cascaded bistable elements.

[0081] The result of the XOR operation can then be fed into a counter. The counting range of this counter can advantageously correspond to the desired phase resolution N or a multiple thereof. In parallel to this counter, another counter, driven by a clock generator, can be operated, which also performs sampling. Both counters can conveniently have the same counting range. When the counter driven by the clock generator overflows, the current count of the counter after the XOR gate is adopted as the new phase value.

[0082] The phase difference value determined in this way, however, contains no information about the sign of the phase difference. Since the XOR operation is always performed at fixed phase values ​​due to the fixed relationship between the clock signal and the reference frequency, it is also possible to determine whether the XOR gate output positive values ​​in the first or second half of the oscillation period, or indeed at all. With this information, the sign can then be directly derived.

[0083] The resolution can also be increased without adjusting the sampling rate. This can be achieved by exploiting frequency fluctuations in the clock signal and the averaging properties of the counter's design. However, the necessary increase in the counter's counting range reduces the frequency at which new phase values ​​are available. This can be mitigated by multiplying the initially calculated sampling rate. Correspondingly to the multiplication of the sampling rate, the counting range can then also be multiplied.

[0084] In a favorable embodiment of the device, the inputs of the exclusive-OR gate can be coupled to a first bistable element, in particular a D flip-flop, for the electrical measurement signal and to a second bistable element, in particular a D flip-flop, for the electrical reference signal, wherein the first and the second bistable elements are triggered by the clock generator. The two bistable elements are used to sample both electrical signals. They thus constitute a 1-bit ADC.

[0085] In a favorable embodiment of the device, the phase comparison unit can include the logic component, in particular an SR flip-flop. For example, an XOR gate can be used as the logic component. The advantage of the SR flip-flop is that the full phase is resolved over a range of 2π. In a favorable embodiment of the device, the logic component can drive at least one optical emitter, and the phase comparison unit can include a camera that determines the intensity of the optical emitter. Phase determination can thus be advantageously performed.

[0086] In another embodiment, a circuit almost identical to the previously described examples, using a bistable element as a logic comparator, can be employed to modulate the duty cycle of an optical emitter such as a light-emitting diode (LED) or a laser diode, for example, a vertical cavity surface-emitting laser (VCSEL). In this case, the pulse width encodes the phase of the system. Furthermore, by doubling the optical emitters in an alternating emission mode (on / off off / on), it can be ensured that the sign is spatially encoded.

[0087] A preferably synchronized camera can record an optical signal that encodes the phase of the element. This avoids scalability limitations due to signal lines or crosstalk between lines. Furthermore, unambiguous identification of signal sources is possible, and with the correct selection of optical emitters, even technically sophisticated high-speed cameras can be used for long-wavelength (NIR / MIR / SWIR) laser radiation. If the laser source emits at 1550 nm to generate the optical signals, for example, the signals can be detected by photodiodes, and the secondary signal can then be output by LEDs at, for example, 850 nm. In this way, standard (silicon) cameras designed for 1550 nm can be used.Alternatively, the radiation emitted by the optical emitter can be propagated back through an optical system to be controlled, in order to directly provide an analog signal at the radiation source to be controlled, which is suitable for control.

[0088] In a favorable design of the device, the phase comparator can have a DC reference input. The DC reference defines the DC component of the signals fed into the phase comparator. The DC reference is applied to so-called bias tees, each consisting of an inductor and a capacitor. In this setup, the DC component of the electrical measurement signal from the transimpedance amplifier is cut off, and the DC reference is applied to the signal as a DC component. This arrangement ensures that the input signals correspond to a substantially symmetrical square wave, meaning that the low and high states of the square wave each have a duty cycle of 50%.

[0089] In a favorable embodiment, the device can further include a control unit which determines a manipulated variable for controlling the phase difference of the measurement signal.

[0090] The determined phase value can be fed as an actual value to any control algorithm. Since the phase value is in digital form, it can also be easily stored, further analyzed, or output via a digital interface. With a favorable device design, an output of the control unit can be electrically coupled to a delta-sigma modulator, which provides an analog control variable for adjusting the phase difference of the measured signal.

[0091] The manipulated variable calculated in the control algorithm can, if the manipulated variable is to be output analogously, be fed to a so-called delta-sigma modulator. This modulator can generate a bitstream from the multi-bit manipulated variable with a bit width of one. If this bitstream is low-pass filtered, an analog manipulated variable signal is obtained. The low-pass filter can be implemented, for example, with an RC circuit. The analog manipulated variable signal can then be passed to an actuator, which can change the phase of a beam as desired.

[0092] In a favorable embodiment, the device can be configured to simultaneously determine a plurality of phase differences of a plurality of measurement signals. Advantageously, this allows the measurement of a large number of beams from an optical beam.

[0093] In a favorable embodiment, the device can be configured to simultaneously control a plurality of phase differences of a plurality of measurement signals. Advantageously, this allows the control of phase differences of a large number of beams within an optical beam.

[0094] With a favorable design of the device, at least the phase comparison unit can be integrated into a single component, in particular an FPGA component. This advantageously enables a compact device design, even for a large number of channels, to determine the phase differences of a large number of beams within a beam.

[0095] With a favorable design of the device, the control unit and / or the delta-sigma modulator can be integrated into the component, in particular the FPGA component. This advantageously enables an even more compact device design, even for a large number of channels for controlling the phase differences of a large number of beams within a beam.

[0096] In a favorable embodiment, the device for determining a phase difference between an optical measurement signal and the reference signal can further comprise a receiving unit that converts the optical measurement signal into an electrical measurement signal. The electrical measurement signal can then be processed in the other units of the device as described above.

[0097] Advantageously, an optical measurement signal, which can be generated from two optical signals using a heterodyne optical arrangement, can be converted into an electrical voltage signal by the receiving unit, for example, by a photodetector and a transimpedance amplifier. The second input signal of the device is the electrical reference signal, which was used, for example, to generate the optical measurement signal. The electrical reference signal can be generated, for example, with a function generator.According to a further aspect of the invention, an optical arrangement for determining a phase difference between a first optical signal and a second optical signal is proposed using a method and device described above, further comprising a laser source generating the first optical signal, a function generator generating an electrical reference signal, an acousto-optic modulator electrically coupled to the function generator to generate a second optical signal with a frequency shift relative to the first optical signal, and a superposition unit that superimposes the first optical signal with the second optical signal to form an optical measurement signal with the phase difference of the two optical signals.

[0098] With the proposed optical arrangement, an optical measurement signal of the two optical signals is generated by means of a heterodyne optical arrangement with an acousto-optic modulator and a superposition unit, and converted into an electrical voltage signal by means of a receiving unit, for example by means of a photodetector and a transimpedance amplifier.

[0099] The second output signal of the optical arrangement is the electrical reference signal used to generate the second optical signal, which is generated, for example, by a function generator. Both signals are sinusoidal. In the binarization unit, these two signals are converted into signals consisting of logic states for further processing. The shape of these signals corresponds to an essentially symmetrical square wave, with a pulse duration of half a period, thus each having a duty cycle of 50%. Furthermore, the frequency of these square waves corresponds to the frequency of the respective output signal. For each channel, i.e., each beam, of an optical beam, a circuit with a comparator as its core element can be used. Here, the unaltered sinusoidal wave of the electrical signal is fed to the first input of the comparator.The second input also receives this sine wave, but it is first low-pass filtered. This ensures that the signal is converted into a substantially symmetrical square wave, regardless of its amplitude or DC component. The comparator's output can then be directly connected to a phase comparator.

[0100] The proposed optical arrangement can be advantageously used for targeted energy transfer. This enables power scaling of lasers with high beam quality, particularly through coherent laser coupling. Optical disturbances can also be advantageously compensated.

[0101] The proposed optical arrangement requires, in addition to an element in which the logic and other digital operations of the phase comparison unit can be implemented (e.g., an FPGA), only one analog comparator and one filter per channel, specifically a low-pass filter, to determine the phase difference from a sinusoidal measurement signal to a reference signal, calculate a setpoint, and output it in analog form. Optionally, a high-pass or band-pass filter can be used instead of the low-pass filter.

[0102] Therefore, compared to conventional setups, neither a multi-bit ADC nor a dedicated multi-bit DAC is required. Furthermore, no additional components such as RF mixers or dedicated logic ICs are used. Since the low-pass filters can be implemented as RC networks, each consisting of only a resistor and a capacitor, the proposed design requires only a few components per channel, in addition to a logic element, unless it is implemented as an integrated circuit. A typical FPGA is capable of processing the logic operations of several hundred channels simultaneously. Because the additional components per channel are very small (less than or equal to 1 mm), the overall design is very efficient. 2Since these components are available, the scaling problems described above are significantly reduced. It is also conceivable to implement the entire system, including the photodiodes and transimpedance amplifiers, as an integrated circuit, which further increases scalability.

[0103] According to a further aspect of the invention, an optical arrangement for determining and controlling a phase between a first optical signal and a second optical signal is proposed using a method and device described above, further comprising a laser source generating the first optical signal, a function generator generating an electrical reference signal, an acousto-optic modulator electrically coupled to the function generator to generate a second optical signal with a frequency shift relative to the first optical signal, a superposition unit that superimposes the first optical signal with the second optical signal to form an optical measurement signal with the phase difference of the two optical signals, and a control unit that determines a manipulated variable for controlling the phase difference of the two optical signals.Advantageously, the proposed optical arrangement allows not only the determination of the phase difference but also its control. Furthermore, the optical arrangement can conveniently control the phase differences of multiple beams within an optical beam.

[0104] drawing

[0105] Further advantages will become apparent from the following description of the drawings. The figures illustrate exemplary embodiments of the invention. The figures, the description, and the claims contain numerous features in combination. A person skilled in the art will expediently consider the features individually and combine them into meaningful further combinations.

[0106] They show, for example:

[0107] Fig. 1 shows an optical arrangement with a device for determining a phase difference between a first and a second optical signal according to a method according to an embodiment of the invention;

[0108] Fig. 2 shows the device of the optical arrangement according to Figure 1;

[0109] Fig. 3 shows a schematic representation of the sampling process using a clock signal to determine the phase difference between a measurement signal and a reference signal;

[0110] Fig. 4 shows a system diagram of a phase comparison unit according to an embodiment of the invention; Fig. 5 shows a block diagram of the device for determining and controlling a phase difference between an optical measurement signal and an electrical reference signal according to an embodiment of the invention;

[0111] Fig. 6 shows a device for determining and controlling a phase difference between an optical signal and an electrical reference signal according to a further embodiment of the invention;

[0112] Fig. 7 shows a device for determining and controlling a phase difference between an optical signal and an electrical reference signal according to a further embodiment of the invention; and

[0113] Fig. 8 shows a device for determining and controlling a phase difference between an optical signal and an electrical reference signal according to a further embodiment of the invention;

[0114] Fig 9 shows a time course of a measurement signal and a reference signal with a determination of a phase difference by evaluating the measurement signal and the reference signal via an XOR operation and an asynchronously edge-triggered SR flip-flop.

[0115] Embodiments of the invention

[0116] In the figures, similar or equivalent components are numbered with the same reference symbols. The figures merely show examples and are not intended to be restrictive. The directional terminology used below, with terms such as "left," "right," "above," "below," "in front," "behind," "after," and the like, serves only to improve the understanding of the figures and is in no way intended to limit their generality. The components and elements shown, their interpretation, and their use may vary according to the considerations of a person skilled in the art and may be adapted to the respective applications.

[0117] Figure 1 shows an optical arrangement 200 with a device 100 for determining a phase difference 6 between a first optical signal 1 and a second optical signal 2 according to a method according to an embodiment of the invention.

[0118] The optical arrangement 200 comprises a laser 10, which generates the first optical signal 1 in the form of a laser beam with a first frequency and a first phase. The optical arrangement 200 further comprises a function generator 14, which generates an electrical reference signal 5, and an acousto-optic modulator 12, which is electrically coupled to the function generator 14, such that a second optical signal 2 is generated, which has a frequency shift relative to the first optical signal 1.

[0119] The first optical signal 1 passes through the acousto-optic modulator 12 on its propagation path. Within the acousto-optic modulator 12, the second optical signal 2 with a second frequency and a second phase is generated from the first optical signal 1 using the electrical reference signal 5. The second optical signal 2 exhibits a frequency shift relative to the first optical signal 1. The first optical signal 1 is then fed to a superposition unit 20, which superimposes the first optical signal 1 and the second optical signal 2 to form an optical beat signal, the measurement signal 3, with the phase difference 6 of the two optical signals 1 and 2. For this purpose, the second optical signal 2 can be guided into the superposition unit 20 by a mirror 18.

[0120] The optical arrangement 200 further comprises a device 100 with a receiver unit 21 with a photodiode 22 and a transimpedance amplifier 24, which converts the optical measurement signal 3 to an electrical measurement signal 4.

[0121] The device 100 further comprises a binarization unit 30, which converts the electrical measurement signal 4 into a first square wave signal 33 and the electrical reference signal 5 into a second square wave signal 35. The two square wave signals 33 and 35 are each essentially symmetrical square wave signals, whose pulse duration corresponds to half a period. The two square wave signals 33 and 35 therefore have a duty cycle of 50%.

[0122] In the embodiment shown in Figure 1, the binarization unit 30 for the electrical measurement signal 4 and the electrical reference signal 5 each comprises a comparator 32, 34 and a filter device 25, 27 with a low-pass filter 26, 28.

[0123] In the following phase comparison unit 38 of the device 100, the phase difference 6 is then determined by means of a phase comparison of the two rectangular signals 33, 35.

[0124] A pulse-width modulated signal is generated by means of a logic module 36, from which the phase difference 6 is extracted. For this purpose, the phase comparison unit 38 of the embodiment shown in Figure 1 comprises an exclusive-OR component as a logic module 36, a component 39 for phase calculation, and a clock generator 40. The phase difference 6 is therefore determined by means of an exclusive-OR (XOR) operation of the two square wave signals 33, 35. The square wave signal 35, converted from the electrical reference signal 5, is combined with a clock signal 7, which has a frequency shift relative to the electrical reference signal 5.

[0125] The receiving unit 21, the binarization unit 30, and the phase comparison unit 38 can be integrated in a compact form within the device 100. At least the phase comparison unit 38 can advantageously be integrated into a component 110, for example, an FPGA component 110.

[0126] According to the proposed method, the optical beat signal 3 of the two optical signals 1, 2 is generated by the heterodyne optical arrangement and converted into an electrical voltage signal 4 by the photodetector 22 and the transimpedance amplifier 24. The second output signal is the electrical reference signal 5, with which the second optical signal 2 is generated, for example, by the function generator 14.

[0127] Both signals 4 and 5 are sinusoidal. These two signals 4 and 5 are converted into signals 33 and 35, which consist of logic states, in order to be processed further. The shape of these signals 33 and 35 corresponds to an essentially symmetrical square wave, whose pulse duration corresponds to half a period, thus each having a duty cycle of, for example, 50%. Furthermore, the frequency of these square wave signals 33 and 35 corresponds to the frequency of the respective output signals 4 and 5. For other signals, e.g., highly nonlinear ones, the duty cycle may deviate from 50%.

[0128] For each of the 80 channels (see Figure 2) of an optical beam, a circuit with a comparator 32 as its core element can be used. The unaltered sinusoidal oscillation of the electrical signal 4 is fed to the first input of the comparator 32. The second input also receives this sinusoidal oscillation, but it is first low-pass filtered. This ensures that the signal is converted into the form of a substantially symmetrical square wave 33, regardless of its amplitude or DC component. The output of the comparator 32 can then be directly connected electrically to a phase comparator 38.

[0129] The proposed method can be advantageously used for targeted energy transfer. This enables power scaling of lasers with high beam quality, particularly through coherent laser coupling. Optical disturbances can also be advantageously compensated.

[0130] The proposed method requires, in addition to an element in which the logic and other digital operations of the phase comparison unit 38 can be implemented (e.g., an FPGA 110), only one analog comparator 32 per channel and a filter device 25 (e.g., a low-pass filter 26) to determine the phase difference from a sinusoidal measurement signal 4 to a reference signal 5, calculate a setpoint, and output it in analog form. Thus, compared to conventional arrangements, neither a multi-bit ADC nor a dedicated multi-bit DAC is required. Furthermore, no additional components such as RF mixers or dedicated logic modules are used.

[0131] Since the low-pass filters 26 can be implemented as RC circuits and thus each only have one resistor and one capacitor, the device 100 required for the proposed method needs only a few components per channel in addition to a logic element, if this setup is not to be implemented as an integrated circuit. A typical FPGA is capable of processing the logic operations of 80 out of several hundred channels simultaneously. Since the other components per channel have very small dimensions (less than or equal to 1 mm), 2 Since these components are available, the scaling problems described above are significantly reduced. It is also conceivable to implement the entire system, including the photodiodes 22 and the transimpedance amplifiers 24, as an integrated circuit, which further increases scalability.

[0132] In such a phase detector, the two signals 33 and 35 can each be fed to the input of the XOR gate 36. The XOR gate 36 outputs a one whenever only one of the two signals 33 and 35 is positive. In all other cases, it outputs a zero. Accordingly, the output of the XOR gate 36 outputs a one for each period of one signal 33 and 35, always for the time that one signal 33 and 35 is shifted relative to the other signal 35 and 33. If a low-pass filter is applied to the resulting digital signal, the resulting output signal represents the phase difference between the two input signals 33 and 35.

[0133] Figure 2 shows the device 100 of the optical arrangement 200 according to Figure 1. The device 100 comprises the receiver unit 21 with photodiode 22 and transimpedance amplifier 24, the binarization unit 30 (dashed box) with the two comparators 32, 34 and the two low-pass filters 26, 28, the phase comparison unit 38 (dashed box) with the XOR gate as logic element 36, the phase calculation component 39 and the clock generator 40.

[0134] The receiving unit 21, the comparator 32 with low-pass filter 26 for processing the electrical measurement signal 4, as well as the XOR gate 36 and the phase calculation component 39 are provided for each channel of a number of 80 channels of an optical beam.

[0135] In this way, the illustrated device 100 can advantageously be designed for the simultaneous determination of a plurality of phase differences 6 of a plurality of optical measurement signals 3.

[0136] The phase comparison unit 38 with the XOR gates 36 and the phase calculation component 39 for the individual channels as well as the clock generator 40 can advantageously be integrated in a component 110, in particular an FPGA component 110, in order to make the device 100 as compact and cost-effective as possible.

[0137] To achieve a sufficiently high phase resolution without requiring very high sampling rates, the reference signal 5 and the measurement signal 4 can be sampled using bistable elements with a clock signal 40 whose clock is slightly frequency-shifted relative to the reference signal 5. Since the input signals 4 and 5 are asynchronous to the generated clock signal 7, a chain of bistable elements can also be used to reduce metastable states. A suitable sampling frequency for this clock signal 7 can be determined as follows: sampling frequency = (1 - (N + 1)- 1 )

[0138] Here, fref is the frequency of the reference signal 5 and N is the desired phase resolution as an integer. The XOR operation between the previously sampled values ​​can be performed at the same clock signal 7.

[0139] Alternatively, the proposed device 100 can also be used to decode at least one phase-coded data transmission. The measurement signal 3, 4 can, for example, be generated using the so-called phase-shift keying method. In this way, a scalable number of phase-coded data signals can be fed as measurement signals 3 via photodiodes, or directly as a voltage signal in the form of an electrical measurement signal 4, into binarization units 30 to generate square wave signals 33. If the measurement signal 4 is transmitted purely electrically, the photodiode 22 and the transimpedance amplifier 24 in Figure 2 can be omitted.

[0140] The reference signal 5 can be generated locally and is not permanently linked to a transmitting side of the measurement signal 3, 4. Alternatively, the reference signal 5 can be transmitted separately or extracted from the data signal 3, 4.

[0141] Figure 3 shows a schematic representation of the sampling process using a clock signal 7 to determine the phase difference 6 between the optical measurement signal 3 and the electrical reference signal 5. The clock signal 7 (dashed line) provided by the clock generator 40 is plotted for sampling the square wave signal 35 (solid line) generated from the reference signal 5 as a function of time 90. Figure 3 illustrates the sampling process using an example signal. The sampling is performed in such a way that six different phase ranges can be distinguished. The sampling times 41 are marked with arrows.

[0142] Figure 4 shows a system representation of a phase comparison unit 38 according to an embodiment of the invention.

[0143] The inputs of the XOR component 36 are coupled to a first bistable element 60, in particular a D flip-flop, for the electrical measurement signal 4 and to a second bistable element 61, in particular a D flip-flop, for the electrical reference signal 5, wherein the first and the second bistable element 60, 61 are triggered by the clock generator 40.

[0144] An output of the XOR component 36 is electrically coupled to a first counter 50, and the clock generator 40 is electrically coupled to a second counter 55. The first and second counters 50 and 55 are coupled to a bistable element 62, in particular a D flip-flop, at whose output Q the phase difference 6 is applied. The first counter 50 and the second counter 55 each have a plurality of cascaded bistable elements 51, 52, 53, 54; 56, 57, 58, 59.

[0145] The result of the XOR operation is then fed into the first counter 50. The counting range of this first counter 50 can advantageously correspond to the desired phase resolution N or a multiple thereof. In parallel to this first counter 50, the second counter 55 is operated by a clock generator 40, which is also used for sampling. The two counters 50 and 55 can conveniently have the same counting range. When the second counter 55, which is operated by the clock generator 40, overflows, the current count of the first counter 50 after the XOR gate 36 is adopted as the new phase value 6.

[0146] The value determined for the phase difference 6, however, contains no information about the sign of the phase difference 6. Since the XOR operation is always performed at fixed phase values ​​due to the fixed relationship between the clock signal 7 and the reference signal 5, it is also possible to determine whether the XOR gate 36 output positive values ​​in the first or second half of the oscillation period, or at all. With this information, the sign can then be directly derived.

[0147] The resolution can also be increased without adjusting the sampling rate. This can be achieved by exploiting frequency fluctuations of the clock signal 7 and the averaging property of the counter structure. However, the necessary increase in the counting range of the first counter 50 reduces the frequency at which new phase values ​​6 are available. This, in turn, can be increased by multiplying the initially calculated sampling rate. Correspondingly to the multiplication of the sampling rate, the counting range can then also be multiplied.

[0148] Figure 5 shows a block diagram of the device 100 for determining and controlling a phase difference 6 between an optical measurement signal 3 and an electrical reference signal 5 according to an embodiment of the invention.

[0149] The optical measurement signal 3 is received by the receiver unit 21 and converted into the electrical measurement signal 4. This is then fed to a first part of the binarization unit 30. A second part of the binarization unit 30 receives the electrical reference signal 5 from the function generator 14. The outputs of the two parts of the binarization unit 30 feed the two square wave signals 33 and 35 to the phase comparison unit 38. From the phase comparison unit, the phase value 6 can be fed to the control unit 42 and a phase actuator 46.

[0150] The dashed outline of the device 100 can advantageously be implemented multiple times in the same way, corresponding to the number 80 of channels of the beam of optical measurement signals 3.

[0151] Figure 6 shows a device 100 for determining and controlling a phase difference 6 between an optical measurement signal 3 and an electrical reference signal 5 according to a further embodiment of the invention.

[0152] The receiving unit 21 and the two parts of the binarization unit 30 correspond in Figure 6 to the embodiment shown in Figure 2, but the phase comparison unit 38 is designed differently.

[0153] The phase comparison unit 38 comprises a logic module 63, in particular an SR flip-flop, which controls an optical emitter 64, and a camera 66, which determines the intensity of the optical emitter 64. The phase difference 6 is determined from the two square wave signals 33 and 35 by means of the bistable element 63, with the optical emitter 64 being operated by an output signal of the bistable element 63. The intensity of the optical signal emitted by the optical emitter 64 is proportional to the phase difference 6. The camera 66 determines the intensity of the optical signal. The output signal of the camera then corresponds to the phase value 6. Furthermore, the device 100 comprises a control unit 42. From the phase difference 6, a manipulated variable for controlling the phase difference 6 between the optical measurement signal 3 and the electrical reference signal 5 in a phase actuator 46 can be determined by means of the control unit 42.

[0154] In this further embodiment, a circuit almost identical to the previously described examples, with the bistable element as a logic comparator 32 and the low-pass filter 26, can be used to modulate the pulse width of the optical emitter 64, such as a light-emitting diode (LED) or a laser diode, for example, a vertical cavity surface-emitting laser (VCSEL). In this case, the pulse width encodes the phase difference 6 of the system. With a duplication of the optical emitters 64 in an alternating emission mode (on / off), (off / on) it can also be ensured that the sign is spatially encoded. The preferably synchronized camera 66 can thus record an optical signal that encodes the phase difference 6 of the element. This avoids a limitation of the scalability due to signal lines or crosstalk between the lines. In addition, a unique assignment of the signal sources is possible, and with the correct selection of the optical emitters 64, even technically sophisticated high-speed cameras can be used for long-wavelength laser radiation. If the laser source 10 emits at 1550 nm, for example, to generate the optical signals, the signals can be detected by the photodiodes and the secondary signal can then be output with LEDs at, for example, 850 nm. Thus, normal (silicon) cameras for 1550 nm can be used.Alternatively, the radiation emitted by the optical emitter 64 can be propagated back through an optical system to be controlled in order to directly provide an analog signal at the radiation source to be controlled, which is suitable for control.

[0155] The receiving unit 21, the part of the binarization unit 30 for determining the first rectangular signal 33, and the part of the phase comparison unit 38 with the bistable element 63 and the optical emitter 64 can again be implemented multiple times in the same manner, corresponding to the number of 80 channels of the beam of optical measurement signals 3. The connection between the control unit 42 and the phase actuator 46 is also implemented multiple times, corresponding to the number of 80 channels.

[0156] Figure 7 shows a device 100 for determining and controlling a phase difference 6 between an optical measurement signal 3 and an electrical reference signal 5 according to a further embodiment of the invention.

[0157] In this embodiment, the phase comparator 38 has a reference DC voltage input 48. The reference DC voltage 8 defines the DC component of the signals 33, 35, which are fed into the phase comparator 38. The reference DC voltage 8 is supplied to so-called bias tees, each consisting of an inductor 70, 71 and a capacitor 72, 73. In this setup, the DC component of the electrical measurement signal 4 from the transimpedance amplifier 24 is cut off, and the reference DC voltage 8 is applied to the signal 4 as a DC component. This arrangement ensures that the input signals 33, 35 correspond to a substantially symmetrical square wave, i.e., the low and high states of the square wave signal 33, 35 each have a duty cycle of 50%.

[0158] The input threshold and switch-off threshold for the other inputs of the module 110 are defined via the reference DC voltage input 48.

[0159] The reference DC voltage 8 is used to define the switch-on threshold or switch-off threshold when sampling the two signals 75, 76 in the phase comparison unit 38.

[0160] In this embodiment for the binarization of input signals 4 and 5, the input signals 4 and 5 can be modified to include an externally determined DC component. This DC component corresponds to the common reference DC voltage 8. This reference DC voltage 8 can also be fed into an FPGA 110 of the phase comparator unit 38. Here, this reference DC voltage 8 is used to define the voltage threshold that distinguishes between the high and low states of signals 75 and 76. One logic standard that supports this is, for example, high-speed transceiver logic (HSTL). As with the previously described binarization units 30, this arrangement ensures that the input signals 4 and 5 correspond to a substantially symmetrical square wave, i.e., the low and high states of the square wave each have a duty cycle of 50%.The two signals 75, 76 are fed to the XOR gate 36, whose output signal is fed to the phase calculation component 39.

[0161] The specified phase value 6 is supplied to the control unit 42. An output of the control unit 42 is electrically coupled to a delta-sigma modulator 44, which provides a one-bit-wide bitstream that is in turn filtered in a low-pass filter 68.

[0162] After the low-pass filter 68, an analog control variable 9 is available for controlling the phase of the optical measurement signal 3.

[0163] The receiving unit 21, the part of the binarization unit 30 for determining the first square wave signal 33, and the part of the phase comparison unit 38 with the XOR gate 36, the phase calculation component 39, the control unit 42, the delta-sigma modulator 44, and the low-pass filter 68 can again be implemented multiple times in the same way according to the number 80 channels of the beam of optical measurement signals 3.

[0164] The phase comparison unit 38 with the clock generator 40, the control unit 42 and the delta-sigma modulator 44 can advantageously be integrated in a component 110, in particular an FPGA component 110, in order to achieve the most compact and cost-effective design possible.

[0165] Figure 8 shows a device 100 for determining and controlling a phase difference 6 between an optical signal 3 and an electrical reference signal 5 according to a further embodiment of the invention. The embodiment shown in Figure 8 is a modification of the embodiment shown in Figure 7 with the reference DC voltage input 48. The phase comparison unit 38 is implemented in the configuration shown in Figure 4 with the two counters 50, 55. The electrical measurement signal 4 and the electrical reference signal 5 are sampled with the clock signal 7 by means of the bistable elements 60, 61, in particular D flip-flops. Optionally, the XOR operation in the XOR gate 36 can be clocked with the clock signal 7.

[0166] The result of the XOR operation is processed using the first counter 50. The clock signal 7 is processed using the second counter 55. When the second counter 55 overflows, the current value of the first counter 50 is used as the value of the phase difference 6.

[0167] The output of the phase comparison unit 38, which provides the phase difference 6, is electrically coupled to a control unit 42, as shown in Figure 7. This control unit 42 is coupled to a delta-sigma modulator 44, which outputs a bit stream that, after filtering in a low-pass filter 68, yields an analog manipulated variable 9. The analog manipulated variable 9 is then supplied to the phase actuator 46.

[0168] The receiving unit 21, the part of the binarization unit 30 for determining the first square wave signal 33, and the part of the phase comparison unit 38 with the XOR gate 36, the two counters 50, 55, the control unit 42, the delta-sigma modulator 44, the low-pass filter 68, and the phase actuator 46 can again be implemented multiple times in the same manner, corresponding to the number 80 channels of the beam of optical measurement signals 3. The clock generator 40 is only required once per module 110, in particular per device 100. In this way, a plurality of phase differences 6 of a plurality of optical measurement signals 3 of a plurality of beams of a beam can be determined and controlled simultaneously.

[0169] The phase comparison unit 38 with the clock generator 40, the control unit 42 and the delta-sigma modulator 44 can advantageously be integrated in a component 110, in particular an FPGA component 110, in order to achieve the most compact and cost-effective design possible.

[0170] Figure 9 shows a time course of a measurement signal 3 and a reference signal 5 with a determination of a phase difference 6 by evaluating measurement signal 3 and reference signal 5 via an XOR operation and via an asynchronously edge-triggered SR flip-flop.

[0171] In the top row of Figure 9, the measurement signal 3 (solid line) and the reference signal 5 (dashed line) are shown as a function of time 90 with a normalized amplitude from -1 to 1.

[0172] Below is shown the phase difference 6 between the two signals 3, 5, which varies between 0 and 2 TT.

[0173] The third row shows a square wave signal 91, which is the result of an XOR operation. The input signals for the XOR operation are the two signals 3 and 5, which were previously converted to square wave signals. The points on the dashed line 93 represent mean values ​​94 of a segment of the pulse-width modulated signal. These mean values ​​were each determined over a segment that is one period wide of the reference signal 5. The dashed line 93 is a linear interpolation between the individual mean values ​​94. The fourth row below shows a square wave signal 92, which is the result of an operation using an asynchronous, edge-triggered SR flip-flop. The input signals for the operation are the two signals 3 and 5, which were previously converted to square wave signals. The points on the dashed line 95 represent mean values ​​96 of a segment of the pulse-width modulated signal.These mean values ​​96 were each determined over a section that is one period wide of the reference signal 5. The dashed line 95 is a linear interpolation between the individual mean values ​​96.

[0174] The maximum of the dashed line 93 in the third row of the variant using the XOR gate arises because, with this type of phase evaluation, which only averages over a single window, the XOR gate can only evaluate a range from 0 to TT without a sign. Since the XOR operation is always performed at fixed phase values ​​due to the fixed relationship between the clock signal and the reference frequency, it is also possible to determine whether the XOR gate output positive values ​​in the first or second half of the oscillation period, or at all. With this information, the sign can then be directly derived.

[0175] An asynchronously edge-triggered SR flip-flop, whose results are shown in the fourth row, can directly resolve the 2 TT using this type of phase evaluation. Therefore, the dashed line 95 is a straight line. Reference symbol

[0176] 1. First optical signal

[0177] 2 second optical signal

[0178] 3 optical measurement signal

[0179] 4 electrical measurement signal

[0180] 5 Reference signal

[0181] 6 Phase difference

[0182] 7 Clock signal

[0183] 8 Reference DC voltage

[0184] 9 analog control variable

[0185] 10 Laser source

[0186] 12 Acousto-optic modulator

[0187] 14 Function Generator

[0188] 18 mirrors

[0189] 20 superposition unit

[0190] 21 receiving unit

[0191] 22 Photodiode

[0192] 24 Transimpedance Amplifiers

[0193] 25 Filter system

[0194] 26 Low-pass filter

[0195] 27 Filter system

[0196] 28 Low-pass filter

[0197] 30 Binary unit

[0198] 32 Comparator

[0199] 33 Rectangular signal

[0200] 34 Comparator

[0201] 35 square wave

[0202] 36 XOR gates

[0203] 38 Phase comparison unit

[0204] 39 Phase value calculation component

[0205] 40-stroke generator

[0206] 41 Sampling time Control unit Delta-sigma modulator Phase actuator Reference DC input First counter Bistable element Bistable element Bistable element Bistable element Second counter Bistable element Bistable element Bistable element Bistable element Bistable element Bistable element Bistable element Bistable element Optical emitter Camera Low-pass filter Coil Coil Capacitor Capacitor Signal Signal Number of channels Time Square wave XOR operation Square wave SR flip-flop Dashed line Average Dashed line Average Device FPGA component Optical arrangement

Claims

Claims 1. Method for determining a phase difference (6) between a measurement signal (4) and a reference signal (5), comprising at least Converting the measurement signal (4) into a first square wave signal (33) and the reference signal (5) into a second square wave signal (35), which are each essentially symmetrical square wave signals (33, 35) whose pulse duration corresponds to half a period; Determining the phase difference (6) by means of a phase comparison of the two square wave signals (33, 35), wherein a pulse-duration modulated signal is generated by means of a logic block (36, 63), from which the Phase difference (6) is extracted.

2. Method according to claim 1, wherein the phase difference (6) is determined by means of an exclusive-OR operation of the two rectangular signals (33, 35).

3. Method according to claim 1 or 2, wherein the output binary states of a logic operation of the logic block are digitally evaluated over a defined time period.

4. Method according to one of the preceding claims, wherein the pulse duration of the pulse duration modulated signal is directly detected.

5. Method according to one of the preceding claims, wherein the result of the operation of the logic block (36, 63) is processed by means of a first counter (50) and wherein the clock signal (7) is processed by means of a second counter (55), wherein, in the event of an overflow of the second counter (55), the current counter value of the first counter (50) is adopted as the value of the phase difference (6).

6. Method according to claim 1, wherein the pulse-duration modulated signal generated by the logic module (36, 63) is output as an optical signal.

7. Method according to one of the preceding claims, wherein the second square wave signal (35) converted from the reference signal (5) is linked to a clock signal (7) which has a frequency shift relative to the reference signal (5).

8. Method according to one of the preceding claims, wherein the electrical measurement signal (4) and the reference signal (5) are sampled with the clock signal (7) by means of bistable elements (60, 61), in particular D flip-flops, in particular by means of chained bistable elements (60, 61).

9. Method according to one of the preceding claims, wherein the phase difference (6) is determined from the two square wave signals (33, 35) by means of the logic block (36, 63), in particular an S R flip-flop.

10. Method according to claim 1, wherein at least one optical emitter (64) is operated with an output signal of the logic module (36, 63), the mean intensity of which is proportional to the phase difference (6), in particular wherein the intensity is determined by means of a camera (66).

11. Method according to one of the preceding claims, wherein the measurement signal (4) and the reference signal (5) are connected with a Reference DC voltage (8) is applied, the reference voltage (8) being used to define a common voltage threshold for the two square wave signals (33, 35).

12. Method according to one of the preceding claims, wherein a control variable for controlling the phase difference (6) of the optical measurement signal (3) is determined from the phase difference (6), in particular by means of a control unit (42), in particular wherein an analog control variable (9) for controlling the phase difference (6) of the optical measurement signal (3) is provided by means of a delta-sigma modulator (44).

13. Method according to one of the preceding claims, wherein a plurality of phase differences (6) of a plurality of measurement signals (4) is determined simultaneously, in particular wherein a plurality of phase differences (6) of a plurality of measurement signals (4) is controlled simultaneously.

14. Method according to any of the preceding claims, wherein the reference signal (5) is an electrical reference signal (5).

15. Method according to one of the preceding claims, wherein the phase difference (6) between an optical measurement signal (3) and the reference signal (5) is determined, wherein the optical measurement signal (3) is converted to an electrical measurement signal (4).

16. The method of claim 15, further comprising Generating a first optical signal (1 ) with a first frequency and a first phase using a laser source (10); Generating a second optical signal (2) with a second frequency and a second phase from the first optical signal (1) from the laser source (10) using the reference signal (5), wherein the second optical signal (2) has a frequency shift relative to the first optical signal (1); Superimposing the first optical signal (1) with the second optical signal (2) to form the optical measurement signal (3).

17. Device (100) for determining a phase difference (6) between a measurement signal (4) and a reference signal (5) using a method according to one of the preceding claims, comprising at least a binarization unit (30) which converts the measurement signal (4) into a first square wave signal (33) and the reference signal (5) into a second square wave signal (35), each of which is essentially a symmetrical square wave signal (33, 35) whose pulse duration corresponds to half a period, a phase comparison unit (38) which determines the phase difference (6) by means of a phase comparison of the two square wave signals (33, 35), and a logic module (36, 63) which generates a pulse duration modulated signal from which the phase difference (6) can be extracted.

18. Device according to claim 17, wherein the binarization unit (30) for the measurement signal (4) and the reference signal (5) each comprises a comparator (32, 34) and a filter device (25, 27), in particular a low-pass filter (26, 28).

19. Device according to claim 17 or 18, wherein the phase comparison unit (38) comprises an exclusive-OR gate (36) and a clock generator (40).

20. Device according to claim 19, wherein an output of the exclusive-OR gate (36) is electrically coupled to a first counter (50) and the clock generator (40) is electrically coupled to a second counter (55), wherein the first and the second counter (50, 55) are coupled to a bistable element (62), in particular a D flip-flop, at whose output (Q) the phase difference (6) is applied, in particular wherein the first counter (50) and the second counter (55) comprise a plurality of cascaded bistable elements (51, 52, 53, 54; 56, 57, 58, 59).

21. Device according to claim 19 or 20, wherein inputs of the exclusive-OR gate (36) are coupled to a first bistable element (60), in particular a D flip-flop, for the electrical measurement signal (4) and to a second bistable element (61), in particular a D flip-flop, for the reference signal (5), wherein the first and the second bistable element (60, 61 ) are triggered by the clock generator (40).

22. Device according to claim 17 or 18, wherein the phase comparison unit (38) comprises the logic component (36, 63), in particular an SR flip-flop, 23. Device according to claim 22, wherein the logic module (36, 63) controls at least one optical emitter (64), and the phase comparison unit (38) has a camera (66) which determines an intensity of the optical emitter (64).

24. Device according to one of claims 17 to 21, wherein the phase comparison unit (38) has a reference DC voltage input (48).

25. Device according to one of claims 17 to 24, further comprising a control unit (42) which determines a manipulated variable for controlling the phase difference (6) of the measurement signal (3), in particular wherein an output of the control unit (42) is electrically coupled to a delta-sigma modulator (44) which provides an analog manipulated variable (9) for controlling the phase difference (6) of the measurement signal (3).

26. Device according to one of claims 17 to 25, configured for the simultaneous determination of a plurality of phase differences (6) of a plurality of measurement signals (3), and / or designed to simultaneously control a plurality of phase differences (6) of a plurality of measurement signals (3).

27. Device according to one of claims 17 to 26, wherein at least the phase comparison unit (38) is integrated in a component (110), in particular an FPGA component (110), and / or wherein the control unit (42) and / or the delta-sigma modulator (44) is integrated in the component (110), in particular the FPGA component (110).

28. Device according to one of claims 17 to 27 for determining a phase difference (6) between an optical measurement signal (3) and the reference signal (5), further comprising a receiving unit (21) which converts the optical measurement signal (3) to an electrical measurement signal (4).

29. Optical arrangement (200) for determining a phase difference (6) between a first optical signal (1) and a second optical signal (2) using a method according to any one of claims 1 to 16, with a device according to any one of claims 17 to 28, further comprising a laser source (10) generating the first optical signal (1), a function generator (14) generating an electrical reference signal (5), an acousto-optic modulator (12) electrically coupled to the function generator (14) so ​​that a second optical signal (2) is generated which has a frequency shift relative to the first optical signal (1), and a superposition unit (20) which combines the first optical signal (1) with the second optical signal (2) to form an optical The measurement signal (3) is superimposed with the phase difference (6) of the two optical signals (1 , 2).

30. Optical arrangement (200) for determining and controlling a phase difference (6) between a first optical signal (1) and a second optical signal (2) with a method according to any one of claims 1 to 16, with a device according to any one of claims 17 to 28, further comprising a laser source (10) generating the first optical signal (1), a function generator (14) generating an electrical reference signal (5), an acousto-optic modulator (12) electrically coupled to the function generator (14) so ​​that a second optical signal (2) is generated which has a frequency shift relative to the first optical signal (1), a superposition unit (20) which superimposes the first optical signal (1) with the second optical signal (2) to form an optical measurement signal (3) with the phase difference (6) of the two optical signals (1, 2), a control unit (42),which determines a control variable for regulating the phase difference (6) of the two optical signals (1).

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