Converter circuit for converting DC to ac

Abstract

Embodiments of the invention relate to a converter circuit (100) for converting a DC to an AC. The converter circuit (100) comprises: a first half-bridge (110) comprising a first (112) and a second (112') high frequency switch, the first half-bridge (110) being coupled between a DC source (102) and a primary winding (142) of a transformer (140); a second half-bridge (120) comprising a first (122) and a second (122') high frequency switch, the second half-bridge (120) being coupled in parallel with a third half-bridge (130) comprising a first (132) and a second (132') low frequency switch, wherein the second half-bridge (120) and the third half-bridge (130) are coupled between a secondary winding (144) of the transformer (140) and an AC load (104); and a control device (150) configured to: control the first (112) and the second (112') high frequency switch of the first half-bridge (110) based on a first control signal (Ctrl_1) with a variable switching frequency, and control the first (122) and the second (122') high frequency switch of the second half-bridge (120) based on a second control signal (Ctrl_2) with a variable switching frequency, wherein the second control signal (Ctrl_2) is phase shifted in relation to the first control signal (Ctrl_1) with a variable phase shift φ.

Description

[0001] CONVERTER CIRCUIT FOR CONVERTING DC TO AC

[0002] TECHNICAL FIELD

[0003] Embodiments of invention relate to a converter circuit for converting a direct current (DC) to and alternating current (AC).

[0004] BACKGROUND

[0005] Currently, great attention is paid to Photovoltaic (PV) inverters due to the increased PV capacity and demand for renewable energy. In small-scale power systems, the use of micro-inverters for PV inverters is getting popular due to their ease of installation and maintenance. Moreover, micro-inverters generate high power even under partially shaded conditions, making them an attractive solution for small-scale PV systems. In micro-inverters DC-AC power conversion that is commonly performed using two-stage configurations. The two-stage topology has its limitations of high efficiency, lifetime, reliability, and power density due to the two-stage power conversion and large size of electrolytic capacitors. Therefore, single-stage converters can be a powerful candidate to satisfy these limitations.

[0006] The control variables optimization is the key element for increasing the efficiency and performance operation of converters such as high-power quality, dynamic tolerance, etc. in particular, for grid-connected applications.

[0007] SUMMARY

[0008] An objective of embodiments of the invention is to provide a solution which mitigates or solves the drawbacks and problems of conventional solutions.

[0009] Another objective of embodiments of the invention is to provide a converter circuit solution having high efficiency in DC-AC conversion.

[0010] The above and further objectives are solved by the subject matter of the independent claims.

[0011] Further embodiments of the invention can be found in the dependent claims.

[0012] According to a first aspect of the invention, the above mentioned and other objectives are achieved with a converter circuit for converting a direct current, DC, to an alternating current, AC, the converter circuit comprising: a first half-bridge comprising a first and a second high frequency switch, the first half-bridge being coupled between a DC source and a primary winding of a transformer; a second half-bridge comprising a first and a second high frequency switch, the second half-bridge being coupled in parallel with a third half-bridge comprising a first and a second low frequency switch, wherein the second half-bridge and the third half-bridge are coupled between a secondary winding of the transformer and an AC load; and a control device configured to: control the first and the second high frequency switch of the first half-bridge based on a first control signal with a variable switching frequency, and control the first and the second high frequency switch of the second half-bridge based on a second control signal with a variable switching frequency, wherein the second control signal is phase shifted in relation to the first control signal with a variable phase shift <p. An advantage of the converter circuit according to the first aspect is that the variable phase shift <p between the first and second half-bridge maintains soft switching for all switches thereby high efficiency achieved in the DC-AC conversion. Further, sinusoidal AC current is also maintained with very low distortion.

[0013] In an implementation form of a converter circuit according to the first aspect, the variable phase shift <p is based on an angular frequency co of a voltage of the AC load.

[0014] In an implementation form of a converter circuit according to the first aspect, the variable phase shift p is based on the angular frequency co of the voltage of the AC load and one or more multiples of the angular frequency co.

[0015] In an implementation form of a converter circuit according to the first aspect, the variable phase shift p is based on a sine function of the angular frequency co of the voltage of the AC load and the one or more multiples of the angular frequency co.

[0016] In an implementation form of a converter circuit according to the first aspect, the variable phase shift p is based on the sine function of the angular frequency co of the voltage of the AC load, the one or more multiples of the angular frequency co of the voltage of the AC load, and one or more constant values.

[0017] In an implementation form of a converter circuit according to the first aspect, the variable phase shift p is expressed as: s(sin(2 * a> * t))+ a2* abs(sin(4 « w « t)) + a3* abs(sin(6 « w « t)) + a4where alta2, a3and a4are constant values, abs is the absolute value and t is time.

[0018] An advantage with this implementation form is that a closed form expression is provided for the variable phase shift (p. Further, the variable phase shift (p is one of the control parameters that controls the amount of power transfer.

[0019] In an implementation form of a converter circuit according to the first aspect, the variable switching frequency is higher than lO kHZ.

[0020] An advantage with this implementation form is that the size of utilized magnetic components in the converter circuit may be reduced.

[0021] In an implementation form of a converter circuit according to the first aspect, the control device is configured to: control the first and the second low frequency switch of the third half-bridge based on a third control signal with a constant switching frequency.

[0022] An advantage with this implementation form is that the switching losses of the third half-bridge are negligible. Only conduction losses of one switch of the third half-bridge are considered.

[0023] In an implementation form of a converter circuit according to the first aspect, the constant switching frequency is a frequency of the voltage of the AC load.

[0024] In an implementation form of a converter circuit according to the first aspect, the frequency of the AC load is less than 60 Hz.

[0025] An advantage with this implementation form is that it covers all mains frequency ranges. In an implementation form of a converter circuit according to the first aspect, wherein the first half-bridge is a current fed half-bridge, the second half-bridge is a current fed half-bridge, and the third half-bridge is an unfolding half-bridge.

[0026] An advantage with this implementation form is that reduced switch count is achieved meaning that less number of switches may be used in the converter circuit compared to conventional solutions. Thereby, reducing the cost of producing the converter circuit and also high power density is achieved.

[0027] In an implementation form of a converter circuit according to the first aspect, wherein the first half-bridge comprises a first capacitor and a second capacitor coupled in parallel with the first and the second switch of the first half-bridge via a node arranged between the first capacitor and the second capacitor of the first halfbridge; and the second half-bridge comprises a first capacitor and a second capacitor coupled in parallel with the first and the second switch of the second half-bridge via a node arranged between the first capacitor and a second capacitor of the second half-bridge.

[0028] The capacitors may be film capacitors with long lifetime.

[0029] In an implementation form of a converter circuit according to the first aspect, the first half-bridge comprises an inductor coupled between a decoupling capacitor and the primary winding of the transformer via a node arranged between the first and the second switch of the first half-bridge.

[0030] An advantage with this implementation form is that the decoupling capacitor decouples the second harmonic pulsating power thus the PV current is maintained pure DC.

[0031] In an implementation form of a converter circuit according to the first aspect, the second half-bridge comprises a power transfer element coupled between the secondary winding of the transformer and the AC load via a node arranged between the first and the second switch of the second half-bridge.

[0032] An advantage with this implementation form is that the power transfer element minimizes the circulating current and guarantees soft switching thereby high efficiency is achieved. This also implies that higher switching frequency may be employed in the converter circuit. Higher switching frequency results in higher power density.

[0033] In an implementation form of a converter circuit according to the first aspect, the power transfer element is an inductor.

[0034] In an implementation form of a converter circuit according to the first aspect, the converter circuit comprises a filter circuit coupled between a node arranged between the first and the second switch of the second half-bridge and a node arranged between the first and the second switch of the third half-bridge, the filter circuit being coupled in parallel with the AC load.

[0035] An advantage with this implementation form is that the filter circuit guarantees sinusoidal load voltage in a stand-alone mode, and the sinusoidal grid current in grid connected mode.

[0036] In an implementation form of a converter circuit according to the first aspect, the DC source is a photovoltaic source. In an implementation form of a converter circuit according to the first aspect, the AC load is an AC load or an AC grid.

[0037] An advantage with this implementation form is that applications of the converter circuit may be extended.

[0038] In an implementation form of a converter circuit according to the first aspect, the transformer is a high frequency transformer.

[0039] An advantage with this implementation form is that the transformer may be of small size and of light weight.

[0040] Further applications and advantages of embodiments of the invention will be apparent from the following detailed description.

[0041] BRIEF DESCRIPTION OF THE DRAWINGS

[0042] The appended drawings are intended to clarify and explain different embodiments of the invention, in which:

[0043] - Fig. 1 shows a converter circuit according to embodiments of the invention;

[0044] - Fig. 2 shows in detail a converter circuit according to embodiments of the invention;

[0045] - Fig. 3 shows a controller block diagram for the stand-alone mode according to embodiments of the invention;

[0046] - Fig. 4 shows operating waveforms according to embodiments of the invention;

[0047] - Fig. 5 shows a controller block diagram for the grid-connected mode according to embodiments of the invention; and

[0048] - Fig. 6 shows ZVS turn on conditions in stand-alone mode according to embodiments of the invention.

[0049] DETAILED DESCRIPTION

[0050] Single-stage converters are current-fed or voltage-fed. In voltage-fed converters a high winding ratio is necessary since the boosting action is performed only by the high frequency transformer (HFT). However, in current-fed converters with switched inductors an active boosting action is attained with relatively low transformer winding ratio. Moreover, current-fed converters decrease the grid current ripple as well as capacitor size. For this reason, current-fed converters are suitable for wide voltage range application. In addition, current-fed topologies enable the interleaving of two or multiple phases. Hence, a good grid current quality is obtained using simple structure and reduced component current stress.

[0051] Furthermore, in voltage-fed topologies zero voltage switching (ZVS) is achieved based on the energy stored in the leakage inductor of the HFT. On the other hand, current-fed topologies offer more flexibility to achieve soft-switching since the ZVS depends on switched inductor and HFT leakage inductance. AC-DC converters employing the conventional phase-shift modulation (PSM) exhibits soft-switching failure under wide voltage and load ranges since it achieves full-range ZVS only at a unity conversion ratio. Therefore, improved modulation schemes with increased degrees of freedom are introduced to facilitate optimal ZVS operation of the AC-DC converters.

[0052] Despite various efforts of securing the ZVS through employing multiple phase shifts and frequency modulation, the aforementioned designs and schemes are complicated to implement in practice, and the controllers are susceptible to instability because of the heavier calculations and numerous sensed parameters.

[0053] Therefore, it is herein presented a single-stage half-bridge topology employing a control device / system being promising in terms of efficiency and power density due to the reduced switch and magnetic component counts and the maintained ZVS-tum ON for all HF switching devices of the converter. Fig. 1 therefore shows a converter circuit 100 for converting a DC to an AC according to embodiments of the invention. The herein disclosed converter circuit 100 comprises a first half-bridge 110 comprising a first 112 and a second 112' high frequency switch. The first half-bridge 110 is coupled between a DC source 102 and a primary winding 142 of a transformer 140. The converter circuit 100 also comprises a second half-bridge 120 comprising a first 122 and a second 122' high frequency switch. The second half-bridge 120 is coupled in parallel with a third half-bridge 130 comprising a first 132 and a second 132' low frequency switch. The second half-bridge 120 and the third half-bridge 130 are coupled between a secondary winding 144 of the transformer 140 and an AC load 104.

[0054] The converter circuit 100 further comprises a control device 150 which is configured to control the first 112 and the second 112' high frequency switch of the first half-bridge 110 based on a first control signal Ctrl_l with a variable switching frequency, and control the first 122 and the second 122' high frequency switch of the second half-bridge 120 based on a second control signal Ctrl_2 with a variable switching frequency. The second control signal Ctrl_2 is phase shifted in relation to the first control signal Ctrl_l with a variable phase shift <p.

[0055] Fig. 2 shows a converter circuit 100 more in detail according to embodiments of the invention. The shown topology in Fig. 2 is based on a Dual Active Bridge (DAB) concept between the primary winding 142 and the secondary winding 144 of the transformer 140 but other suitable converter topologies may be used. The transformer 140 of the converter circuit 100 is assumed to be a high frequency transformer which means that a small and lightweight transformer may be used. As a result, high power density is achieved.

[0056] The first half-bridge 110 is a current fed half-bridge, the second half-bridge 120 is a current fed half-bridge, and the third halfbridge 130 is an unfolding half-bridge according to embodiments of the invention. Furthermore, in embodiments of the invention, it is assumed that the DC source 102 is a PV source. As for load, the AC load may be an AC load 104 in which the converter circuit 100 is operating in a so-called stand-alone mode or an AC grid 104' in which the converter circuit 100 is operating in a so-called grid-connected mode. Thus, the converter circuit 100 may either operate in the stand-alone mode or in the grid-connected mode according to embodiments of the invention.

[0057] In the disclosed topology in Fig. 2, the first half-bridge 110 comprises a first capacitor 114 and a second 114' capacitor coupled in parallel with the first 112 and the second 112' switch of the first half-bridge 110 via a node denoted b arranged between the first capacitor 114 and the second 114' capacitor of the first half-bridge 110. The second half-bridge 120 comprises a first capacitor 124 and a second capacitor 124' coupled in parallel with the first 122 and the second 122' switch of the second halfbridge 120 via a node denoted d arranged between the first capacitor 124 and a second capacitor 124' of the second half-bridge 120. Exemplary values for the first capacitor 114 and the second 114' capacitor of the first half-bridge 110 and the first capacitor 124 and the second capacitor 124' of the second half-bridge 120 may be in the micro-Farad range.

[0058] Moreover, the first half-bridge 110 further comprises an inductor 116 which is coupled between a decoupling capacitor 106 and the primary winding 142 of the transformer 140 via a node denoted a arranged between the first 112 and the second 112' switch of the first half-bridge 110. The decoupling capacitor 106 may have a value in the milli-Farad range. The second halfbridge 120 on the other hand comprises a power transfer element 126 coupled between the secondary winding 144 of the transformer 140 and a filter circuit 160 via a node c arranged between the first 122 and the second 122' switch of the second half-bridge 120. In embodiments of the invention, the power transfer element 126 is an inductor.

[0059] Thus, in embodiments of the invention, at the DC side, the active current-fed first half-bridge 110 is made up of inductor 116, first 112 and a second 112' high frequency switches and first 114 and second 114' capacitors which are interfaced with the AC sidel04 through an HF link (HFL). The first 114 and a second 114' capacitors of the first half-bridge 110 may be small film capacitors, while the electrolytic decoupling capacitor 106 of few milli-farads decouples the second harmonic pulsating power. The HFL consists of a small HF transformer with a series inductor 126 acting as a power transfer element. The power transfer element 126 may be a combination of a small discrete inductor and the leakage inductance of the transformer 140. At the AC load side, two line-frequency first 132 and second 132' low frequency switches make up the unfolding third half-bridge 130. First 122 and second 122' high frequency switches, and small first 124 and second film capacitors 124' make up the current- fed second half-bridge 120.

[0060] Therefore, the adopted PV converter circuit 100 shown in Fig. 2 comprises of current-fed dual active half-bridges, i.e., the first 110 and second 120 half-bridges, with an unfolding half-bridge, i.e., the third half-bridge 130. Hence, the disclosed topology uses four high switching frequency switches, i.e., 112, 112', 122, 122', in the first 110 and second 120 half-bridges and two low frequency switches, i.e., 132, 132' in the third half-bridge 130.

[0061] Thus, in embodiments of the invention the variable switching frequency for switches of the first 110 and second 120 halfbridge is higher than 10 kHZ, while the frequency of the AC load 104 is less than 60 Hz which is the same as the switching frequency of the switches of the third half-bridge 130. The control device 150 controls the first 132 and the second 132' low frequency switch of the third half-bridge 130 based on a third control signal Ctrl_3 with a constant switching frequency as illustrated in Fig. 2. As mentioned, the constant switching frequency is a frequency of the voltage of the AC load 104.

[0062] Furthermore, a filter circuit 160 may be coupled between a node denoted c arranged between the first 122 and the second 122' switch of the second half-bridge 120 and a node denoted e arranged between the first 132 and the second 132' switch of the third half-bridge 130. The filter circuit 160 is coupled in parallel with the AC load 104 in embodiments of the invention.

[0063] In the standalone mode, the converter circuit 100 is connected to the AC load 104 through the filter circuit 160 in the form of a LC filter made up of inductor Li and capacitor Ci. However, in the grid-connected mode the converter circuit 100 is connected to the AC grid 104' through a LCL filter made up of the filter circuit 160 and an inductor Lg, where inductor Lgis coupled between the filter circuit 160 and the AC grid 104'. The values of the inductors in the converter circuit 100 depend on the converter application.

[0064] In embodiments of the invention, the first half-bridge 110 is switched with 50% duty cycle meaning that switches 112 and 112' commutates in complimentary manner with 50% duty cycle. As a result, bidirectional symmetrical core excitation at the DC side is achieved. Similarly, the second half-bridge 120 is switched with 50% duty cycle meaning that switches 122 and 122' commutate in complimentary manner with 50% duty cycle.

[0065] For grid / load voltage positive half-cycle switch 132' of the third half-bridge 130 is turned ON. However, for grid / load voltage negative half-cycle switch 132 of the third half-bridge 130 is turned ON which means that the unfolding is done with a single switch of the third half-bridge 130 conducting during each half-cycle. Hence, the disclosed topology is promising in terms of conduction losses.

[0066] Fig. 3(a) shows a simplified electrical model of the converter circuit 100 in the stand-alone mode, while Fig. 3(b) shows a control block diagram for the adopted single-stage power conversion circuit for a PV micro-inverter in stand-alone mode i.e., when the DC source 102 is a PV source. Fig. 4(a) to Fig. 4(e) show operating waveforms according to embodiments of the invention. Table 1 gives the definitions of terms used in Fig. 3(b). Table. 1

[0067] An objective of the control device 150 in the stand-alone mode is to feed the AC load 104 with a desired sinusoidal voltage with specified d, q components and angle A.

[0068] The control device 150 comprises two control variables, the switching frequency and the outer phase-shift which is herein also denoted the variable phase shift. The secondary AC source side is phase-shifted with respect to the primary DC source side which means that second control signal Ctrl_2 is phase shifted in relation to the first control signal Ctrl_l with the variable phase shift

[0069] Furthermore, the control device 150 uses two proportional integral (PI) controllers, i.e., PI controller and PI controller. Inverse park transformation is applied where the two signals out of the two PI controllers (d controller and q controller) are considered as d and q and components, respectively. It is worth mentioning that <: / > <» / ,„„ v and are the upper and lower limits of PI controller whereas are the upper and lower limits of PI controller. The signal obtained from the inverse park transformation is sinusoidal in shape and its absolute value is calculated and then is subtracted from the absolute value of A feedforward term FFVthat represents the minimum switching frequency is included in the control device 150.

[0070] Hence, in the stand-alone mode the switching frequency control signal is plotted in Fig. 4(d) and defined as:

[0071] Therefore, in embodiments of the invention, the variable phase shift control signal is based on an angular frequency of a voltage of the AC load 104. This may imply that the variable phase shift is dependent on or is a function of the angular frequency Thus, the variable phase shift may beafunction of the angular frequency and one or more multiples of the angular frequency

[0072] In a particular example, the variable phase shift control signal is based on a sine function of the angular frequency and the one or more multiples of the angular frequency In this respect, one or more constant values may also be considered for the variable phase shift Thus, in embodiments of the invention, the variable phase shift (pv.CLrt is given by the equation below and is plotted in Fig. 4(c),

[0073] Vvfitri = sin2( <n « t) f a abs(sin(2 * a> * t))+ a2* abs(sin(4 * a> * t)) + a3* abs(sin(6 * a> * t)) + a4where sin is the sine function, a1;a2, a3and a4are constant values, abs is the absolute value, and t is the time variable.

[0074] In the carrier generation block of the control device 150, using the two voltage control signals fvptriand <pv_ciri, two variable frequency, phase-shifted <p carrier functions with amplitude vcarare generated. For 50% duty cycle generation the compare value is 0.5vCar.

[0075] In order to verify the performance of the present solution on the particular micro-inverter topology shown in Fig. 2, a simulation test was performed considering the following specifications: output power = 300W, load voltage VL = 220 VAC, load voltage frequency ft = 50 Hz, PV voltage Vpv= 35~65V. The main key waveforms are given in Fig. 4. The continuous conduction mode (CCM) operation of the converter circuit 100 is proven in the boost inductor current ILPV waveform in Fig. 4(e). ILPV mainly include is DC and second harmonic components and switching frequency components.

[0076] In Fig. 4(b), the primary side voltage r,, / , of the transformer 140 is two-level low-frequency-free waveform. The 100Hz envelope of Vcd in Fig. 4(b) implies the small size of the film clamping capacitors 124 and 124'. Moreover, the voltage vcd includes the switching frequency and its multiples. Therefore, a lightweight HF transformer is used. Thus, the disclosed converter topology is promising in terms of power density. The load current and voltage in Fig. 4(a) is sinusoidal with Total Harmonic distortion (THD) < 5% for power e [40w, 350w] .

[0077] Fig. 5(a) shows a simplified electrical model of the overall system in grid-connected mode while Fig. 5(b) shows a control block diagram for the adopted single-stage power conversion circuit for PV micro-inverter in grid-connected mode. Table 2 gives the definitions used in Fig. 5(b). Fig. 6 shows ZVS turn ON conditions for the converter circuit 100 according to embodiments of the invention. Along with the adopted switching scheme, some design considerations need to be taken into account in order to maintain the ZVS turn ON for the switches of the first half-bridge 110. The soft commutation of switches 112 and 112' depends on the relationship between the input inductor current ILPand the current i.,r!as illustrated in Fig . 6. The ZVS turn ON condition for switch 112' is ILPV - iPri < 0 and its ZVS turn ON current is izvs_s2_dc = ILPV - iPri. The ZVS turn ON for switch 112' is inherently achieved since it operates as a synchronous rectifier.

[0078] An obj ective of the control device 150 in the grid-connected mode is to inj ect a good current quality to the AC grid 104 ' which * may be understood as a sinusoidal current with a very low harmonic distortion. It is worth to mention that the term ' is calculated in a maximum power point tracking (MPPT) algorithm. Similar to the control device 150 in the stand-alone mode, the control device 150 uses two control variables, the switching frequency fi,ciri and the outer phase-shift <pt, ctri. In grid-connected mode the controller block diagram for the control device 150 is not detailed since the processes are similar to the one in the stand-alone mode.

[0079] Table. 2

[0080] Finally, it should be understood that the invention is not limited to the embodiments described above, but also relates to and incorporates all embodiments within the scope of the appended independent claims.

Claims

CLAIMS1. A converter circuit (100) for converting a direct current, DC, to an alternating current, AC, the converter circuit (100) comprising: a first half-bridge (110) comprising a first (112) and a second (112') high frequency switch, the first half-bridge (110) being coupled between a DC source (102) and a primary winding (142) of a transformer (140); a second half-bridge (120) comprising a first (122) and a second (122') high frequency switch, the second half-bridge (120) being coupled in parallel with a third half-bridge (130) comprising a first (132) and a second (132') low frequency switch, wherein the second half-bridge (120) and the third half-bridge (130) are coupled between a secondary winding (144) of the transformer (140) and an AC load (104); and a control device (150) configured to: control the first (112) and the second (112') high frequency switch of the first half-bridge (110) based on a first control signal (Ctrl_l) with a variable switching frequency, and control the first (122) and the second (122') high frequency switch of the second half-bridge (120) based on a second control signal (Ctrl_2) with a variable switching frequency, wherein the second control signal (Ctrl_2) is phase shifted in relation to the first control signal (Ctrl_l) with a variable phase shift <p.

2. The converter circuit (100) according to claim 1, wherein the variable phase shift <p is based on an angular frequency co of a voltage of the AC load (104).

3. The converter circuit (100) according to claim 2, wherein the variable phase shift <p is based on the angular frequency co of the voltage of the AC load (104) and one or more multiples of the angular frequency co.

4. The converter circuit (100) according to claim 3, wherein the variable phase shift <p is based on a sine function of the angular frequency co of the voltage of the AC load (104) and the one or more multiples of the angular frequency co.

5. The converter circuit (100) according to claim 4, wherein the variable phase shift <p is based on the sine function of the angular frequency co of the voltage of the AC load (104), the one or more multiples of the angular frequency co of the voltage of the AC load (104), and one or more constant values.

6. The converter circuit (100) according to claim 5, wherein the variable phase shift <p is expressed as:<p = sin2( <n « t) + aj * abs(sin(2 * a> * t))+ a2* abs(sin(4 * a> * t)) + a3* abs(sin(6 * a> * t)) + a4where a , . a2. a3and a4are constant values, abs is the absolute value, and t is time.

7. The converter circuit (100) according to any one of the preceding claims, wherein the variable switching frequency is higher than 10 kHZ.

8. The converter circuit (100) according to any one of the preceding claims, wherein the control device (150) is configured to: control the first (132) and the second (132') low frequency switch of the third half-bridge (130) based on a third control signal (Ctrl_3) with a constant switching frequency.

9. The converter circuit (100) according to claim 8, wherein the constant switching frequency is a frequency of the voltage of the AC load (104).

10. The converter circuit (100) according to claim 9, wherein the frequency of the AC load (104) is less than 60 Hz.

11. The converter circuit (100) according to any one of the preceding claims, wherein the first half-bridge (110) is a current fed half-bridge, the second half-bridge (120) is a current fed half-bridge, and the third half-bridge (130) is an unfolding half-bridge.

12. The converter circuit (100) according to any one of the preceding claims, wherein the first half-bridge (110) comprises a first capacitor (114) and a second (114') capacitor coupled in parallel with the first (112) and the second (112') switch of the first half-bridge (110) via a node (b) arranged between the first capacitor (114) and the second (114') capacitor of the first half-bridge (110); and the second half-bridge (120) comprises a first capacitor (124) and a second capacitor (124') coupled in parallel with the first (122) and the second (122') switch of the second half-bridge (120) via a node (d) arranged between the first capacitor (124) and a second capacitor (124') of the second half-bridge (120).

13. The converter circuit (100) according to claim 12, wherein the first half-bridge (110) comprises an inductor (116) coupled between a decoupling capacitor (106) and the primary winding (142) of the transformer (140) via a node (a) arranged between the first (112) and the second (112') switch of the first half-bridge (110).

14. The converter circuit (100) according to claim 12 or 13, wherein the second half-bridge (120) comprises a power transfer element (126) coupled between the secondary winding (144) of the transformer (140) and the AC load (104) via a node (c) arranged between the first (122) and the second (122') switch of the second half-bridge (120).

15. The converter circuit (100) according to claim 14, wherein the power transfer element (126) is an inductor.

16. The converter circuit (100) according to any one of the preceding claims, comprising a filter circuit (160) coupled between a node (c) arranged between the first (122) and the second (122') switch of the second half-bridge (120) and a node (e) arranged between the first (132) and the second (132') switch of the third half-bridge (130), the filter circuit (160) being coupled in parallel with the AC load (104).

17. The converter circuit (100) according to any one of the preceding claims, wherein the DC source (102) is a photovoltaic source.

18. The converter circuit (100) according to any one of the preceding claims, wherein the AC load (104) is an AC load or an AC grid.

19. The converter circuit (100) according to any one of the preceding claims, wherein the transformer (140) is a high frequency transformer.

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