Power converter and power conversion system

Abstract

This power converter is provided with a first resonance inductor LA and a second resonance inductor LB connected to a leg A and a leg B, wherein: four or more switching elements are connected in series, and further, one or more resonance capacitors CresoA and resonance capacitors CresoB are connected, thereby enabling a plurality of levels of voltage to be generated; and negative side ends and positive side ends of the leg A and the leg B are connected to input / output terminals on a high voltage side, high frequency voltage terminals are connected to one ends of the first resonance inductor LA and the second resonance inductor LB, and the other ends of the first resonance inductor LA and the second resonance inductor LB are connected to input / output terminals on a low voltage side. The power converter can perform two or more types of operations having different characteristics between a first mode in which soft switching operation is performed by resonating the first resonance inductor LA and the second resonance inductor LB with the resonance capacitors CresoA and the resonance capacitors CresoB, and a second mode in which chopper operation is performed without resonating the first resonance inductor LA and the second resonance inductor LB with the resonance capacitors CresoA and the resonance capacitors CresoB.

Description

Power converters and power conversion systems

[0001] The present invention relates to a power converter and a power conversion system.

[0002] Boost converters are used in the powertrains of various types of electric vehicles. For example, in hybrid electric vehicles (HEVs) and plug-in hybrid electric vehicles (PHEVs), they boost battery voltage and supply it to the inverter that drives the motor. Because of the diverse operating conditions depending on acceleration and speed, boost converters have various requirements depending on the operating conditions. For example, during normal driving with little acceleration or deceleration, it is desirable to operate at an optimal boost ratio that minimizes system loss to reduce fuel and power consumption, requiring variable boost operation and high efficiency. Furthermore, during full acceleration, maximum output operation is required to ensure the necessary acceleration force. Because maximum output operation lasts for a short period of time, it has little impact on fuel and power consumption, but low loss and high cooling capacity are required to prevent overheating and damage to power devices.

[0003] High-power resonant switched capacitor converters are being considered for use in heavy-load electric mobility vehicles such as airplanes and trucks (Non-Patent Document 1). These high-power resonant switched capacitor converters perform bidirectional power conversion between 800 V and 1600 V at a high output of 500 kW. Soft switching using LC resonance enables highly efficient operation with reduced switching losses even at high frequencies of 100 kHz, with an estimated efficiency of 98.7% at 500 kW output.

[0004] Also, a three-level boost converter using a flying capacitor system has been proposed (Non-Patent Document 2), which uses legs consisting of four switching elements capable of generating three-level voltages. This technology allows the voltage applied to the boost inductor to be set to three levels: 0, Vo / 2, and Vo. This reduces current ripple compared to a conventional boost chopper that uses two levels: 0 and Vo, thereby achieving higher efficiency and a more compact inductor. Furthermore, because the applied voltage to the power device is halved, lower-voltage elements can be used.

[0005] Also, in a two-phase interleaved chopper circuit, a configuration in which the inductors of each phase are magnetically coupled inductors is known (Non-Patent Document 3). Appropriate magnetic coupling reduces current ripple, achieving high efficiency, and integrating the two elements allows for fewer components and a smaller size.

[0006] X. Liu, M. Qiu, K. Hobbs, A. Dahneem, H. Meng and D. Cao, "Experimental Verification of 500kW Resonant Switched-Capacitor Converter for Electric Trucks and Electric Aircraft Application," 2024 IEEE Applied Power Electronics Conference and Exposition (APEC), Long Beach, CA, USA, 2024, pp. 830-837, doi: 10.1109 / APEC48139.2024.10509275.H. Keyhani and HA Toliyat, "Flying-capacitor boost converter," 2012 Twenty-Seventh Annual IEEE Applied Power Electronics Conference and Exposition (APEC), Orlando, FL, USA, 2012, pp. 2311-2318, doi: 10.1109 / APEC.2012.6166145Pit-Leong Wong, Peng Xu, P. Yang and FC Lee, "Performance improvements of interleaving VRMs with coupling inductors," in IEEE Transactions on Power Electronics, vol. 16, no. 4, pp. 499-507, July 2001, doi: 10.1109 / 63.931059.

[0007] Boost converters in powertrains must be able to achieve variable boost and high-efficiency operation over continuous light to medium loads to improve fuel economy and power consumption, as well as low-loss operation under short-term heavy loads to ensure acceleration during full acceleration. Achieving this with a conventional boost converter requires switching losses, making it difficult to miniaturize the converter by increasing its frequency. This makes it impossible to increase the switching frequency under heavy loads due to cooling requirements, and the need for large passive components increases the system size.

[0008] Furthermore, when using a resonant switched capacitor system, switching losses can be reduced by soft switching using LC resonance, and the system can be made smaller by increasing the frequency. However, the boost ratio is generally fixed to an integer such as 2x or 3x, which means that the benefits of optimizing the boost ratio for improving fuel economy and power efficiency cannot be achieved.

[0009] One aspect of the present invention is a power converter comprising a leg and an inductor connected to the leg, wherein the leg is capable of generating multiple levels of voltage with four or more switching elements connected in series and one or more resonant capacitors connected thereto, the negative and positive ends of the leg are connected to high-voltage input / output terminals, the high-frequency voltage terminal of the leg is connected to one end of the inductor, and the other end of the inductor is connected to a low-voltage input / output terminal, and the converter is capable of two or more types of operation with different characteristics: a first mode in which soft switching operation is performed by resonating the inductor and the resonant capacitor, and a second mode in which chopper operation is performed without resonating the inductor and the resonant capacitor.

[0010] Another aspect of the present invention is a power conversion system in which a plurality of the above-described power converters are connected in parallel and operated out of phase with each other.

[0011] According to the present invention, it is possible to provide a power converter that enables both a fixed boost ratio and high-efficiency operation using a resonant state, and a variable boost ratio operation using a non-resonant state.

[0012] 1 is a diagram showing a basic configuration of a power converter in an embodiment of the present invention. FIG. 2 is a diagram showing an equivalent circuit of the power converter in in-phase mode. FIG. 3 is a diagram explaining the operation of the power converter in in-phase mode. FIG. 4 is a diagram explaining the operation of the power converter in non-common-phase mode. FIG. 5 is a diagram showing a switching pattern of the power converter in non-common-phase mode. FIG. 6 is a diagram showing a switching pattern of the power converter in non-common-phase mode. FIG. 7 is a diagram showing the transition over time of signals of each leg in non-common-phase mode when there is no magnetic coupling. FIG. 8 is a diagram showing the transition over time of signals of each leg in non-common-phase mode when there is magnetic coupling. FIG. 9 is a diagram showing the transition over time of signals of each leg in in-phase mode when there is no magnetic coupling. FIG. 10 is a diagram showing the transition over time of signals of each leg in in-phase mode when there is magnetic coupling. FIG. 11 is a diagram showing the trial calculation results of the efficiency of the power converter in an embodiment of the present invention. FIG. 12 is a diagram showing a modified example of the power converter in an embodiment of the present invention. FIG. 13 is a diagram showing a modified example of the power converter in an embodiment of the present invention. FIG. 14 is a diagram showing the configuration of a power conversion system in an embodiment of the present invention. FIG. 15 is a diagram showing a modified example of the power converter in an embodiment of the present invention.

[0013] As shown in FIG. 1, the power converter 100 according to the embodiment of the present invention includes an input capacitor Cin, a first resonant inductor L Ａ , second resonant inductor L Ｂ , leg A, first resonant capacitor C ｒｅｓｏＡ , leg B, second resonant capacitor C ｒｅｓｏＢ , and an output capacitor Cout.

[0014] The power converter 100 has a DC power supply Vin connected to its low-voltage input / output terminals (T1, T2) and a load 200 connected to its high-voltage input / output terminals (T3, T4). The power converter 100 is a power converter for efficiently supplying power from the DC power supply Vin to the load 200. It mainly uses a resonant inductor L Ａ , resonant inductor L Ｂ and resonance capacitor C ｒｅｓｏＡ , resonant capacitor C ｒｅｓｏＢ This is used to achieve highly efficient energy transfer.

[0015] In the input stage of the power converter 100, the output of a DC power supply Vin is connected to an input capacitor Cin, which suppresses ripples in the DC voltage from the power supply Vin, stabilizes it, and supplies it to the next stage.

[0016] The output from the positive terminal of the input capacitor Cin is connected to the first resonant inductor L Ａ and the second resonant inductor L Ｂ That is, the first resonant inductor L Ａ and the second resonant inductor L Ｂ One end of the first resonant inductor L is connected to the low-voltage side input / output terminal T1. Ａ The output terminal of the first resonant inductor L is connected to leg A. Ａ The other end of the second resonant inductor L is connected to the high frequency power supply terminal TA of leg A. Ｂ The output terminal of the second resonant inductor L is connected to leg B. Ｂ The other end of the first resonant inductor L is connected to the high frequency power supply terminal TB of the leg B. Ａ and the second resonant inductor L Ｂ stores and transfers energy as part of a resonant circuit.

[0017] Leg A is connected to the first switching element S １Ａ , second switching element S ２Ａ , third switching element S ３Ａ , the fourth switching element S ４Ａ and a resonant capacitor C ｒｅｓｏＡ The first switching element S １Ａ One end of the first switching element S is connected to the output capacitor Cout and the input / output terminal T3. １Ａ The other end of the first resonant capacitor C ｒｅｓｏＡ and one end of the second switching element S ２Ａ The second switching element S ２Ａ The other end of the first resonant inductor L Ａ the output terminal of the third switching element S ３Ａ The third switching element S ３Ａ The other end of the first resonant capacitor C ｒｅｓｏＡ and the other end of the fourth switching element S４Ａ The fourth switching element S ４Ａ The other end of leg A is connected to the input / output terminal T4 and the negative electrode of the DC power supply Vin (the negative electrode of the input capacitor Cin). Leg A controls the energy transfer of the resonant circuit by a switching operation.

[0018] Leg B is connected to the fifth switching element S １Ｂ , the sixth switching element S ２Ｂ , the seventh switching element S ３Ｂ , the eighth switching element S ４Ｂ and a resonant capacitor C ｒｅｓｏＢ The fifth switching element S １Ｂ One end of the fifth switching element S is connected to the output capacitor Cout and the input / output terminal T3. １Ｂ The other end of the second resonant capacitor C ｒｅｓｏＢ and one end of the sixth switching element S ２Ｂ The sixth switching element S ２Ｂ The other end of the second resonant inductor L Ｂ the output terminal of the seventh switching element S ３Ｂ The seventh switching element S ３Ｂ The other end of the second resonant capacitor C ｒｅｓｏＢ and the other end of the eighth switching element S ４Ｂ The eighth switching element S ４Ｂ The other end of leg B is connected to the input / output terminal T4 and the negative electrode of the DC power supply Vin (the negative electrode of the input capacitor Cin). Leg B controls the energy transfer of the resonant circuit by switching operation.

[0019] In the output stage of the power converter 100, the positive electrode of the output capacitor Cout is connected to the input / output terminal T3, and the negative electrode is connected to the input / output terminal T4. The output capacitor Cout smoothes the supplied electrical energy and supplies it to the load 200. The load 200 is connected between the input / output terminal T3 and the input / output terminal T4 of the power converter 100.

[0020] The power converter 100 has a circuit configuration in which a two-phase interleaved configuration with magnetic coupling is applied to a flying capacitor type three-level chopper. A power converter with a normal interleaved configuration operates to reduce current ripple by symmetrically shifting the phase of each phase, but the power converter 100 can operate in two different modes by switching the phase shift. Specifically, it uses a mode in which the voltages at the high-frequency voltage terminals of each leg are phase-shifted as in a normal interleaved configuration (out-of-phase mode), and a mode in which the phase shift is zero and the voltages at the high-frequency voltage terminals of all legs are in phase (in-phase mode).

[0021] 2 shows an equivalent circuit of the power converter 100 in common mode. The first resonant inductor L Ａ and the second resonant inductor L Ｂ Therefore, the inductance of the coupled inductor decreases depending on the magnetic coupling coefficient k, and the magnetic flux of the first resonant inductor L Ａ and the second resonant inductor L Ｂ are the common-mode inductors L ｃｏｍ In other words, the inductance can be changed equivalently by switching the mode.

[0022] First resonant capacitor C ｒｅｓｏＡ and a second resonant capacitor C ｒｅｓｏＢ and the common-mode inductance L ｃｏｍ By adjusting the resonant frequency to the switching frequency, in the common-phase mode, a switched capacitor operation using resonance with a small inductance can be realized. On the other hand, in the out-of-phase mode, the first resonant capacitor C ｒｅｓｏＡ and the second resonant capacitor C ｒｅｓｏＢ By utilizing the magnetic coupling between the two modes, it is possible to utilize two modes having different characteristics from the common mode, thereby realizing a power converter 100 that is smaller, more efficient, and less expensive than conventional power converters.

[0023] For example, in applications such as powertrains, where there are requirements for high output, light load, and variable boost, the high output requirement is met by using a switched capacitor, while only the variable boost operation is performed using chopper operation. This relaxes the output requirements for chopper operation, enabling the realization of a compact power converter 100 that can use higher frequencies.

[0024] Another application is the standardization of boost converters across different systems. For example, a boost converter requiring high output and fixed boost can be used in one system, while a boost converter requiring variable boost can be used in another. By using a switched capacitor for the former and a chopper for the latter, the power converter 100 can be applied as a common boost converter to different systems. This reduces development costs and achieves cost reductions through economies of scale.

[0025] The following describes the details of operation in common-mode and non-common-mode. Figure 3 shows a general shape of the operating waveform in common-mode. In common-mode, the switching of the two legs A and B occurs at the same timing. Therefore, only leg A is shown as a representative in Figure 3. Note that the upper first switching element S is shown for simplicity. １Ａ , second switching element S ２Ａ The lower third switching element S is a diode. ３Ａ , the fourth switching element S ４Ａ This is represented by an ideal switch. In reality, the upper first switching element S １Ａ , second switching element S ２Ａ By actively switching the second switching element S as a switching element, it is possible to operate with reduced conduction loss. The operating waveform is expressed in four operating modes, including a mode in which current is blocked by a diode. In MODE 1, the second switching element S ２Ａ and the fourth switching element S ４Ａ is conductive, and the first resonant capacitor C ｒｅｓｏＡ and the common-mode inductance L ｃｏｍ The DC power supply Vin is connected in series with the inductance L, causing LC resonance. ｃｏｍ Current i flowing through ＬＡThe current of the second switching element S becomes a sine wave, starting from 0 A and returning to 0 A after half a cycle. ２Ａ The current is blocked by the diode action of , and MODE 1 ends. During MODE 1, the current i ＬＡ The first resonant capacitor C ｒｅｓｏＡ In MODE 3, the first switching element S １Ａ and the third switching element S ３Ａ is conducted, and the DC power supply Vin and the common-mode inductance L ｃｏｍ , first resonant capacitor C ｒｅｓｏＡ , and the load 200 are connected in series. At this time, LC resonance occurs as in MODE 1, and the first resonant capacitor C ｒｅｓｏＡ The energy stored in the device is discharged as the output voltage Vout of the load 200. As a result, the power converter 100 transmits power from the low-voltage side to the high-voltage side.

[0026] In a steady state, the output voltage Vout is twice the DC power supply Vin, and a boost operation is performed at a fixed boost ratio. Each switch is a zero-current switch (ZCS) that switches with 0A of current at all times, which suppresses switching losses.

[0027] Next, the operation of the non-in-phase mode will be explained. Figure 4 shows a list of switching modes for leg A and leg B. Leg A and leg B are switched in non-in-phase, and in a power converter 100 with two legs, leg A and leg B are switched in opposite phases.

[0028] The voltage generated by leg A, v Ｌ In MODE (I) and MODE (IV), the first resonant capacitor C ｒｅｓｏＡ Terminal voltage V ＣＡ In MODE (II) and MODE (III), the output voltage Vout and 0 V and the first resonant capacitor C ｒｅｓｏＡ Terminal voltage V ＣＡ The terminal voltage V ＣＡ If the average output voltage is Vout / 2, the generated voltage v ＬIt can take on three voltage levels: 0, output voltage Vout / 2, and output voltage Vout.

[0029] DC power supply Vin and generated voltage v Ｌ The difference is the common-mode inductance L ｃｏｍ When applied to the inductor, current ripple is generated. To reduce the current ripple, the generated voltage v Ｌ Of the three available voltage levels, it is desirable to use the two levels closest to the DC power supply Vin. In addition, the first resonant capacitor C ｒｅｓｏＡ To balance the capacitor voltage, the switching pattern shown in Figure 5 is obtained by using MODE (II), where the capacitor voltage is discharged, and MODE (III), where it is charged, in a symmetrical manner. Note that this is shown for the case where the DC power supply Vin > output voltage Vout / 2.

[0030] First switching element S １Ａ and the fourth switching element S ４Ａ The two elements are switched exclusively, and the second switching element S ２Ａ and the third switching element S ３Ａ The first switching element S is switched exclusively. １Ａ and the second switching element S ２Ａ The same pulse width and phase difference are switched at 180 degrees. MODE(II) and MODE(III) occur alternately for the same amount of time, with MODE(I) in between.

[0031] Figure 6 shows the switching pattern when DC power supply Vin < output voltage Vout / 2. The waveform is almost the same as when DC power supply Vin > output voltage Vout / 2, except that MODE (IV) is applied instead of MODE (I).

[0032] In both the case of DC power supply Vin>output voltage Vout / 2 in Fig. 5 and the case of DC power supply Vin<output voltage Vout / 2 in Fig. 6, a gate signal can be generated by comparing two carrier waves that are 180 degrees out of phase with the duty ratio D. In addition, the step-up ratio characteristic is output voltage Vout=DC power supply Vin / duty ratio D.

[0033] 7 shows the results of a simulation of the time transition of various signals in leg A and leg B in the out-of-phase mode when there is no magnetic coupling between the resonant inductors L. FIG. 8 shows the results of a simulation of the time transition of various signals in leg A and leg B in the out-of-phase mode when there is magnetic coupling between the resonant inductors L. In FIGS. 7 and 8, the generated voltage v ＬＡ and the generated voltage v ＬＢ , resonant capacitor C ｒｅｓｏＡ and a resonant capacitor C ｒｅｓｏＢ The resonant capacitor voltage V ＣＡ and the resonant capacitor voltage v ＣＢ , the generated current i of the resonant inductor L ＬＡ and the generated current i ＬＢ Here, the DC power supply Vin is set to 125 V, the duty ratio D is set to 0.4, and the magnetic coupling coefficient k when magnetic coupling is present is set to 0.57.

[0034] Generated voltage V ＬＡ and the generated voltage v ＬＢ are waveforms with a phase shift of 180 degrees, and the current i generated by the difference between these and the DC power supply Vin and the magnetic coupling between the phases ＬＡ and the generated current i ＬＢ The resonant capacitor voltage v ＣＡ and the resonant capacitor voltage v ＣＢ is the generated voltage v ＬＡ and the generated voltage v ＬＢ The waveform is repeated charging and discharging at half the frequency of the output voltage Vout, and the average value is the output voltage Vout / 2.

[0035] 7 and 8, it can be seen that the current ripple is smaller when there is magnetic coupling than when there is no magnetic coupling. The reason why the current ripple is larger when there is no magnetic coupling than when there is magnetic coupling is because the self-inductance of the boost inductor is set small in order to perform resonant operation in the common mode.

[0036] Even when magnetic coupling is present, the common-mode inductance L ｃｏｍAlthough the value of is small, the current ripple is low because the current change is suppressed by the interleaving operation and magnetic coupling.

[0037] 9 shows the results of a simulation of the time transition of various signals in leg A and leg B in the common mode when there is no magnetic coupling between the resonant inductors L. FIG. 10 shows the results of a simulation of the time transition of various signals in leg A and leg B in the common mode when there is magnetic coupling between the resonant inductors L. In FIGS. 9 and 10, the generated voltage v ＬＡ and the generated voltage v ＬＢ , resonant capacitor C ｒｅｓｏＡ and a resonant capacitor C ｒｅｓｏＢ The resonant capacitor voltage V ＣＡ and the resonant capacitor voltage v ＣＢ , the generated current i of the resonant inductor L ＬＡ and the generated current i ＬＢ Here, the DC power supply Vin is set to 125 V, the duty ratio D is set to 0.4, and the magnetic coupling coefficient k when magnetic coupling is present is set to 0.57.

[0038] In the common-mode, the operation is the same regardless of whether magnetic coupling is present or not, and both realize resonant switched capacitor operation that realizes soft switching.

[0039] These waveforms show that when using a conventional power converter like power converter 100, which does not use magnetically coupled inductors or switch between common-phase and non-common-phase modes, achieving both resonant switched-capacitor operation and chopper operation in a single device configuration results in large current ripple in chopper operation. By using the configuration of power converter 100 and switching between common-phase and non-common-phase modes, the current ripple in chopper operation can be reduced. Therefore, dual-mode operation can be achieved in a single device configuration while maintaining high efficiency in both modes.

[0040] 11 shows the results of trial calculations of the efficiency of a boost converter using power converter 100. The figures show the results of trial calculations of the efficiency when an operating point with a boost ratio of 2 is switched capacitor operation using common mode, and when operating points with other boost ratios are chopper operation using non-common mode.

[0041] In the high load region of an input current of 1000 A, the efficiency in non-common-phase mode is around 96.1 to 97.6%, while in common-phase mode, a high efficiency of 98.4% is achieved due to the soft switching effect of the switched capacitor. In light to medium load conditions of an input current of 500 A or less, conduction loss is small, so a high efficiency of at least 97.8% or more can be achieved even in non-common-phase mode.

[0042] If chopper operation alone were to cover an operating range up to 1000 A and ensure an efficiency of 98.4%, equivalent to that of a switched capacitor, the design would require a significantly lower switching frequency to reduce switching loss, which would result in an increased size of the power converter 100. In the power converter 100 of this embodiment, by covering the high load range with switched capacitor operation and reducing switching loss, the operating range covered by chopper operation can be limited to light to medium loads, and a higher frequency can be used to reduce the size of the power converter 100.

[0043] In the above description, the power converter 100 is explained with two legs, but the number of legs can be further increased. Fig. 12 shows an example of the configuration of the power converter 100 provided with N legs that operate at three levels. In this power converter 100 as well, resonant switched capacitor operation can be realized by aligning the phases of all legs, and three-level chopper operation with reduced ripple can be realized by shifting the phase between each leg. In a configuration with N legs, the switching elements S corresponding to each other in each leg are mn By switching (m=1 to 4, n=A to N) with a phase difference of 360 deg / N, chopper operation in a non-in-phase mode can be realized.

[0044] Figure 13 shows an example configuration that expands the output voltage levels in each leg to four or more. In each leg, the output voltage level can be increased by increasing the number of switching elements connected in series and increasing the number of resonant capacitors connected in parallel. Increasing the number of output voltage levels reduces current ripple in chopper operation. Furthermore, the boost ratio of the switched-capacitor operation can be improved to an integer multiple of two or more.

[0045] Figure 14 shows a power conversion system 300 in which multiple power converters 100 are connected in parallel. That is, the power conversion system 300 is a multiphase configuration in which multiple power converters 100 are connected in parallel, with each power converter 100, which consists of an inductor and multiple legs, considered as one phase. By applying the power conversion system 300, the input and output capacitors can be miniaturized.

[0046] Furthermore, the power converter 100 is configured to switch between a common-phase mode in which the switching elements are controlled to make the voltages of each high-frequency voltage terminal in multiple legs in phase, and a non-common-phase mode in which the switching elements are controlled to phase-shift the voltages of each high-frequency voltage terminal in each leg. However, it is sufficient to have a configuration that switches between a first mode in which a soft switching operation is performed by resonating the inductor and the resonant capacitor, and a second mode in which a chopper operation is performed without resonating the inductor and the resonant capacitor.

[0047] For example, Fig. 15 shows a configuration of a power converter 102 including a combination of leg A and inductor L. The power converter 102 switches between the first mode and the second mode by switching the operating frequency. In the resonant switched capacitor operation, the inductor L and the capacitor C ｒｅｓｏＡ By switching the switching element of leg A near the resonant frequency, LC resonance is generated, achieving soft switching operation. Furthermore, during chopper operation, the switching pattern shown in Figures 5 and 6 is used, and switching is performed at a frequency higher than the resonant frequency. This makes it possible to achieve low-ripple chopper operation with suppressed resonance.

[0048] As shown in FIG. 16, an inductor L and a capacitor C ｒｅｓｏＡ The power converter 104 may have a circuit configuration similar to that of the power converter 102 shown in FIG. 15, but the inductor L and the capacitor C ｒｅｓｏＡ During resonant operation, the inductance of the inductor L and the capacitor C are adjusted using variable elements to match the switching frequency and the resonant frequency. ｒｅｓｏＡ In addition, during chopper operation, the inductance of inductor L and the capacitance of capacitor C are reduced to suppress resonance by lowering the resonance frequency below the switching frequency. ｒｅｓｏＡ Increase the capacitance of

[0049] In addition, the inductor L and the capacitor C ｒｅｓｏＡ Alternatively, only one of the power converters 102 and 104 may be configured as a variable element. Alternatively, a hybrid configuration may be formed by combining the power converter 102 and the power converter 104 and also using a variable switching frequency. Also, in the configuration of the power converters 102 and 104, a leg configuration with three or more voltage levels may be applied, as shown in FIG. 13. Alternatively, a power conversion system may be formed in which a plurality of power converters 102 and 104 are connected in parallel, as shown in FIG. 14.

[0050] Furthermore, in this embodiment, the explanation is based on the assumption of voltage boost operation, but since both the resonant switched capacitor and chopper circuit are circuits that can transmit power in both directions, voltage drop operation in which power is transmitted from the high voltage side to the low voltage side is also possible.

[0051] 100, 102, 104 power converter, 200 load, 300 power conversion system.

Claims

1. A power converter comprising a leg and an inductor connected to the leg, wherein the leg is capable of generating multiple levels of voltage with four or more switching elements connected in series and one or more resonant capacitors connected thereto, the negative and positive ends of the leg are connected to high-voltage input / output terminals, the high-frequency voltage terminal of the leg is connected to one end of the inductor, and the other end of the inductor is connected to a low-voltage input / output terminal, and the converter is capable of two or more types of operation with different characteristics: a first mode in which soft switching operation is performed by resonating the inductor and the resonant capacitor, and a second mode in which chopper operation is performed without resonating the inductor and the resonant capacitor.

2. A power converter as claimed in claim 1, comprising a plurality of combinations of the legs and the inductors, the inductors being magnetically coupled to each other, the negative and positive ends of each of the legs being connected to a high-voltage side input / output terminal, the high-frequency voltage terminal of each of the legs being connected to one end of the inductor, and the other end of the inductor being connected to a low-voltage side input / output terminal, and two or more types of operation with different characteristics being possible by switching the switching phase of the legs.

3. A power converter as claimed in claim 2, wherein the two or more types of operation are performed by switching between an in-phase mode in which the switching elements are controlled to switch so that the voltages at the high frequency voltage terminals of the legs are in phase, and an out-of-phase mode in which the switching elements are controlled to switch so that the voltages at the high frequency voltage terminals of the legs are phase shifted.

4. A power converter according to claim 3, wherein in the common mode, soft switching operation is performed utilizing resonance between the inductor and the resonant capacitor.

5. A power converter according to claim 4, wherein in the common mode, soft switching operation is performed at a boost ratio of an integer multiple of 2 or more.

6. A power converter according to claim 3, wherein in the out-of-phase mode, a chopper operation is performed by magnetic coupling and phase shift of the inductors.

7. A power converter according to claim 4, wherein in the out-of-phase mode, a chopper operation is performed by magnetic coupling and phase shift of the inductors.

8. A power converter according to claim 5, wherein in the out-of-phase mode, a chopper operation is performed by magnetic coupling and phase shift of the inductors.

9. A power converter according to claim 6, wherein in the non-common-phase mode, a variable boost chopper operation with three or more levels is performed.

10. A power converter according to claim 7, wherein in the non-common-phase mode, a variable boost chopper operation with three or more levels is performed.

11. A power converter according to claim 8, wherein in the non-common-phase mode, a variable boost chopper operation with three or more levels is performed.

12. A power conversion system in which a plurality of power converters according to claim 1 are connected in parallel and operated out of phase with each other.

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