Ac electric motor control device and ac electric motor control method

Abstract

Disclosed are an AC electric motor control device and an AC electric motor control method that make it possible to attain a highly accurate and stable control response without complicating a control system. The AC electric motor control device controls a power converter for driving an AC electric motor, and comprises: a second current command computation unit (201, 202) that generates a second current command value (Id **, Iq **) such that a first current command value (Id *, Iq *) and a current detection value (Idc, Iqc) of the AC electric motor match each other in a rotating coordinate system; a magnetic flux command computation unit (203, 204) that calculates, for the AC electric motor, a magnetic flux command value (Φd **, Φq **) in the rotating coordinate system; and a voltage command computation unit (205) that, on the basis of the second current command value and the magnetic flux command value, generates, for the power converter, a voltage command value (Vd **, Vq **) in the rotating coordinate system.

Description

Control device for AC motor, and method for controlling AC motor

[0001] The present invention relates to a control device for an AC motor that controls a power converter that drives an AC motor, and to a method for controlling an AC motor.

[0002] AC motors exhibit nonlinearity due to magnetic saturation, but are typically controlled based on a linear model in which magnetic flux increases or decreases in proportion to the current.

[0003] Permanent magnet synchronous motors (hereinafter referred to as "PM motors"), which are widely used as AC motors, are being made smaller and lighter by reducing the amount of magnetic material in order to increase efficiency and power density. As a result, magnetic saturation is more likely to occur, and the nonlinearity of PM motors is becoming more pronounced.

[0004] Under these circumstances, control technologies that take into account the nonlinearity of AC motors are required.

[0005] As a conventional control technique that takes nonlinearity into consideration, the technique described in Patent Document 1 is known.

[0006] In the technology described in Patent Document 1, magnetic flux and magnetic flux deviation are calculated using nonlinear functions of magnetic flux and current that represent magnetic saturation and interaxial interference. A voltage command value is generated from these calculated values.

[0007] Japanese Patent Publication No. 2010-239730

[0008] In the conventional technology described above, in order to obtain a highly accurate and stable control response in the high-speed range of an AC motor, the control system becomes complex, the number of parameters used in magnetic flux calculations increases, and it becomes difficult to identify the parameters.

[0009] Therefore, the present invention provides a control device for an AC motor and a control method for an AC motor that can obtain a highly accurate and stable control response without complicating the control system.

[0010] To solve the above problems, the AC motor control device according to the present invention controls a power converter that drives an AC motor and comprises: a second current command calculation unit that generates a second current command value in a rotating coordinate system such that the first current command value and the current detection value of the AC motor match; a magnetic flux command calculation unit that calculates the magnetic flux command value of the AC motor in a rotating coordinate system; and a voltage command calculation unit that generates a voltage command value for the power converter in a rotating coordinate system based on the second current command value and the magnetic flux command value.

[0011] To solve the above problems, the AC motor control method according to the present invention is a control method in which a control device controls a power converter that drives an AC motor, and generates a second current command value in a rotating coordinate system such that the first current command value and the detected current value of the AC motor match, calculates a magnetic flux command value of the AC motor in the rotating coordinate system, and generates a voltage command value for the power converter in the rotating coordinate system based on the second current command value and the magnetic flux command value.

[0012] According to the present invention, a highly accurate and stable control response can be obtained without complicating the control system.

[0013] Other issues, configurations, and effects not mentioned above will be clarified by the following description of the embodiments.

[0014] This is a system configuration diagram showing the AC motor drive system of Embodiment 1. This is a diagram showing the coordinate axes of the coordinate system used in the control of the PM motor 103 in Embodiment 1. This is a functional block diagram showing the configuration of the vector control unit 112 (Figure 1). This is a functional block diagram showing the configuration of the second d-axis current command calculation unit 201 (Figure 3). This is a functional block diagram showing the configuration of the second q-axis current command calculation unit 202 (Figure 3). This is a functional block diagram showing the configuration of the voltage vector calculation unit 205 (Figure 3). This is a current-magnetic flux characteristic diagram showing an example of the relationship between magnetic flux and current in the PM motor 103. This is a system configuration diagram showing the AC motor drive system of Embodiment 2. This is a functional block diagram showing the configuration of the vector control unit 112b (Figure 8). This is a functional block diagram showing the configuration of the voltage vector calculation unit 205b (Figure 9).

[0015] Embodiments of the present invention will be described below with reference to Examples 1 and 2, using the drawings. In each figure, elements with the same reference number represent the same or similar functional elements.

[0016] Figure 1 is a system configuration diagram showing an AC motor drive system, which is an embodiment 1 of the present invention.

[0017] The AC motor drive system of Embodiment 1 includes a PM motor 103 as the AC motor to be driven, a power converter 102 that drives the PM motor 103, a control device 101 that controls the power converter 102, and a torque command value T for the PM motor 103. ｍ ＊ The system includes a command generator 105 that generates a command, a phase current detector 121 that detects the current flowing through the PM motor 103, and a rotor position detector 124 that detects the rotor position of the PM motor 103.

[0018] The PM motor 103 is driven by three-phase AC power output from the power converter 102.

[0019] The power converter 102 includes input terminals 123a and 123b for supplying DC power, a main circuit section 132 composed of six switching elements Sup to Swn, a gate driver 133 for driving the main circuit section 132, a DC resistor 134 for overcurrent protection, and a smoothing capacitor 131. Based on the gate command signal generated by the control device 101, the power converter 102 converts the DC power supplied from the input terminals 123a and 123b into AC power and supplies the AC power to the PM motor 103.

[0020] The phase current detector 121 detects the alternating current of two phases (U phase and W phase) of the three-phase alternating current flowing from the power converter 102 to the PM motor 103 (i ｕ , i ｗ ). As the phase current detector 121, for example, a current sensor using a Hall element is applied.

[0021] The phase current detector 121 may also detect three-phase (U-phase, V-phase, and W-phase) alternating current. Alternatively, instead of the phase current detector 121, a DC resistor 134 may be used to detect three-phase alternating current in a so-called single-shunt method.

[0022] The rotor position detector 124 outputs an angular signal φ corresponding to the rotor position (rotation angle) of the PM motor 103. As the rotor position detector 124, for example, a resolver, a rotary encoder, a magnetic sensor, or the like is applied.

[0023] Instead of the rotor position detector 124, a speed detector (for example, a rotary encoder) may be used to detect the rotation speed, and the rotor position may be calculated based on the detected value of the rotation speed. Further, the rotation speed and rotor position of the PM motor 103 may be estimated sensorless based on a voltage command value, a current detection value, or the like.

[0024] The command generator 105 is a higher-level control device that generates a torque command value T for the PM motor 103. ｍ ＊ which is.

[0025] In this embodiment, the torque generated by the PM motor 103 is controlled, but the present invention is not limited to this, and any one of the current, rotation speed, and position of the PM motor 103 may be controlled. In this case, the command generator 105 generates a current command value, a speed command value, and a position command value, respectively.

[0026] The control device 101 is based on the angular signal φ corresponding to the rotor position (rotation angle) of the PM motor 103 from the rotor position detector 124, the detected value I of the alternating current flowing through the PM motor 103, ｕ , I ｗ and the torque command value T from the command generator 105, ｍ ＊ and generates a control command signal for driving the switching elements Sup to Swn (IGBT in FIG. 1) constituting the main circuit section 132 by a current control system and a phase control system, and outputs the control command signal to the gate driver 133.

[0027] The gate driver 133 performs on / off control of the switching elements Sup to Swn according to the control command signal output by the control device 101.

[0028] In the control device 101 in this embodiment, so-called vector control is applied. Therefore, the coordinate system used for the control in this embodiment will be described.

[0029] Figure 2 shows the coordinate axes of the coordinate system used in the control of the PM motor 103 in Example 1. Note that all of these coordinate systems are known coordinate systems.

[0030] The ab-axis coordinate system, defined by mutually orthogonal a-axis and b-axis, is a stator coordinate system representing the phase of the stator windings of the PM motor 103. The a-axis is based on the phase of the u-phase winding of the PM motor 103.

[0031] The dq-axis coordinate system, defined by mutually orthogonal d-axis and q-axis, is a rotating coordinate system representing the magnetic pole position of the rotor of the PM motor 103, and rotates in synchronization with the magnetic pole position of the PM motor 103's rotor. The d-axis is based on the north pole direction of the magnetic poles provided by the permanent magnets on the rotor of the PM motor 103. The d-axis is also called the magnetic pole axis.

[0032] The DC-QC axis coordinate system, defined by mutually orthogonal DC and QC axes, is a rotating coordinate system used by the control device 101 for vector control. The DC and QC axes are also called control axes.

[0033] The z-axis represents the phase of the angle signal φ, which corresponds to the rotor position (rotation angle) of the PM motor 103 output by the rotor position detector 124.

[0034] The phase information between the coordinate axes shown in Figure 2 will be described later as appropriate.

[0035] The configuration of the control device 101 in Example 1 will be described below.

[0036] As shown in Figure 1, the control device 101 includes a current command calculation unit 111, a vector control unit 112, a current detection unit 113, a dq coordinate transformation unit 114, a magnetic pole position transformation unit 115, a control phase / frequency calculation unit 116, a polar coordinate transformation unit 117, a UVW coordinate transformation unit 118, and a PWM controller 119.

[0037] The current command calculation unit 111 calculates the electrical angular frequency ω of the rotational speed of the PM motor 103, which is output from the control phase / frequency calculation unit 116. １ And the voltage detection value E of the smoothing capacitor 131 ｃｆ The torque command value T output from the command generator 105.ｍ ＊ The current command value I on the dc-qc axis coordinate system is calculated based on this. ｄ ＊ , I ｑ ＊ Outputs.

[0038] Furthermore, the current command calculation unit 111 is T ｍ ＊ , E ｃｆ and ω １ Then, using means (reference table, function formula, approximation formula, design formula, theoretical formula, etc.) that express the relationship with the optimal dq axis current command value, the current command value I ｄ ＊ , I ｑ ＊ This is calculated. Such methods are obtained beforehand through testing and analysis.

[0039] The vector control unit 112 receives the current detection value I on the dc-qc axis coordinate system output by the dq coordinate transformation unit 114. ｄｃ , I ｑｃ The current command value I on the dc-qc axis coordinate system output by the current command calculation unit 111. ｄ ＊ , I ｑ ＊ To match the torque current component (q-axis current component) and the excitation current component (d-axis current component), current control is performed on each component separately. The vector control unit 112 uses the result of the current control and the electrical angular frequency ω of the rotational speed of the PM motor 103. １ Based on this, the voltage command value V on the dc-qc axis coordinate system, which is a rotating coordinate system. ｄ ＊＊ , V ｑ ＊＊ Perform the calculation and output the result.

[0040] The current detection unit 113 detects the alternating current i flowing through the PM motor 103 detected by the phase current detector 121. ｕ , i ｗ From the three-phase current detection value I ｕ , I ｖ , I ｗ The calculation is performed and output to the dq coordinate transformation unit 114.

[0041] The dq coordinate transformation unit 114 processes the three-phase current detection value I output by the current detection unit 113.ｕ , I ｖ , I ｗ The control phase θ output from the control phase / frequency calculation unit 116 is then used to determine the control phase θ. ｄｃ Based on this, the current detection value I on the dc-qc axis coordinate system ｄｃ , I ｑｃ Convert and output.

[0042] The magnetic pole position conversion unit 115 converts the magnetic pole position information θ of the PM motor 103 based on the angle signal φ corresponding to the rotor position of the PM motor 103 output from the rotor position detector 124. ｄ Output '.

[0043] The control phase / frequency calculation unit 116 receives magnetic pole position information θ output by the magnetic pole position conversion unit 115. ｄ Based on ', the electrical angular frequency ω of the rotational speed of the PM motor 103 １ This involves the dq coordinate transformation of the AC current detection value and the voltage command value V. ｄ ＊＊ , V ｑ ＊＊ The control phase θ of the dc-qc axis coordinate system used in the UVW coordinate transformation. ｄｃ (Figure 2) is calculated and output.

[0044] The polar coordinate transformation unit 117 receives the voltage command value V on the dc-qc axis output by the vector control unit 112. ｄ ＊＊ , V ｑ ＊＊ The voltage amplitude command value V １ ＊＊ It is then converted to a voltage phase command value δ and output.

[0045] The UVW coordinate transformation unit 118 receives the voltage amplitude command value V output by the polar coordinate transformation unit 117. １ ＊＊ The control phase θ output by the control phase / frequency calculation unit 116 is used to determine the voltage phase command value δ. ｄｃ Based on this, the three-phase AC voltage command value V ｕ ＊＊ , V ｖ ＊＊ , V ｗ ＊＊ It is converted to and output to the PWM controller 119.

[0046] The PWM controller 119 uses a triangular wave signal generated based on an arbitrary carrier frequency fc and the voltage detection value Ecf of the smoothing capacitor 131 as a carrier wave, and uses the three-phase AC voltage command values V ｕ ＊＊ , V ｖ ＊＊ , V ｗ ＊＊ output by the UVW coordinate conversion unit 118 as a modulation wave, generates a control command signal for driving the switching elements Sup to Swn by pulse width modulation, and outputs it to the gate driver 133.

[0047] FIG. 3 is a functional block diagram showing the configuration of the vector control unit 112 (FIG. 1).

[0048] As shown in FIG. 3, the vector control unit 112 includes a second d-axis current command calculation unit 201, a second q-axis current command calculation unit 202, a second d-axis magnetic flux command calculation unit 203, a second q-axis magnetic flux command calculation unit 204, and a voltage vector calculation unit 205.

[0049] The second d-axis current command calculation unit 201 is based on the first d-axis current command value I ｄ ＊ on the dc axis and the d-axis current detection value I ｄｃ on the dc axis, and makes I ｄｃ coincide with I ｄ ＊ to calculate and output the second d-axis current command value I ｄ ＊＊ on the dc axis.

[0050] The second q-axis current command calculation unit 202 is based on the first q-axis current command value I ｑ ＊ on the qc axis and the q-axis current detection value I ｑｃ on the qc axis, and makes I ｑｃ coincide with I ｑ ＊ to calculate and output the second q-axis current command value I ｑ ＊＊ on the qc axis.

[0051] The second d-axis magnetic flux command calculation unit 203 is based on I ｄ ＊＊ output by the second d-axis current command calculation unit 201 and I output by the second q-axis current command calculation unit 202ｑ ＊＊ Based on this, the second d-axis flux command value Φ on the dc axis ｄ ＊＊ is calculated and output.

[0052] The second q-axis flux command calculation unit 204 uses the I output by the second d-axis current command calculation unit 201 ｄ ＊＊ and the I output by the second q-axis current command calculation unit 202 ｑ ＊＊ Based on this, the second q-axis flux command value Φ on the qc axis ｑ ＊＊ is calculated and output.

[0053] The voltage vector calculation unit 205 uses the I output by the second d-axis current command calculation unit 201 ｄ ＊＊ and the I output by the second q-axis current command calculation unit 202 ｑ ＊＊ and the Φ output by the second d-axis flux command calculation unit 203 ｄ ＊＊ and the Φ output by the second q-axis flux command calculation unit 204 ｑ ＊＊ and the ω output by the control phase / frequency calculation unit 116 (Fig. 1) １ Based on this, the d-axis voltage command value V on the dc axis ｄ ＊＊ and the q-axis voltage command value V on the qc axis ｑ ＊＊ are calculated and output.

[0054] Fig. 4 is a functional block diagram showing the configuration of the second d-axis current command calculation unit 201 (Fig. 3).

[0055] As shown in Fig. 4, the second d-axis current command calculation unit 201 includes a proportionality unit 301, a d-axis integrator 302, and an adder / subtractor 801.

[0056] The adder / subtractor 801 calculates the difference ΔI ｄ ＊ between the first d-axis current command value I ｄｃ and the d-axis current detection value I ｄｃ (= I ｄ ＊ - I ｄｃ ).

[0057] The proportionality unit 301 uses ΔIｄｃ Gain K ｉｄ Multiply by .

[0058] The d-axis integrator 302 calculates the second d-axis current command value I by integrating the multiplication value from the proportionalizer 301, as shown in equation (1). ｄ ＊＊ Calculate.

[0059]

[0060] Here, gain K ｉｄ This is set, for example, by equation (2).

[0061]

[0062] In equation (2), ω ａｃｒ ＊ This is the control response angular frequency for current control.

[0063] Figure 5 is a functional block diagram showing the configuration of the second q-axis current command calculation unit 202 (Figure 3).

[0064] As shown in Figure 5, the second q-axis current command calculation unit 202 includes a proportional control unit 303, a q-axis integrator 304, and an adder / subtractor 802.

[0065] The adder / subtractor 802 controls the first q-axis current command value I ｑ ＊ And the q-axis current detection value I ｑｃ The difference ΔI ｑｃ (=I ｑ ＊ -I ｑｃ Calculate the result.

[0066] The proportionality unit 303 is ΔI ｑｃ Gain K ｉｑ Multiply by .

[0067] The q-axis integrator 304 calculates the second q-axis current command value I by integrating the multiplication value from the proportionalizer 303, as shown in equation (3). ｑ ＊＊ Calculate.

[0068]

[0069] Here, gain K ｉｑ This is set, for example, by equation (4).

[0070]

[0071] In equation (4), ω ａｃｒ ＊ This is the control response angular frequency for current control.

[0072] Note that the proportional gain K ｉｄ , K ｉｑ This can be a constant, or it can be a function of the current command value, current detection value, speed, vector calculation period, etc.

[0073] As described above, in Example 1, as shown in formulas (1) and (3), I ｄ ＊＊ and I ｑ ＊＊ This is calculated by integral (I) control, but is not limited to this; it may also be calculated by proportional (P) control or proportional-integral (PI) control. ｄ ＊＊ and I ｑ ＊＊ This may be calculated using a first-order lag controller that constitutes a feedforward control system.

[0074] The second d-axis magnetic flux command calculation unit 203 is I ｄ ＊＊ and I ｑ ＊＊ And, Φ ｄ ＊＊ Refer to the table data that shows the correspondence with the entered I ｄ ＊＊ and I ｑ ＊＊ Based on, Φ ｄ ＊＊ This calculates the following. This table data is obtained in advance through testing or analysis. Alternatively, a function formula, approximation formula, design formula, etc., may be used instead of the table data.

[0075] The second q-axis magnetic flux command calculation unit 204 is I ｄ ＊＊ and I ｑ ＊＊ And, Φ ｑ ＊＊ Refer to the table data that shows the correspondence with the entered I ｄ ＊＊and I ｑ ＊＊ Based on, Φ ｑ ＊＊ This calculates the following. This table data is obtained in advance through testing or analysis. Alternatively, a function formula, approximation formula, design formula, etc., may be used instead of the table data.

[0076] Figure 6 is a functional block diagram showing the configuration of the voltage vector calculation unit 205 (Figure 3).

[0077] As shown in Figure 6, the voltage vector calculation unit 205 includes proportionals 401, 403, 405, 406, 407, a d-axis differentiator 402, a q-axis differentiator 404, adders 803, 804, 806, 807, an adder / subtractor 805, and multipliers 901, 902.

[0078] In the voltage vector calculation unit 205, V ｄ ＊＊ is, I ｄ ＊＊ R proportional to the gain １ ＊ Multiply by Φ ｄ ＊＊ Differentiate the result, I ｑ ＊＊ Gain L ｑ ＊ and ω １ It is calculated by multiplying by and then adding or subtracting these. That is, V ｄ ＊＊ This is calculated using a voltage equation like equation (5).

[0079]

[0080] Furthermore, in the voltage vector calculation unit 205, V ｑ ＊＊ is, I ｑ ＊＊ Gain R １ ＊ Multiply by Φ ｑ ＊＊ Differentiate the result, I ｄ ＊＊ Gain L ｄ ＊ and ω １ Multiply by ω １ Gain K ｅ ＊It is calculated by multiplying by and adding these together. That is, V ｑ ＊＊ This is calculated using a voltage equation like equation (6).

[0081]

[0082] In equations (5) and (6), R １ ＊ L is the control setting value for the primary resistance of the PM motor 103. ｄ ＊ L is the control setting value for the d-axis inductance of the PM motor 103. ｑ ＊ This is the control setting value for the q-axis inductance of the PM motor 103, K ｅ ＊ This is the control setting value for the induced voltage constant of the PM motor 103. Note that the gain R in the voltage vector calculation unit 205 is... １ ＊ , L ｄ ＊ , L ｑ ＊ , K ｅ ＊ This can be a constant, or it can be represented by a function or reference table such as the current command value, current detection value, or magnet temperature.

[0083] Next, we will explain how to improve the accuracy of the control response according to Example 1.

[0084] First, the voltage equation for a PM motor, considering the nonlinearity of the magnetic circuit, is expressed as shown in equation (7).

[0085]

[0086] In equation (7), L ｄ , L ｑ L represents the static inductance of the PM motor. ｄｈ , L ｑｈ This represents the dynamic inductance of the PM motor. These inductances will be explained using Figure 7.

[0087] Figure 7 is a current-magnetic flux characteristic diagram showing an example of the relationship between magnetic flux and current in the PM motor 103.

[0088] As shown in Figure 7, due to the nonlinear characteristics of the magnetic flux caused by the saturation phenomenon of the magnetic circuit, the q-axis current I ｑ The larger the q-axis magnetic flux Φ, the greater the q-axis magnetic flux Φ. ｑ The rate of increase will slow down.

[0089] In Figure 7, the static inductance L ｑ is the q-axis current I ｑ A point where is 0 and a certain operating point (I ｑ , Φ ｑ This is the slope of the straight line connecting the points. Dynamic inductance L ｑｈ is a certain operating point (I ｑ ,Φ ｑ This corresponds to the slope of the magnetic flux change with respect to the current change in the vicinity.

[0090] Furthermore, the relationship between d-axis magnetic flux and d-axis current, and static inductance L are also discussed. ｄ , dynamic inductance L ｄｈ The same applies to this matter.

[0091] In the calculation of the voltage command value for vector control of a PM motor based on equation (7), static inductance L ｄ , L ｑ And, dynamic inductance L ｄｈ , L ｑｈ Considering the current dependence, these inductances can be represented by table data or functional expressions with d-axis current and q-axis current as variables.

[0092] Here, static inductance corresponds to the slope of the change in magnetic flux from the point of zero current to a certain current operating point; therefore, it is uniquely determined once the operating point is determined. Furthermore, when attempting to identify static inductance from actual measurements, it is possible to identify static inductance from a steady-state test in which the motor speed and torque (d-axis current, q-axis current) are kept constant.

[0093] However, since dynamic inductance corresponds to the slope of the magnetic flux change with respect to the current change near a certain operating point, it is difficult to uniquely determine the value because it fluctuates depending on the definition of the magnitude of the current change even when the operating point is determined. Furthermore, when attempting to identify dynamic inductance from actual measurements, it is difficult to identify it through actual machine testing because it measures transient changes using a current step response test, and even if it can be identified, the value of the dynamic inductance fluctuates depending on the magnitude of the current step. As a result, constant errors occur in the dynamic inductance depending on how the motor is operated, causing variability in the transient response of the current control system.

[0094] Therefore, in Example 1, the voltage command value for vector control is calculated based on a voltage equation that takes into account the nonlinearity of the motor magnetic circuit, without using dynamic inductance. In Example 1, the voltage equation for the PM motor 103 is expressed as equation (8) below.

[0095]

[0096] As shown in equation (8), the dynamic inductance L in equation (7) ｄｈ , L ｑｈ The transient induced electromotive force (dynamic inductance × derivative of current) is generated by the d-axis magnetic flux Φ ｄ and q-axis magnetic flux Φ ｑ The transient induced electromotive force, i.e., Φ ｄ and Φ ｑ This can be replaced by the derivative of .

[0097] Here, the d-axis magnetic flux Φ ｄ and q-axis magnetic flux Φ ｑ When considering the nonlinearity of the motor magnetic circuit, the relationship between magnetic flux and current can be expressed using table data or a functional equation. This relationship between magnetic flux and current is a steady-state characteristic, similar to static inductance, and is uniquely determined for each current operating point (d-axis current, q-axis current).

[0098] Furthermore, when obtaining static inductance table data from steady-state tests by actual measurement, static inductance L ｄ , L ｑIf information such as the dq-axis voltage command value, motor rotation speed, dq-axis current detection value, and magnet temperature that formed the basis for the calculation is available, then the magnetic flux Φ can be calculated based on this information. ｄ , Φ ｑ The reference table can be derived.

[0099] Furthermore, in Example 1, the second current command value I ｄ ＊＊ , I ｑ ＊＊ and the second magnetic flux command value Φ ｄ ＊＊ , Φ ｑ ＊＊ Using this as the inverse model of the motor, the voltage command value V on the dc-qc axis coordinate system ｄ ＊＊ , V ｑ ＊＊ By calculating this, the current command value I on the dc-qc axis coordinate system can be calculated even in the high-speed range. ｄ ＊ , I ｑ ＊ Current detection value I ｄｃ , I ｑｃ It can be matched with high precision, meaning tracking performance is improved. Furthermore, the motor can be controlled stably even in high-speed ranges where the effects of dq-axis interference become significant.

[0100] As explained above, according to Example 1, the first current command value I in the rotating coordinate system (dc-qc axis coordinate system) ｄ ＊ , I ｑ ＊ And the current detection value I in the rotating coordinate system (dc-qc axis coordinate system) ｄｃ , I ｑｃ The second current command value I in the rotating coordinate system (dc-qc axis coordinate system) generated so that it matches the above. ｄ ＊＊ , I ｑ ＊＊ And the second current command value I ｄ ＊＊ , I ｑ ＊＊ The second magnetic flux command value Φ in the rotating coordinate system (dc-qc axis coordinate system) is generated based on this. ｄ ＊＊ ,Φ ｑ ＊＊Based on this, the voltage command value V in the rotating coordinate system (dc-qc axis coordinate system) ｄ ＊＊ ,V ｑ ＊＊ This is generated.

[0101] This enables highly accurate and stable control of the AC motor (PM motor 103) while taking into account the nonlinearity of the AC motor (PM motor 103). In addition, the second magnetic flux command value Φ is used instead of the dynamic inductance. ｄ ＊＊ ,Φ ｑ ＊＊ By using this method, control that takes nonlinearity into account becomes possible without complicating the control system. This makes it easier to identify parameters, reference tables, and function equations used in the control, and shortens the time required for actual machine fitting.

[0102] Furthermore, the AC motor control device according to Embodiment 1 is not limited to PM motors, but may also be applied to other AC motors such as wound-wound synchronous motors and reluctance synchronous motors. The AC motor control device according to Embodiment 1 can also be applied when the AC motor operates as a generator.

[0103] Below, we will mainly explain the differences between Example 2 and Example 1.

[0104] Figure 8 is a system configuration diagram showing an AC motor drive system, which is an embodiment 2 of the present invention.

[0105] In this second embodiment, as will be described later, the vector control unit 112b provided in the control device 101 has a different configuration from the vector control unit 112 in the first embodiment (Figure 1).

[0106] Furthermore, the vector control unit 112b, similar to the vector control unit 112 (Figure 1), controls the current command value I on the dc-qc axis coordinate system. ｄ ＊ , I ｑ ＊ And the current detection value I on the dc-qc axis coordinate system ｄｃ , I ｑｃ And the electrical angular frequency ω of the rotational speed of the PM motor 103 １ Enter the voltage command value V on the dc-qc axis coordinate system. ｄ＊＊ , V ｑ ＊＊ Perform the calculation and output the result.

[0107] Figure 9 is a functional block diagram showing the configuration of the vector control unit 112b (Figure 8).

[0108] As shown in Figure 9, the vector control unit 112b includes a second d-axis current command calculation unit 201 and a second q-axis current command calculation unit 202, similar to Embodiment 1 (Figure 3). Furthermore, unlike Embodiment 1 (Figure 3), the vector control unit 112b includes a first d-axis magnetic flux command calculation unit 206b, a d-axis magnetic flux estimation calculation unit 207b, a first q-axis magnetic flux command calculation unit 208b, a q-axis magnetic flux estimation calculation unit 209b, and a voltage vector calculation unit 205b.

[0109] The second d-axis current command calculation unit 201 calculates the d-axis current command value I on the dc axis, similar to the first embodiment (Figure 3). ｄ ＊ And the detected d-axis current value I on the dc axis ｄｃ Based on that, I ｄｃ to I ｄ ＊ The second d-axis current command value I on the dc axis is set to match this. ｄ ＊＊ Calculate and output the result.

[0110] The second q-axis current command calculation unit 202 calculates the q-axis current command value I on the qc axis, similar to the first embodiment (Figure 3). ｑ ＊ And the detected q-axis current value I on the qc axis ｑｃ Based on that, I ｑｃ to I ｑ ＊ The second q-axis current command value I on the qc axis is set to match this. ｑ ＊＊ Calculate and output the result.

[0111] The first d-axis magnetic flux command calculation unit 206b calculates the current command value I on the dc-qc axis coordinate system output by the current command calculation unit 111 (Figure 8). ｄ ＊ , I ｑ ＊ Based on this, the first d-axis magnetic flux command value Φ ｄ ＊ Perform the calculation and output the result.

[0112] The first d-axis magnetic flux command calculation unit 206b is I ｄ ＊ and I ｑ ＊ And, Φ ｄ ＊ Refer to the table data that shows the correspondence with the entered I ｄ ＊ and I ｑ ＊ Based on, Φ ｄ ＊ This calculates the following. This table data is obtained in advance through testing or analysis. Alternatively, a function formula, approximation formula, design formula, etc., may be used instead of the table data.

[0113] The d-axis magnetic flux estimation calculation unit 207b calculates the current detection value I on the dc-qc axis coordinate system output by the dq coordinate transformation unit 114 (Figure 8). ｄｃ , I ｑｃ Based on this, the estimated d-axis magnetic flux Φ ｄｃ Perform the calculation and output the result.

[0114] The d-axis magnetic flux estimation calculation unit 207b is I ｄｃ and I ｑｃ And, Φ ｄｃ Refer to the table data that shows the correspondence with the entered I ｄｃ and I ｑｃ Based on, Φ ｄｃ This calculates the following. This table data is obtained in advance through testing or analysis. Alternatively, a function formula, approximation formula, design formula, etc., may be used instead of the table data.

[0115] The first d-axis magnetic flux command calculation unit 206b and the d-axis magnetic flux estimation calculation unit 207b may use the same table data, or they may use different table data. However, using the same table data can reduce the amount of work required to create the table data.

[0116] The first q-axis magnetic flux command calculation unit 208b calculates the current command value I on the dc-qc axis coordinate system output by the current command calculation unit 111 (Figure 8). ｄ ＊ , I ｑ ＊ Based on this, the first q-axis magnetic flux command value Φ ｑ ＊Perform the calculation and output the result.

[0117] The first q-axis magnetic flux command calculation unit 208b is I ｄ ＊ and I ｑ ＊ And, Φ ｑ ＊ Refer to the table data that shows the correspondence with the entered I ｄ ＊ and I ｑ ＊ Based on, Φ ｑ ＊ This calculates the following. This table data is obtained in advance through testing or analysis. Alternatively, a function formula, approximation formula, design formula, etc., may be used instead of the table data.

[0118] The q-axis magnetic flux estimation calculation unit 209b calculates the current detection value I on the dc-qc axis coordinate system output by the dq coordinate transformation unit 114 (Figure 8). ｄｃ , I ｑｃ Based on this, the estimated q-axis magnetic flux Φ ｑｃ Perform the calculation and output the result.

[0119] The q-axis magnetic flux estimation calculation unit 209b is I ｄｃ and I ｑｃ And, Φ ｑｃ Refer to the table data that shows the correspondence with the entered I ｄｃ and I ｑｃ Based on, Φ ｑｃ This calculates the following. This table data is obtained in advance through testing or analysis. Alternatively, a function formula, approximation formula, design formula, etc., may be used instead of the table data.

[0120] The first q-axis magnetic flux command calculation unit 208b and the q-axis magnetic flux estimation calculation unit 209b may use the same table data, or they may use different table data. However, using the same table data can reduce the amount of work required to create the table data.

[0121] The voltage vector calculation unit 205b receives the I output from the second d-axis current command calculation unit 201. ｄ ＊＊ The second q-axis current command calculation unit 202 outputs I ｑ ＊＊The first d-axis magnetic flux command calculation unit 206b outputs Φ ｄ ＊ The d-axis magnetic flux estimation calculation unit 207b outputs Φ ｄｃ The first q-axis magnetic flux command calculation unit 208b outputs Φ ｑ ＊ The q-axis magnetic flux estimation calculation unit 209b outputs Φ ｑｃ Then, the control phase / frequency calculation unit 116 (Figure 8) outputs ω １ Based on this, the d-axis voltage command value V on the d-axis ｄ ＊＊ And the q-axis voltage command value V on the qc axis ｑ ＊＊ Calculate and output.

[0122] Figure 10 is a functional block diagram showing the configuration of the voltage vector calculation unit 205b (Figure 9).

[0123] As shown in Figure 10, the voltage vector calculation unit 205b includes proportionals 401, 403, 405, 406, 407, 408b, and 409b, adders 803, 804, 806, and 807, subtractors 805, 808b, and 809b, and multipliers 901 and 902.

[0124] In the voltage vector calculation unit 205b, V ｄ ＊＊ is, I ｄ ＊＊ Gain R １ ＊ Multiply by Φ ｄ ＊ and Φ ｄｃ The sum of the addition and subtraction values ​​(=Φ) ｄ ＊ -Φ ｄｃ ) Gain K ｄｄ Multiply by I ｑ ＊＊ Gain L ｑ ＊ and ω １ It is calculated by multiplying by and then adding or subtracting these. That is, V ｄ ＊＊ This is calculated using a voltage equation like equation (9).

[0125]

[0126] Here, gain Kｄｄ This is set, for example, by formula (10).

[0127]

[0128] In equation (10), ω ａｃｒ ＊ This is the control response angular frequency for current control.

[0129] Furthermore, in the voltage vector calculation unit 205b, V ｑ ＊＊ is, I ｑ ＊＊ Gain R １ ＊ Multiply by Φ ｑ ＊ and Φ ｑｃ The sum of the addition and subtraction values ​​(=Φ) ｑ ＊ -Φ ｑｃ ) Gain K ｄｑ Multiply by I ｄ ＊＊ proportional gain L ｄ ＊ and ω １ Multiply by ω １ Gain K ｅ ＊ It is calculated by multiplying by and adding these together. That is, V ｑ ＊＊ This is calculated using a voltage equation like equation (11).

[0130]

[0131] Here, gain K ｄｑ This is set, for example, by formula (12).

[0132]

[0133] In equation (12), ω ａｃｒ ＊ This is the control response angular frequency for current control.

[0134] Furthermore, in the voltage vector calculation unit 205b, the gain R １ ＊ , L ｄ ＊ , L ｑ ＊ , K ｅ＊ This can be a constant, or it can be expressed as a function or reference table for current command value, current detection value, magnet temperature, etc.

[0135] In equation (9), Φ in equation (5) ｄ ＊＊ The derivative of, i.e., the transient induced electromotive force, is Φ ｄ ＊ and Φ ｄｃ It is replaced by the transient induced electromotive force expressed as the difference between them. Also, in equation (11), the Φ in equation (6) is replaced. ｑ ＊＊ The derivative of, i.e., the transient induced electromotive force, is Φ ｑ ＊ and Φ ｑｃ It is replaced by a transient induced electromotive force expressed as the difference. This improves the stability of the control against disturbances.

[0136] As explained above, according to Example 2, the current command value I in the rotating coordinate system (dc-qc axis coordinate system) ｄ ＊ , I ｑ ＊ And the current detection value I in the rotating coordinate system (dc-qc axis coordinate system) ｄｃ , I ｑｃ The second current command value I in the rotating coordinate system (dc-qc axis coordinate system) generated so that it matches the above. ｄ ＊＊ , I ｑ ＊＊ And, current command value I ｄ ＊ , I ｑ ＊ The first magnetic flux command value Φ in the rotating coordinate system (dc-qc axis coordinate system) generated based on this is ｄ ＊ , Φ ｑ ＊ And, current detection value I ｄｃ , I ｑｃ The estimated magnetic flux value Φ in the rotating coordinate system (dc-qc axis coordinate system) generated based on this ｄｃ , Φ ｑｃ Based on this, the voltage command value V in the rotating coordinate system (dc-qc axis coordinate system) ｄ ＊＊ ,V ｑ ＊＊This is generated.

[0137] This enables high-precision control that takes nonlinearity into account without complicating the control system, and also improves the stability of the control system against disturbances.

[0138] Furthermore, the AC motor control device according to Embodiment 2 is not limited to PM motors, but may also be applied to other AC motors such as wound-wound synchronous motors and reluctance synchronous motors. The AC motor control device according to Embodiment 2 can also be applied when the AC motor operates as a generator.

[0139] In addition to the embodiments described above (AC motor drive system), there are other embodiments such as electric vehicles (automobiles, railway vehicles, etc.) and power generation systems (wind power, diesel power, etc.). In all of these, the control devices described in Embodiments 1 and 2 above are applied. In electric vehicles, the AC motor drive systems described in Embodiments 1 and 2 above are applied. Among electric vehicles, automobiles may be equipped with energy storage devices such as batteries that supply DC power to the power converter.

[0140] The present invention is not limited to the embodiments described above, and includes various modifications. For example, the embodiments described above are described in detail to make the present invention easier to understand, and are not necessarily limited to those having all the configurations described. Furthermore, it is possible to add, delete, or replace some of the configurations in the embodiments with other configurations.

[0141] For example, the switching elements Sup to Swn (Figures 1 and 8) are not limited to IGBTs; they can also be power MOSFETs or other types.

[0142] 101 Control device, 102 Power converter, 103 PM motor, 105 Command generator, 111 Current command calculation unit, 112, 112b Vector control unit, 113 Current detection unit, 114 dq coordinate transformation unit, 115 Magnetic pole position transformation unit, 116 Control phase / frequency calculation unit, 117 Polar coordinate transformation unit, 118 UVW coordinate transformation unit, 119 PWM controller, 121 Phase current detector, 123a, 123b Input terminals, 124 Rotor position detector, 131 Smoothing capacitor, 132 Main circuit unit, 133 Gate driver, 134 DC resistor, 201 Second d-axis current command calculation unit, 202 Second q-axis current command calculation unit, 203 Second d-axis magnetic flux command calculation unit, 204 Second q-axis magnetic flux command calculation unit, 205, 205b Voltage vector calculation unit, 206b First d-axis magnetic flux command calculation unit, 207b d-axis magnetic flux estimation calculation unit, 208b First q-axis magnetic flux command calculation unit, 209b q-axis magnetic flux estimation calculation unit, 301, 303 Proportional unit, 302 d-axis integrator, 304 q-axis integrator, 401, 403, 405, 406, 407 Proportional unit, 402 d-axis differentiator, 404 q-axis differentiator, 801, 802, 805, 808b, 809b Adder / subtractor, 803, 804, 806, 807 Adder, 901, 902 Multiplier

Claims

1. An AC motor control device for controlling a power converter that drives an AC motor, comprising: a second current command calculation unit that generates a second current command value in a rotating coordinate system such that a first current command value matches the current detection value of the AC motor; a magnetic flux command calculation unit that calculates a magnetic flux command value of the AC motor in the rotating coordinate system; and a voltage command calculation unit that generates a voltage command value for the power converter in the rotating coordinate system based on the second current command value and the magnetic flux command value.

2. The control device for an AC motor according to claim 1, wherein the voltage command calculation unit calculates the transient induced electromotive force at the voltage command value based on the magnetic flux command value.

3. The control device for an AC motor according to claim 2, characterized in that the transient induced electromotive force is the derivative of the magnetic flux command value.

4. The control device for an AC motor according to claim 1, characterized in that the magnetic flux command calculation unit calculates the magnetic flux command value based on the second current command value.

5. An AC motor control device according to claim 1, wherein the voltage command calculation unit calculates the transient induced electromotive force at the voltage command value based on the magnetic flux command value and the estimated magnetic flux value of the AC motor in the rotating coordinate system.

6. The control device for an AC motor according to claim 5, wherein the magnetic flux command calculation unit comprises a magnetic flux estimation calculation unit that calculates the magnetic flux command value based on the first current command value and calculates the estimated magnetic flux value based on the current detection value.

7. The control device for an AC motor according to claim 5, characterized in that the transient induced electromotive force is proportional to the difference between the magnetic flux command value and the magnetic flux estimate value.

8. A control method for an AC motor in which a control device controls a power converter that drives an AC motor, characterized in that: a second current command value is generated in a rotating coordinate system such that a first current command value matches the current detection value of the AC motor; a magnetic flux command value of the AC motor is calculated in the rotating coordinate system; and a voltage command value for the power converter in the rotating coordinate system is generated based on the second current command value and the magnetic flux command value.

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