Broadband weak current signal detection system and detection method

Abstract

The broadband weak current signal detection system comprises a signal acquisition mother board and an upper computer system, the signal acquisition mother board is provided with a signal input interface, a channel switching circuit, a transimpedance amplification circuit, an analog-to-digital conversion circuit, a control core board and the upper computer system, and a three-coaxial BNC interface is adopted. Electromagnetic noise interference in the environment is shielded by a metal noise shielding structure; detecting the switching of a wide measuring range and the switching of an alternating current and direct current coupling mode; weak current signals are converted into measurable voltage signals through a trans-impedance amplifier, and the voltage signals obtained after conditioning are converted into digital quantized values through an analog-to-digital converter (ADC); after being processed by a digital noise reduction algorithm, the data are stored; therefore, the real-time measurement of the broadband weak current signal is realized. The broadband weak current signal detection system has the characteristics of high resolution, good repeatability, high measurement accuracy and good real-time performance, and has good detection capability on weak current signals.

Description

Technical Field

[0001] The present invention relates to the field of aerosol detection, and in particular to a ZYNQ-based broadband weak current signal detection system and a detection method. Background Art

[0002] With the continuous progress of science and technology and the continuous improvement of industrial demand, the research and development of sensors is moving towards ultra-miniaturization and high integration. MEMS sensors and semiconductor sensors have attracted much attention due to their small size and high performance, and have become a research hotspot in the field of instrument science. Physical quantities such as electromagnetism, optics, mechanics, thermodynamics and acoustics need to be converted into measurable electrical signals with the help of sensors. A large amount of testing and verification is required in the process of sensor development, and the electrical signals output by miniaturized sensors are usually very weak. It is becoming increasingly difficult to accurately measure the output signals of miniaturized sensors, which greatly restricts the development of micro sensors. Existing weak current signal detection is usually targeted at extremely low frequency signals. Therefore, achieving accurate measurement of small-amplitude, high-frequency weak current signals has become a difficulty in the development of micro sensors.

[0003] Foreign countries have conducted relatively early research on weak current signal detection. In-depth research has been conducted on DC and lower-frequency weak current signals, and the technical system is relatively complete, and corresponding commercial instruments have been launched. In 1997, DC Thelen et al. from the University of New Mexico in the United States developed a low-noise weak current readout detection chip for detecting transient signals of photodiode pixel array chips, achieving nanoampere current detection at a bandwidth of 100Hz to 20kHz, with a circuit gain of 1.8MV / A and a resolution of 6nA. In 2016, Elnaz Ghodsevali et al. from Laval University in Canada realized weak current detection based on a potentiostat with a fully differential amplifier. When a 1MΩ feedback resistor was used, the current detection resolution was 900fA. In 2018, Minhyun Jin et al. from Dongguk University in South Korea proposed a weak current measurement method for organic light-emitting diode mobile displays. The system consists of a current integrator and an 8-bit single-slope analog-to-digital converter with a measurement range of 0 to 102.2nA and a resolution of 399.21pA. In 2021, Gaku Ogihara and others from Gunma University in Japan proposed a weak current measurement method for I / V conversion and DC-AC conversion. The DC current measurement resolution is 20pA and the measurement time is about 40ms. In 2022, Cornelius Wendt and others from the University of Hannover in Germany realized weak current measurement at a 3dB bandwidth of 48Hz based on a capacitive transimpedance amplifier, and the standard deviation of the measurement results was 3.4fA. In the field of commercial instrumentation, foreign companies currently have relatively mature weak current detection instruments, such as the Keithley 6485 produced by Keithley in the United States, the 7482 produced by ADCMT in Japan, and the SM7810 produced by Hioki in Japan. The resolution of weak current measurement of some products can reach 10 -14 A, with a wide range of measurement range, the gear can cover pA to mA current, and has a variety of calibration and compensation functions. The NIPXI-4071 digital multimeter board produced by National Instruments also has the ability to detect weak currents. The corresponding current resolution under the minimum range is 1pA, which is equivalent to traditional measuring instruments and equipment. The form of the acquisition board improves the measurement flexibility. The advent of the weak current detector has popularized weak current measurement.

[0004] In 2021, Yifan Wang and others from the University of Electronic Science and Technology of China designed a weak current measurement circuit based on an LC resonant amplifier, achieving a current amplification of 130dB gain in the range of 15Hz to 35Hz, but the capacitance of the capacitor needs to be manually adjusted to compensate for the drift of the resonant frequency. In the same year, Yan Jingchun and others from the Beijing Institute of Space Mechanics and Electronics designed a high-precision multi-channel weak current signal acquisition circuit, which achieved the measurement of weak current signals with a frequency of 1kHz and an amplitude of 4nA. In 2022, Dahai Jiang and others from the Institute of Microelectronics of the Chinese Academy of Sciences proposed an analytical model of a switched capacitor transimpedance amplifier based on charge distribution, and designed a transimpedance amplifier based on the analysis results of the model using correlated double sampling technology and cross-connection technology, with a gain of 206dB and a circuit bandwidth of 3kHz. In 2023, Liu Zhenping and others from Shandong University designed a weak current detection circuit based on a transimpedance amplifier with a range of 100nA, a circuit bandwidth of 33Hz, and a measurement relative error of less than 0.1%. At the same time, Liu Zhenping and others also analyzed and simulated the DC bias and noise levels of the transimpedance amplifier using a T-type feedback resistor network and a single feedback resistor. The results showed that under the premise of the same equivalent resistance, the DC bias and noise levels of the T-type feedback resistor are greater than those of a single resistor. In the same year, Min Huang and others from the University of Electronic Science and Technology explored the influence of different resistor materials and installation methods on the stability of weak current detection and the minimum measurable current of microcurrent detection based on the transimpedance amplifier circuit experiment. The experimental results showed that when detecting currents above 100pA, using surface mount resistors can achieve similar or better indicators. In the same year, Nan Zhou and others from Beijing Oriental Metrology and Testing Research proposed a fA-level weak current measurement circuit design and software filtering algorithm based on the transimpedance measurement method, realizing DC current measurement of 2pA to 200nA, with a resolution of 1fA and a relative error of ±1%, reaching the international level. In my country's commercial weak current detector market, the fA-level microcurrent tester HB-321 produced by Nanjing Hongbin Company has a measurement range of 200fA to 200μA, a resolution better than 0.1% of the range, and a measurement bandwidth upper limit of 30kHz. The Huace-122 picoammeter and femtoammeter tester developed by Beijing Huace Company has a measurement range of 10fA to 20mA, a 4.5-digit LED display, and a display error of 1%.

[0005] Since the circuit noise is positively correlated with the measurement bandwidth, the noise level of the measurement circuit will increase significantly as the measurement bandwidth increases. Therefore, in order to ensure a sufficient signal-to-noise ratio, the above-mentioned existing weak current detectors are mostly DC or narrow-band AC detection, and the detection bandwidth index has great limitations and cannot meet the needs of broadband weak current signal detection, such as the output signal detection of the new ionization three-electrode gas sensor under high-frequency excitation. At the same time, the existing weak current detectors are large in size and high in cost, making it difficult to integrate and promote them with new sensors.

[0006] Therefore, a broadband nanoampere-level weak current signal detection system prototype is needed to meet the wide bandwidth and high-precision weak current signal detection requirements of some new sensors, while adapting to the development trend of small integration and low cost of sensor systems. Summary of the invention

[0007] In order to solve the above-mentioned defects existing in the prior art, the purpose of the present invention is to provide a broadband weak current signal detection system and detection method based on ZYNQ. First, the stability and noise error analysis of the weak signal conditioning circuit are carried out by combining theoretical calculation with simulation, and the influence of space electromagnetic field coupling and printed circuit board (Print Circuit Board, PCB) leakage current on the measurement system is studied at the same time, and a low-noise weak current signal conditioning acquisition transimpedance amplifier circuit with reasonable protection and shielding scheme is designed. Secondly, the broadband weak signal extraction algorithm based on ZYNQ and the on-board implementation of the control core board are studied, and the collected signal is subjected to noise reduction processing. Then the weak current signal comprehensive detection system is designed and manufactured, and the hardware development of the system, the lower computer software function design and the upper computer interactive software development based on LabVIEW are completed. Finally, the detection system is calibrated with the help of the standard current source Keithley 6221, the system performance indicators are analyzed, and the system's detection ability for weak current signals is experimentally verified.

[0008] The present invention is achieved through the following technical solutions.

[0009] One aspect of the present invention provides a broadband weak current signal detection system, including a signal acquisition motherboard and a host computer system, wherein:

[0010] The signal acquisition motherboard is configured with:

[0011] The signal input interface adopts a triaxial BNC interface and a triaxial cable protection ring driving circuit with a metal noise shielding structure to shield electromagnetic noise interference in the environment; it is used to reduce the leakage current of the transmission cable;

[0012] Channel switching circuit, used for switching between wide range detection and AC / DC coupling mode;

[0013] The transimpedance amplifier circuit is used to convert a weak current signal into a measurable voltage signal by using a transimpedance amplifier, and to perform weak current signal conditioning;

[0014] The analog-to-digital conversion circuit is used to convert the conditioned voltage signal into a digital quantized value using an analog-to-digital converter ADC;

[0015] The control core board is used to send the analog-to-digital converted voltage signal to the digital signal processing system, and store the data after being processed by the digital noise reduction algorithm;

[0016] The host computer system is used to communicate with the host computer through a Gigabit Ethernet transmission path, thereby realizing real-time measurement of broadband weak current signals.

[0017] Preferably, the transimpedance amplifier converts the signal, including a feedback current integration type I / F conversion that converts the current signal into a frequency signal for measurement by time integration, and an I / V conversion scheme that converts the current signal into a voltage signal. The voltage signal obtained by the I / V conversion is further converted into a frequency pulse signal for measurement by a voltage-frequency converter.

[0018] Preferably, the feedback current integration type I / F conversion circuit comprises a current integrator, a voltage comparator and a monostable trigger connected in sequence, and the monostable trigger is connected to the current integrator via a switch circuit.

[0019] Preferably, the I / V conversion circuit first uses a transimpedance amplifier TIA to convert the weak current signal to be measured into a voltage signal, and then uses an analog-to-digital converter ADC to collect the voltage signal and digitize the signal.

[0020] As a preferred embodiment, the operational amplifier of the I / V conversion circuit based on TIA has its non-inverting input terminal grounded, its inverting input terminal virtual grounded, and its feedback resistor R F Connect the inverting input and output of the operational amplifier, and the feedback resistor R F Parallel feedback capacitor C F , the total input capacitance C is connected in parallel between the non-inverting input and the inverting input T .

[0021] Preferably, the metal noise shielding structure sets a wrapping area around the weak current signal input path, and drives the voltage of the area to the signal input voltage, so that the potential difference between the guard ring and the signal line is zero.

[0022] As a preferred embodiment, in a three-coaxial cable protection ring driving circuit, the in-phase input terminals of the two amplifiers are connected and then grounded, the inverting input terminal of the first amplifier is connected to the output terminal and then connected to the inner skin layer of the coaxial cable, and the outer skin shielding layer of the coaxial cable is grounded; the inner core of the coaxial cable is connected to the inverting input terminal of the second amplifier, and the reverse input terminal of the second amplifier is connected to the output terminal in parallel with the feedback resistor R F .

[0023] Another aspect of the present invention provides a detection method of the broadband weak current signal detection system, comprising:

[0024] A three-coaxial BNC interface is used as the detection system's broadband weak target detection current signal input interface, and a metal noise shielding structure is used to shield electromagnetic noise interference in the environment;

[0025] A channel switching circuit is set to switch the detection range and the AC / DC coupling mode;

[0026] Adopting a feedback current integration type I / F conversion that converts a weak current signal into a frequency signal for measurement by time integration, and an I / V conversion scheme that converts a weak current signal into a voltage signal, and obtaining a measurable voltage signal after I / V conversion;

[0027] The conditioned voltage signal is converted into a digital quantized value using an analog-to-digital converter (ADC) and sent to a digital signal processing system. After being processed by a digital noise reduction algorithm, the data is stored.

[0028] Set up Gigabit Ethernet transmission to communicate with the host computer to achieve wideband measurement and high-speed data transmission, thereby performing real-time measurement.

[0029] The present invention adopts the above technical solution, which has the following beneficial effects:

[0030] 1. Considering the weakness of the target detection current signal, in order to reduce the leakage current of the transmission cable, a three-coaxial BNC interface is used as the signal input interface of the detection system, and a corresponding metal noise shielding structure is designed to shield the electromagnetic noise interference in the environment.

[0031] 2. In view of the wide range and high resolution of system indicators and the AC / DC measurement requirements, a channel switching circuit is set to switch the detection range and the AC / DC coupling mode.

[0032] 3. Perform weak current signal conditioning and use a transimpedance amplifier to convert the weak current signal into a measurable voltage signal.

[0033] 4. Use the analog-to-digital converter ADC to convert the conditioned voltage signal into a digital quantization value and send it to the digital signal processing system. After being processed by the digital noise reduction algorithm, the data is stored.

[0034] 5. In view of the system's wide-band measurement characteristics and high-speed data transmission requirements, a Gigabit Ethernet transmission path is set up to communicate with the host computer to achieve real-time measurement. BRIEF DESCRIPTION OF THE DRAWINGS

[0035] The drawings described herein are used to provide a further understanding of the present invention, constitute a part of the present application, and do not constitute an improper limitation of the present invention. In the drawings:

[0036] Figure 1 It is a schematic diagram of the overall scheme architecture of the broadband weak current signal detection system of the present invention;

[0037] Figure 2 It is a block diagram of a feedback current integration type I / F conversion circuit of a broadband weak current signal detection system of the present invention;

[0038] Figure 3 It is an I / V conversion circuit solution of the broadband weak current signal detection system based on a transimpedance amplifier of the present invention;

[0039] Figure 4 It is a schematic diagram of capacitance distribution parameters of an I / V conversion circuit based on a transimpedance amplifier in a broadband weak current signal detection system of the present invention;

[0040] FIG5 is a schematic diagram of a compensation scheme for the TIA circuit stability of a broadband weak current signal detection system of the present invention, (a) is a schematic diagram of a compensation circuit, and (b) is a schematic diagram of introducing a feedback capacitor C F Bode plot after compensation;

[0041] Figure 6 It is the noise model of the transimpedance amplifier circuit of the broadband weak current signal detection system of the present invention;

[0042] Figure 7 It is a schematic diagram of AC electric field coupling of the TIA circuit of the present invention;

[0043] Figure 8 It is a schematic diagram of a three-coaxial cable protection ring driving circuit of a broadband weak current signal detection system of the present invention;

[0044] FIG9 is a diagram of the power supply ripple of the broadband weak current signal detection system OPA858 of the present invention, (a) is a +2.5V power supply ripple, (b) is a -2.5V power supply ripple;

[0045] Fig.10 This is a schematic diagram of the connection of the circuit for testing the working correctness of the broadband weak current signal detection system ADS9110 of the present invention;

[0046] Fig.11 It is the ADC flicker code value test result of the broadband weak current signal detection system of the present invention under the condition of 0 input;

[0047] Fig.12 This is the Gigabit Ethernet bandwidth test result of the broadband weak current signal detection system of the present invention;

[0048] Fig.13 The working characteristic curves and fitting results of the weak current signal detection system of the broadband weak current signal detection system of the present invention under the DC and AC coupling input modes, (a) is the DC coupling 2μA range, (b) is the DC coupling 20μA range, (c) is the DC coupling 200μA range, (d) is the AC coupling 2μA range, (e) is the AC coupling 20μA range, and (f) is the AC coupling 200μA range;

[0049] Fig.14 These are the static characteristic test results of the broadband weak current signal detection system under DC and AC coupling modes of the present invention, (a) is a DC coupling 2μA range, (b) is a DC coupling 20μA range, (c) is a DC coupling 200μA range, (d) is an AC coupling 2μA range, (e) is an AC coupling 20μA range, and (f) is an AC coupling 200μA range;

[0050] Fig.15 The amplitude-frequency characteristic curve of the AC coupling input of the broadband weak current signal detection system of the present invention, (a) is 2μA range, (b) is 20μA range, and (c) is 200μA range. DETAILED DESCRIPTION

[0051] The present invention will be described in detail below in conjunction with the accompanying drawings and specific embodiments. The exemplary embodiments and descriptions of the present invention are used to explain the present invention but are not intended to limit the present invention.

[0052] like Figure 1 As shown, a broadband weak current signal detection system provided by an embodiment of the present invention includes a signal acquisition motherboard and a host computer system, wherein: the signal acquisition motherboard is configured with a signal input interface, a channel switching circuit, a transimpedance amplifier circuit, an analog-to-digital conversion circuit and a control core board. Among them:

[0053] The signal input interface adopts a three-coaxial BNC interface and is equipped with a metal noise shielding structure to shield the electromagnetic noise interference in the environment; it is used to detect weak target current signals and reduce the leakage current of the transmission cable.

[0054] The channel switching circuit is used to switch between wide detection ranges and AC and DC coupling modes.

[0055] The transimpedance amplifier circuit is used to convert a weak current signal into a measurable voltage signal by using a transimpedance amplifier, so as to perform weak current signal conditioning.

[0056] The analog-to-digital conversion circuit is used to convert the conditioned voltage signal into a digital quantized value using an analog-to-digital converter ADC.

[0057] The control core board is used to send the voltage signal after analog-to-digital conversion to the digital signal processing system, and store the data after being processed by the digital noise reduction algorithm.

[0058] The host computer system is used to communicate with the host computer through a Gigabit Ethernet transmission path, thereby realizing real-time measurement of broadband weak current signals.

[0059] There are two main measurement schemes for the weak current signal transimpedance amplifier circuit of the signal detection system of the present invention: one is to convert the current signal into a frequency signal for measurement by time integration, namely the I / F conversion scheme; the other is to convert the current signal into a voltage signal, namely the I / V conversion scheme, and the voltage signal obtained by the I / V conversion is further converted into a frequency pulse signal through a voltage-frequency converter for measurement.

[0060] The I / F conversion circuit is composed of Figure 2 As shown, the feedback current integration type I / F conversion circuit includes a current integrator, a voltage comparator, a monostable trigger and a switch circuit.

[0061] The main working principle of the circuit is: when the switch is disconnected, the weak current I to be measured charges the feedback capacitor C in the current integrator, and the integrator outputs a voltage signal U1. The U1 input voltage comparator circuit is compared with the threshold voltage Uth. It is stipulated that when U1>Uth, the output U2 of the comparator is high level, and when U1<Uth, U2 is low level. When the charging process continues, at the moment when U1 reaches Uth, U2 changes from low level to high level, and the monostable trigger outputs a pulse signal U3, triggering the switch circuit to operate. When the switch is closed, the feedback capacitor C in the current integrator is quickly discharged to zero, and the U2 level flips again, waiting for the next charging voltage to reach the threshold voltage. This charging and discharging process makes the measurement circuit output a pulse square wave with a frequency proportional to the current I to be measured. The larger I is, the shorter the time required for the capacitor to charge to the threshold voltage Uth, and the higher the frequency of the pulse signal output by the measurement circuit. Therefore, the amplitude of the weak current signal I to be measured can be obtained by counting pulses in the subsequent processing link. The feedback current integration type I / F conversion circuit realizes the digitization of weak current signals by means of pulse counting. It can maintain the advantage of linear conversion while successfully avoiding the limitation of ADC on the measurement system. Moreover, since the feedback element used in the circuit is capacitor, it has a lower temperature coefficient than resistor and thus has better stability. In addition, the integral measurement form can also suppress low-frequency noise interference to a certain extent.

[0062] I / V conversion scheme: The I / V conversion scheme is to first use a transimpedance amplifier to convert the weak current signal to be measured into a voltage signal, and then use an ADC to collect the voltage signal and digitize the signal. The basic principle of the I / V conversion circuit based on TIA is as follows: Figure 3 shown.

[0063] Figure 3 In the circuit shown, the non-inverting input of the operational amplifier is grounded. For an ideal operational amplifier, its inverting input is virtual grounded. The measured current I S Isolation from the measurement circuit, I S All flows through the feedback resistor R F , so the output voltage Uo=R F ×I S But in reality, due to the input bias current I B- And input offset voltage Uos, the actual output voltage Uo of the transimpedance amplifier is:

[0064] U o =I S ·R F +I B- ·R F +U os

[0065] In the formula, I S is the current to be measured, i.e., the input current of the transimpedance amplifier; IB It is the bias current of the inverting input of the operational amplifier.

[0066] Basics of TIA circuit stability and noise analysis: There are parasitic capacitances introduced by non-ideal op amps and signal transmission cables in TIA circuits, which will affect the circuit stability. Generally, the feedback resistor R F The feedback capacitor C is connected in parallel at both ends F The capacitance distribution parameters of the transimpedance amplifier circuit are as follows: Figure 4 shown.

[0067] Figure 4 In the circuit shown, the inverting input of the operational amplifier is connected to a capacitor C S The back is grounded, and the capacitor C is connected in parallel to the ground terminal of the common-mode input terminal. DIFF and capacitor C CMI , capacitor C CMI One end is grounded, capacitor C DIFF Connect capacitor C CM2 Then ground.

[0068] The total capacitance at the input of the transimpedance amplifier is the parasitic capacitance of the signal transmission cable C S Differential C DIFF And the common mode input capacitor C CM1 , C CM2 The op amp's in-phase terminal is connected to the AC ground, C CM1 is short-circuited, so the total input capacitance C of the transimpedance amplifier circuit T for

[0069] C T =C s +C DIFF +C CM2

[0070] In the formula, C S is the parasitic capacitance of the signal transmission cable, C DIFF is the differential amplifier capacitance, C CM2 is the common-mode amplifier capacitance.

[0071] TIA circuit input capacitance C T It acts as a shunt, same as the feedback resistor R F The zero point formed on the noise gain curve will cause the op amp open-loop gain and noise gain curves to intersect at an approach speed of -40dB / dec, causing the op amp to self-oscillate. To ensure circuit stability, the feedback capacitor C F With R F The compensation scheme is shown in Figure 5(a) and Figure 5(b) shows the introduction of feedback capacitor C. F Bode plot after compensation.

[0072] In order to ensure the stability of the I / V conversion circuit based on TIA under the specified bandwidth, the feedback capacitor C F The following equation gives the value of C that meets the TIA bandwidth requirement: F Calculation formula:

[0073]

[0074] In the formula, f BW is the required closed-loop bandwidth of the TIA circuit, R F is the feedback resistor.

[0075] The three main noise sources of the TIA circuit are the op amp's n 、e n Noise and feedback resistance R F The Johnson noise has a noise model such as Figure 6 As shown, the op amp’s non-inverting input and output terminals are connected in parallel with feedback capacitor C F With the feedback resistor R F , the reverse input terminal parallel capacitor C CM2 With C DIFF And the parasitic capacitance C of the signal transmission cable S , the reverse input terminal is connected to the current signal source, R S is the internal resistance of the current signal source, generally greater than 10 8 Ω.

[0076] In the I / V conversion scheme, the ADC can maintain a relatively stable quantization resolution within a wide frequency range, thereby realizing real-time acquisition of high-frequency input signals and meeting the 100kHz wide-band measurement requirements. Therefore, an I / V conversion scheme based on a transimpedance amplifier is used to condition weak current signals.

[0077] Figure 7 The effect of the alternating electric field coupling to the TIA detection input terminal is shown. To simplify the calculation, it is assumed that the power line l2 carrying the 220V AC current is parallel to the transmission line l1 of the input current signal i in the weak current signal detection system. The distributed capacitance between the two is represented by C d Indicates that the input resistance of the op amp in TIA is represented by R i express.

[0078] Assuming that the distance D between l1 and l2 is much larger than the radius r of the current transmission line, the distributed capacitance Cd can be expressed as:

[0079] C d =πε r ε0l / ln(D / r)

[0080] Due to the existence of AC power supply, the distribution environment of electromagnetic field in space is constantly changing. The changing electric field strength will bring coupling noise interference of different voltage levels to the detection system, resulting in fluctuations in measurement results, which cannot be removed by linear circuit calculation methods. Therefore, when detecting weak current signals, shielding design is required to minimize interference from coupling noise of electromagnetic field in space.

[0081] It uses a three-coaxial BNC interface and is equipped with a metal noise shielding structure to shield electromagnetic noise interference in the environment; the shielding and protection solution: a low leakage current PCB guard ring solution must be used when detecting weak current signals at nA and below levels, and PCB protection technology is used to deal with PCB leakage current problems. The PCB guard ring is also called drive protection, which is connected to a low impedance voltage source in the circuit to eliminate the leakage current generated by the high impedance node. For this detection system, the idea is to set a wrapping area around the weak current signal input path, and drive the voltage in this area to the signal input voltage, so that the potential difference between the guard ring and the signal line is zero, thereby ensuring that the current in the guard ring is not disturbed by the PCB leakage current.

[0082] Figure 8 The three-coaxial cable protection ring driving circuit is shown. The in-phase input terminals of the two amplifiers are connected and then grounded. The inverting input terminal of the first amplifier is connected to the output terminal and then connected to the inner layer of the coaxial cable. The outer shielding layer of the coaxial cable is grounded. The inner core of the coaxial cable is connected to the inverting input terminal of the second amplifier. The reverse input terminal of the second amplifier is connected to the output terminal in parallel with the feedback resistor R F .

[0083] Figure 8 In the circuit, the endothelial signal layer is connected to the PCB guard ring, and the signal input path voltage is driven to the guard ring by a follower, thereby constructing an equipotential body around the signal transmission line to isolate the introduction of PCB leakage current. In addition to the leakage current interference from the transmission medium, the weak current signal may also be interfered by the noise of electromagnetic field coupling in the space where the system is located during the transmission process. Alternating electric field coupling is a common capacitive coupling interference, which is mainly transmitted through the distributed capacitance between the signal transmission line and the alternating power supply in space, causing the voltage change of the power supply to be superimposed on the input end of the weak current acquisition circuit, thereby causing the output of the detection circuit to be offset or saturated, affecting the performance of the detection system.

[0084] The weak signal detection method based on wavelet denoising of the present invention: Wavelet analysis, as a time-frequency localized signal processing method, can take into account the noise reduction requirements of periodic signals, transient signals and multi-frequency signals, and can perform real-time calculations and processing. It shows unique advantages in the noise reduction detection of broadband weak current signals targeted by the detection system of the present invention. The principle of wavelet denoising is that in the wavelet domain, when the decomposition scale increases, the amplitude of the wavelet coefficient of f(i) is basically constant, while the amplitude of the wavelet coefficient of n(i) decays rapidly. The corresponding denoising criterion is constructed to minimize or remove the n(i) wavelet coefficient while retaining the f(i) wavelet coefficient as much as possible. The processed wavelet coefficients are reconstructed to obtain an optimal approximation of f(i) under the error estimation criterion. Thus, signal denoising is achieved. The present invention adopts a threshold method. According to the difference in the threshold setting method, there are the following two wavelet analysis denoising methods: (1) Forced denoising: all high-frequency coefficients of each level of wavelet decomposition are set to zero, and then the remaining wavelet coefficients are used to reconstruct the signal. This method is simple to operate and the reconstructed signal is smooth, but the high-frequency components of the original signal may be lost. (2) Threshold denoising: according to the empirical rule or the statistical characteristics of the original signal, a global threshold or a single-layer threshold at different scales is set, and threshold value processing is performed on the high-frequency coefficients of each level in the signal wavelet decomposition result, such as the classic hard threshold function, soft threshold function and various improved threshold functions. This method utilizes the characteristics of uniform and dense noise signals in the high-frequency part, which is closer to the actual denoising requirements. For the broadband weak current detection system designed by the present invention, the measurement results only focus on the measurement bandwidth specified by the system. For the high-frequency signal components that may exist outside the band, they are uniformly regarded as noise. Therefore, by controlling the number of wavelet decomposition layers to match the measurement bandwidth, and then combining the forced denoising processing method to limit the out-of-band signal, it can meet the system design requirements.

[0085] The overall scheme architecture of the weak current signal detection system is as follows Figure 8 As shown in the figure, the arrows indicate the direction of signal transmission. As the brain of the system, the system control core needs to command and dispatch all functional units in the system in a unified manner, so that each module in the system can cooperate with each other according to strict logic and become a unified organic whole. Figure 8 The control core of the detection system shown needs to complete the switching control of the measurement channel, ADC driving, data reception and processing core logic design, data storage and on-chip transmission control, and Gigabit Ethernet communication control between the host computer. Based on the above-mentioned function demand analysis for control core computing resources, bus resources, storage resources and peripheral resources, the weak current signal detection system designed by the present invention intends to use the ZYNQ XC7Z020 chip produced by Xilinx as the main control chip. The chip is equipped with FPGA and dual-core Cortex-A9 processors at the same time, has rich I / O resources and programmable logic resources, and can flexibly respond to system development needs.

[0086] The weak current detection experimental system of the present invention consists of four parts: a weak current source, an external power supply, a broadband weak current signal detection system acquisition board and a PC host computer. In the experimental system, the external power supply provides a +12V DC voltage input for the detection system, and the weak current source provides an AC / DC standard test current with a frequency of DC to 100kHz and an amplitude of 1nA to 200μA.

[0087] The weak current signal to be measured enters the detection circuit through the three-coaxial BNC interface, and the transimpedance amplifier, as the first-stage conditioning circuit, converts it into a measurable voltage signal. Different measurement channels correspond to different measurement modes and measurement ranges, and their selection is manually controlled by the host computer. The host computer sends the channel switching instruction to the FPGA, and then the FPGA controls the relay group through the general-purpose input / output (GPIO), thereby completing a measurement channel switch. Since the input bias voltage of the op amp will cause the zero point offset of the TIA circuit, and the circuit zero point will drift with the increase of usage time and the change of device temperature, a zero calibration module is designed to eliminate zero bias and zero drift, and is connected to the in-phase input terminal of the transimpedance amplifier. Its working mechanism is that after the host computer sends the zero calibration code value and zero calibration instruction to the FPGA, the FPGA controls the zero calibration module through the SPI interface, and the detection system completes the zero calibration. The fully differential amplifier is used as the second-stage conditioning circuit to drive the ADC, complete the matching of the TIA output signal with the ADC range, and convert the single-ended signal to be measured into the differential input signal required by the ADC. This differential measurement method can reduce the influence of common-mode noise and has a strong suppression ability for interference signals, thereby improving the anti-interference ability of the entire system. At the same time, in order to further attenuate the circuit noise without adding active devices, the fully differential amplifier also realizes the function of second-order Bessel low-pass filtering. An anti-aliasing low-pass filter circuit is set before the analog-to-digital converter. Its main purpose is to avoid the phenomenon of spectrum aliasing in sampling and to suppress high-frequency noise interference to a certain extent. The ADC converts the output voltage of the fully differential amplifier into a digital quantization value, transmits data with the PL end of the master control ZYNQ7020 through the multiSPI interface, and sends the sampling result to the FPGA for further digital noise reduction processing. Based on the physical layer network chip, an Ethernet communication interface is added between the acquisition board of the detection system and the host computer.

[0088] The following is the measurement of the voltage value of each power supply in the system and the ripple of the simulated power supply.

[0089] 1. Power module test:

[0090] The main purpose is to measure the voltage value of each power supply in the system and the ripple of the analog power supply to ensure that the power supply requirements of each module in the system are met. In the design process of the acquisition board of the detection system, considering the convenience of testing, test pads for each power supply, analog signal ground, and digital signal ground in the power module are reserved when drawing the PCB board.

[0091] When measuring the analog power supply ripple, the digital oscilloscope MDO4104C is set as follows: set the coupling mode to AC coupling; set the input impedance to 1MΩ; turn on the 20M bandwidth limit to filter out high-frequency noise. Use the oscilloscope spring probe to measure the power supply ripple to make the measurement loop as short as possible and reduce the impact of external interference on the measurement results. Adjust the appropriate amplitude and time scale, save the measured power supply ripple waveform and read it. Taking the power supply ±2.5V of the core of the analog circuit, OPA858, as an example, the power supply ripple waveform obtained by testing is shown in Figure 9(a) and (b). As shown in Figure 9(a) and (b), the ripple of the +2.5V power supply of OPA858 is 1mV, and the ripple of -2.5V is 1.36mV.

[0092] According to the above test method, the ripple test is performed on the power supply of the analog circuit in the system, and the test results are recorded in Table 1.

[0093] Table 1 Test results of power module of weak current signal detection system

[0094]

[0095]

[0096] The test results in Table 1 show that the ripple error of the power supply of the analog circuit of the weak current signal detection system is within 0.1%.

[0097] 2. Functional test of zero calibration circuit:

[0098] This includes the correctness test of the linear variable digital potentiometer MAX5482 driver and the correctness test of the digital zeroing circuit. During the test, the weak current signal detection system acquisition board and the host computer are connected using a 10G Ethernet cable. After the acquisition board is powered on, first check whether the MAX5482 chip's 1-pin VDD and 8- and 16-pin VSS power supply are normal. According to the device manual, in the bipolar power supply mode, MAX5482 requires the VDD input range to be +2.5V to +5.25V, the VSS input range to be -2.5V to -0.2V, and requires (VDD-VSS) ≤ +5.25V. The actual voltages of the VDD and VSS pins measured in the test were 2.49V and -2.48V, respectively, indicating that the MAX5482 power supply meets the device requirements. Then, the ARM control program and FPGA programmable logic program are burned to the acquisition board of the detection system, the host computer is started and communication with the acquisition board is established, the zero calibration code value sent to the digital zero calibration module is changed by the host computer, and the output voltage value of pin 11 of MAX5482 under different zero calibration code values ​​is recorded, and the measurement results are listed in Table 2.

[0099] Table 2 Digital zero calibration circuit test results

[0100]

[0101] The test results show that the MAX5482 driver program written by the present invention can correctly drive the digital zeroing circuit to output the expected voltage, the relative error between the actual output voltage and the target output voltage does not exceed 5%, and the digital zeroing circuit can work normally as expected.

[0102] 3. Relay drive circuit test:

[0103] The current working state of the acquisition board of the detection system is closely related to the on-off state of the relay. The switching of the signal input coupling mode and the switching of the measurement range are inseparable from the on-off control of the relay G3VM-21MT. Specifically, the switching action of the relay is driven by the transistor array TPL7407, and the control switching is realized by changing the FPGA pin level through the ARM control program. In order to verify whether the relay is correctly operating according to the system preset, the host computer and the acquisition board are started and the communication channel is waited for to be established. The control instruction frame is set through the front panel of the host computer, and the system is specified to work in the 2μA measurement range and DC coupling input mode. If the system functions normally, the relay state in the multi-function TIA circuit should be K1 closed, K2, K3, K4 and K5 disconnected. After the setting is completed, the system communication is started on the host computer, and the above preset instruction frame is sent to the system acquisition board. At this time, the digital multimeter VICTORVC890D is adjusted to the ohm range to measure the path resistance between the 4th and 6th pins of the relay G3VM-21MT, and then the relay conduction is judged by the path resistance. The test results are recorded in Table 3.

[0104] Table 3 Relay on / off state test results

[0105]

[0106]

[0107] Since the upper limit of the ohm range measurement of the digital multimeter VICTOR VC890D is 20MΩ, the over-range result is recorded as “>20MΩ” in Table 3 during the test. According to the device manual of the relay G3VM-21MT, the standard value of its on-resistance in the on state is 0.4Ω, and the minimum value of its input and output insulation resistance in the off state is 100MΩ. The test results in Table 3 show that when the system works under the specified test conditions, the relay K1 in the multi-function TIA circuit is closed and the other relays are disconnected, which is consistent with the expected state. Therefore, it can be concluded that the relay drive circuit designed in the present invention can work normally, and the relay action control program is correct and reliable.

[0108] 4. Analog-to-digital conversion circuit test:

[0109] The analog-to-digital conversion circuit plays a key role in the detection system. Its conversion error directly affects the accuracy of the detection result. The accuracy of the conversion result needs to be verified by actual measurement. The experimental circuit connection diagram is as follows: Fig.10 shown.

[0110] The analog-to-digital conversion module of the detection system of the present invention is built based on the unipolar fully differential analog input device ADS9110 chip, and a +5V standard reference voltage UREF is provided to it through a reference power supply chip REF6050 outside the chip. The device manual shows that the common mode input voltage allowed by the chip is UREF / 2, and the differential mode input voltage range is ±UREF. In order to verify whether the chip and its peripheral circuits are working properly and test the flash code value of the ADC at the same time, the output of the +2.5V reference power supply chip REF6025 in the detection system is connected to the differential input pins AINP and AINM of the ADS9110 through a flying wire, providing an input signal with a differential mode voltage of 0V and a common mode voltage of UREF / 2 for the analog-to-digital conversion circuit.

[0111] Fig.10 Under the circuit connection mode shown, the theoretical digital conversion code value of the analog-to-digital conversion circuit should be 0. Connect the host computer and the acquisition board and establish communication, provide the setting instruction frame to run the ADS9110 driver IP core program in ZYNQ 7020, and then start a single sampling on the host computer and record the original data of the ADC conversion code value with a sampling time of 1s. Fig.11 The resulting data of 0.5μs captured from this is shown.

[0112] Depend on Fig.11It can be seen that when the differential input is 0V, the decimal value conversion result obtained by the analog-to-digital conversion circuit jumps in the range of -2 to 1, the ADC flicker code value error is 3LSB, and the corresponding analog voltage value is about 114.44μV, which represents the peak-to-peak level of the ADC conversion voltage noise. It is converted back to the input end of the detection system of the present invention, and the equivalent input current error corresponding to the system 2μA range is 0.05nA, which will not affect the target resolution of the system. At the same time, the experimental results also mean that the ADS9110 driver IP core program written by the present invention can drive the analog-to-digital conversion circuit to work normally and the system data transmission path can operate normally as expected.

[0113] 5. Gigabit Ethernet interface circuit bandwidth test:

[0114] In order to verify that the communication rate of the Gigabit Ethernet interface circuit designed in the present invention can meet the design requirements, the Gigabit network port of the PC is connected to the network port of the acquisition board using a 10 Gigabit network cable. After the communication link is established, the network performance test tool Iperf is used to test the TCP communication bandwidth. In order to cooperate with the Iperf tool to complete the Ethernet bandwidth test, the LwIPTCP Perf Server routine provided by LwIP is run on the acquisition board. The program creates a TCP server on the acquisition board and listens to TCP client connections. At the beginning of the test, the routine displays the client connection information and temporary and average TCP statistics for data transmission through the serial port.

[0115] In the Iperf tool, execute the command "bin / iperf.exe-c 192.168.0.13-P 1-i 1-p 1013-Cf m-t30" to start the Ethernet bandwidth test with a communication duration of 30 seconds. The transmission bandwidth test results are as follows: Fig.12 shown.

[0116] The test results show that the average maximum communication rate of the Gigabit Ethernet designed by the present invention is 943.67 Mbps, which fully meets the required 64 Mbps transmission requirement.

[0117] 6. Range calibration:

[0118] The detection system of the present invention is a linear system, so the system working characteristic equation can be obtained by performing least square linear fitting on the calibration data. Fig.10As shown, connect the current source, system acquisition board and host computer, use Keithley6221 current source to generate standard current signal I, set the corresponding instruction frame on the host computer to switch the system input coupling channel and range channel, read the system sampling voltage value U and record it. Considering the coverage between the ranges and the uniformity within the range, the selection scheme of the calibration points of each range is determined as follows: Under the 2μA range, set the number of calibration points of the detection system to 22 points, and the specific current values ​​are 1, 3, 5, 7, 9, 10, 15, 20, 25, 50, 75, 100, 150, 200, 250, 500, 750, 1000, 1250, 1500, 1750, 1999, 2000 (in nA); under the 20μA range, set the number of calibration points to 20 points, and the specific current values ​​are 10, 20, 25, 50, 100, 200, 500, 1000, 1 250, 1500, 1750, 2000, 3500, 5000, 7500, 10000, 12500, 15000, 17500, 20000 (in nA); Under the 200μA range, the calibration points are set to 26 points. The specific current values ​​are 0.01, 0.02, 0.25, 0.5, 0.1, 0.2, 0.5, 1, 2, 5, 10, 12.5, 15, 17.5, 20, 35, 50, 65, 75, 85, 100, 110, 125, 140, 150, 160, 175, 185, 200 (in μA).

[0119] When the system is calibrated in DC coupling input mode, the Keithley 6221 is controlled to output the DC current signal corresponding to each calibration point. When the system is calibrated in AC coupling input mode, the Keithley 6221 is controlled to output a sinusoidal signal with an amplitude of the calibration point value and a frequency of 100 Hz. The system calibration results are plotted on Fig.13 In (a)-(f), the fitting equations are given together with the figures.

[0120] 7. Forward and reverse stroke test experiment:

[0121] The static characteristics of the detection system characterize the relationship between the system output and input when the measured value is in a constant or slowly changing state. In order to accurately evaluate the measurement capability and error level of the broadband weak current signal detection system, the static characteristics test experiment of the system was carried out under DC coupling and AC coupling input modes. The DC coupling input controls the Keithley6221 to output a DC current signal, and the AC coupling input controls the Keithley6221 to output a 100Hz standard sinusoidal current signal. Before the experiment begins, the fitting straight line parameters obtained from the calibration experiment are input into the host computer processing program, so that the detection current I decThe target resolution of the detection system of the present invention in the three ranges of 2μA, 20μA and 200μA is 0.1nA, 1nA and 10nA respectively. According to the definition of the number of digits displayed by the digital instrument, the current detection system of the present invention belongs to a four-and-a-half-digit meter. Considering the uniformity of the range coverage, 9 current detection points are selected in each range, among which the current values ​​of the detection points in the 2μA range are 0.0001, 0.0010, 0.0100, 0.0500, 0.1000, 0.5000, 1.0000, 1.50 00, 2.0000 (unit: μA); the current values ​​of the detection points in the 20μA range are 0.001, 0.010, 0.100, 0.500, 1.000, 5.000, 10.000, 15.000, 20.000 (unit: μA); the current values ​​of the detection points in the 200μA range are 0.010, 0.100, 1.000, 5.000, 10.000, 50.000, 100.000, 150.000, 200.00 (unit: μA). Five forward and reverse stroke test experiments were carried out in three ranges, and the measurement results of the broadband weak current signal detection system were plotted on Fig.14 (a)-(f). Fig.14 The test results (a)-(f) show that the detection system of the present invention has good measurement repeatability and system linearity in all measuring ranges.

[0122] 8. System bandwidth test:

[0123] In order to test the measurable signal bandwidth of three different ranges of the weak current signal detection system designed by the present invention, a Keithley 6221 current source is used to output a standard sinusoidal AC current signal of specified amplitude and different frequencies, and the sampling voltage value corresponding to each current measurement point of the detection system is read and recorded on the host computer. Considering the range coverage, the current test point selection scheme is as follows: the signal amplitude is half of the test range, namely 1μA, 10μA and 100μA, and the signal frequency is set to 100Hz, 1kHz, 5kHz, 10k Hz, 25kHz, 50k Hz, 75k Hz and 100kHz. After the test is completed, the sampling voltage value U corresponding to the 100Hz electrical input current signal of the detection system is used. 100Hz For reference, the relative gain G(f) of the detection system at the remaining frequency points is calculated according to formula (7) and the results are plotted on Fig.15 (a)-(c).

[0124] G(f)=20lg(U f / U 100Hz ) (7)

[0125] Where U fRepresents the sampling voltage value corresponding to the current test point of the detection system at a frequency other than 100Hz.

[0126] Fig.15 The test results (a)-(c) show that, in the 2μA range, the relative gain of the detection system at the 100kHz frequency current test point is -3.08dB, and the system bandwidth is about 100kHz, which is consistent with the target bandwidth specification. In the 20μA range and 200μA range, the relative gain of the detection system at the 100kHz frequency current test point is -1.08dB and -1.26dB respectively, and the system bandwidth exceeds the theoretical design bandwidth target. After analysis, the difference comes from the nominal error of the feedback capacitor in the 20μA range and 200μA range channels of the transimpedance circuit. This result does not affect the system from achieving the expected 100kHz bandwidth design target.

[0127] The metrological capability of the detection system of the present invention is specifically evaluated from aspects such as system range, system error, accuracy level, and measurement uncertainty.

[0128] 1) The range of each range of the broadband weak current signal detection system X FS

[0129] The measuring range of the detection system is X FS Refers to the algebraic difference between the upper and lower limits of measurement. The detection system of the present invention has three range gears, corresponding to the measurement ranges of -2μA~2μA, -20μA~20μA and -200μA~200μA, respectively. The corresponding measurement range is X FS They are 4μA, 40μA and 400μA respectively, and the switching of the range is controlled by the host computer instructions.

[0130] 2) Maximum reference error γ and accuracy level of each range of broadband weak current signal detection system

[0131] According to the relevant provisions of the national standard "GB / T 13283-91", the reference error of the instrument can be used to measure the accuracy level of the instrument. For instruments with multiple measuring ranges, each range can belong to the same accuracy level or different accuracy levels. The maximum reference error refers to the maximum absolute error of the measurement value of the detection system |Δx| max With system range X FS The ratio of, usually expressed as a percentage:

[0132] γ=|Δx| max / X FS ×100% (1)

[0133] Substituting the system static characteristic test data into formula (1), the maximum reference error γ of each range of the weak current signal detection system is calculated, and the results are statistically summarized in Table 4.

[0134] Table 4 Maximum reference error γ of broadband weak current signal detection system

[0135]

[0136] The calculation results in Table 4 show that the reference error of the weak current signal detection system in the 2μA range does not exceed 0.03%, the reference error in the 20μA range does not exceed 0.05%, and the reference error in the 200μA range does not exceed 0.04%. Referring to the "GB / T 13283-91" standard, the accuracy level of the detection system of the present invention is 0.05.

[0137] 3) Resolution R of broadband weak current signal detection system x

[0138] According to the relevant provisions of the national standard "JJF 1094-2002 Measuring Instrument Evaluation Characteristics", the resolution of the measuring instrument can be evaluated by the minimum indication difference that can be recognized by the display or reading device. If there is a digital display device, the indication difference when the lowest digit changes by one step can be regarded as the resolution. The detection system of the present invention is a four-and-a-half-digit meter. When the range is 2μA, 20μA and 200μA, the indication difference when the lowest digit changes by one word is 0.0001μA, 0.001μA and 0.01μA respectively, and the corresponding resolution is 0.1nA, 1nA and 10nA respectively. The average of ten test points of the system's response to 0.1nA, 1nA and 10nA input signals under three ranges is taken as the measurement result, and the corresponding indication error, maximum allowable error, relative error and reference error are calculated and listed in Table 5.

[0139] Table 5 Summary of the resolution test results of the broadband weak current signal detection system

[0140]

[0141]

[0142] The calculation results in Table 5 show that the indication errors of the resolution test of each range of the detection system of the present invention are all within the maximum allowable error limit. However, when measuring the input current of 1nA (the resolution of this range is 0.001μA) under the DC coupled input 20μA range, the relative error of the measurement result is relatively large. The possible reason is that the human operation introduced gross error interference during the system calibration process, resulting in a large offset between the calibration point and the actual working characteristics of the system. The indication errors of the remaining test points are all 0, and the overall resolution test results of the system show high accuracy.

[0143] 4) Uncertainty of broadband weak current signal detection system

[0144] When evaluating the measurement uncertainty of the detection system, the present invention adopts the standard source method and refers to "JJF1059.1-2012 Evaluation and Expression of Measurement Uncertainty" to evaluate the AC current measurement and DC current measurement processes under three ranges.

[0145] Similarly, the measurement uncertainty of the broadband weak current signal detection system for the specified AC and DC currents in the remaining ranges is calculated according to the above method and summarized in Table 6.

[0146] Table 6 Measurement uncertainty of broadband weak current signal detection system

[0147]

[0148]

[0149] 5) Conformity assessment of indication error of broadband weak current signal

[0150] According to the relevant provisions of the national standard "JJF1094-2002", when evaluating the conformity of the characteristics of the measuring instrument, if the expanded uncertainty of the indication error U 95 The ratio of the absolute value of the maximum permissible error of the instrument MPEV satisfies:

[0151]

[0152] The influence of the measurement uncertainty of the indication error can be ignored. At this time, the qualified criterion for the indication error of the evaluated instrument is:

[0153] |Δ|≤MPEV(3)

[0154] Where Δ represents the indication error of the instrument being evaluated.

[0155] According to the maximum allowable error requirement of "±(0.2% indication + 0.02% range)", the maximum allowable error absolute values ​​MPEV of the detection system of the present invention for the three half-range test current points of 1μA, 10μA and 100μA are 2.8nA, 28nA and 280nA respectively. From Table 6, the expanded uncertainty U of the corresponding current point under DC coupling 95 They are 2.3nA, 18nA and 200nA respectively. The expanded uncertainty U of the corresponding current point under AC coupling 95 The expanded uncertainty U of the six test current points is 2.3nA, 17nA and 170nA respectively. 95 and the maximum allowable error absolute value MPEV do not satisfy formula (2). At this time, the qualified criterion for the indication error of the detection system is changed to:

[0156] |Δ|≤MPEV-U 95 (4)

[0157] The criterion for the unqualified system indication error is:

[0158] |Δ|≥MPEV+U 95 (5)

[0159] The area to be determined for system indication error compliance is:

[0160] MPEV-U 95 ≤|Δ|≤MPEV+U 95 (6)

[0161] According to equations (4) to (6), the conformity of the indication error of the half-scale current test point of the broadband weak current signal detection system is determined, and the determination results are summarized in Table 7.

[0162] Table 7. Indication error of half-scale current test point of broadband weak current signal detection system

[0163]

[0164] The calculation results in Table 7 show that the indication error of the half-scale current test point of the detection system of the present invention in each range meets the error design expectation. This result is highly representative for the overall performance evaluation of the detection system. The evaluation results show that the broadband weak current signal detection system designed by the present invention meets the expected target of the maximum allowable error of "±(0.2% indication + 0.02% range)".

[0165] The present invention is not limited to the above-mentioned embodiments. On the basis of the technical solution disclosed in the present invention, technicians in this field can make some substitutions and deformations to some technical features therein according to the disclosed technical content without creative labor, and these substitutions and deformations are all within the protection scope of the present invention.

Claims

1. A broadband weak current signal detection system, characterized in that: It includes a signal acquisition motherboard and a host computer system, including: The signal acquisition motherboard is configured with: The signal input interface adopts a triaxial BNC interface and a triaxial cable protection ring driving circuit with a metal noise shielding structure to shield electromagnetic noise interference in the environment; it is used to reduce the leakage current of the transmission cable; Channel switching circuit, used for switching between wide range detection and AC / DC coupling mode; The transimpedance amplifier circuit is used to convert a weak current signal into a measurable voltage signal by using a transimpedance amplifier, and to perform weak current signal conditioning; The analog-to-digital conversion circuit is used to convert the conditioned voltage signal into a digital quantized value using an analog-to-digital converter ADC; The control core board is used to send the analog-to-digital converted voltage signal to the digital signal processing system, and store the data after being processed by the digital noise reduction algorithm; The host computer system is used to communicate with the host computer through a Gigabit Ethernet transmission path, thereby realizing real-time measurement of broadband weak current signals.

2. The broadband weak current signal detection system according to claim 1, characterized in that: The transimpedance amplifier converts the signal, including a feedback current integration type I / F conversion that converts the current signal into a frequency signal for measurement by time integration, and an I / V conversion scheme that converts the current signal into a voltage signal. The voltage signal obtained by the I / V conversion is further converted into a frequency pulse signal for measurement through a voltage-frequency converter.

3. The broadband weak current signal detection system according to claim 2, characterized in that: The feedback current integration type I / F conversion circuit comprises a current integrator, a voltage comparator and a monostable trigger which are connected in sequence, and the monostable trigger is connected to the current integrator through a switch circuit.

4. The broadband weak current signal detection system according to claim 2, characterized in that: The I / V conversion circuit first uses a transimpedance amplifier TIA to convert the weak current signal to be measured into a voltage signal, and then uses an analog-to-digital converter ADC to collect the voltage signal and digitize the signal.

5. The broadband weak current signal detection system according to claim 4, characterized in that: The operational amplifier of the I / V conversion circuit based on TIA has its non-inverting input terminal grounded, its inverting input terminal virtual grounded, and its feedback resistor R F Connect the inverting input and output of the operational amplifier, and the feedback resistor R F Parallel feedback capacitor C F , the total input capacitance C is connected in parallel between the non-inverting input and the inverting input T .

6. The broadband weak current signal detection system according to claim 5, characterized in that: Total input capacitance C T for C T =C S +C DIFF +C CM2 In the formula, C S is the parasitic capacitance of the signal transmission cable, C DIFF is the differential amplifier capacitance, C CM2 is the common-mode amplifier capacitance.

7. The broadband weak current signal detection system according to claim 5, characterized in that: Satisfy the feedback capacitor C F The capacitance and the gain-bandwidth product of the operational amplifier C F Calculation formula: In the formula, f BW is the required closed-loop bandwidth of the TIA circuit, R F is the feedback resistor.

8. The broadband weak current signal detection system according to claim 1, characterized in that: The metal noise shielding structure sets a wrapping area around the weak current signal input path and drives the voltage of this area to the signal input voltage, making the potential difference between the guard ring and the signal line zero.

9. The broadband weak current signal detection system according to claim 1, characterized in that: The three-coaxial cable protection ring driving circuit has two amplifiers with the same-phase input terminals connected and then grounded. The inverting input terminal of the first amplifier is connected to the output terminal and then connected to the inner layer of the coaxial cable. The outer shielding layer of the coaxial cable is grounded. The inner core of the coaxial cable is connected to the inverting input terminal of the second amplifier. The reverse input terminal of the second amplifier is connected to the output terminal in parallel with the feedback resistor R. F .

10. A detection method of a broadband weak current signal detection system according to any one of claims 1 to 9, characterized in that: include: A three-coaxial BNC interface is used as the detection system's broadband weak target detection current signal input interface, and a metal noise shielding structure is used to shield electromagnetic noise interference in the environment; A channel switching circuit is set to switch the detection range and the AC / DC coupling mode; Adopting a feedback current integration type I / F conversion that converts a weak current signal into a frequency signal for measurement by time integration, and an I / V conversion scheme that converts a weak current signal into a voltage signal, and obtaining a measurable voltage signal after I / V conversion; The conditioned voltage signal is converted into a digital quantized value using an analog-to-digital converter (ADC) and sent to a digital signal processing system. After being processed by a digital noise reduction algorithm, the data is stored. Gigabit Ethernet transmission is set up to communicate with the host computer to achieve wideband measurement and high-speed data transmission, thereby performing real-time measurement.

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