Receiver

Abstract

The present disclosure relates to a receiver, comprising: an I-branch including an I-branch mixer that outputs an I component by down-converting a radio frequency (RF) signal, and an I-branch channelization filter that separates a base band signal from an output signal of the I-branch mixer, and an I-branch channel filter that separates the base band signal from the output signal of the I-branch mixer; a Q-branch including a Q-branch mixer that outputs a Q component by down-converting the radio frequency (RF) signal and a Q-branch channelization filter that separates a baseband signal from an output signal of the Q-branch mixer; a frequency-dependent mismatch estimation unit that calculates a frequency-dependent mismatch between the I-branch and the Q-branch; and a frequency-dependent mismatch compensation unit that compensates for the frequency-dependent mismatch between the I-branch and the Q-branch on the basis of the calculation result of the frequency-dependent mismatch estimation unit.

Description

Technical Field

[0001] The present technology generally relates to a receiver. Background Art

[0002] A direct down-conversion receiver uses quadrature mixing to downconvert RF signals to baseband (BB) all at once. Quadrature mixing utilizes two mixers to multiply two local oscillator (LO) signals with a 90-degree phase shift with the received signal. The resulting I and Q signals, separated into baseband I and Q branches, pass through a lowpass filter (LPF) and amplifier, then are converted into discrete-time signal samples by an analog-to-digital converter (ADC) for processing. Summary of the Invention

[0003] Technical problem to be solved by the invention

[0004] An ideal receiver operates under three prerequisites: a 90-degree phase difference between the two local oscillator (LO) signals, identical gain, and identical frequency characteristics of the low-pass filters (LPFs) of the two I / Q branches, as well as the amplifiers and digital-to-analog converters (DACs). However, in actual semiconductor engineering, component mismatches, unrelated to the design, can occur at the component level, preventing these three prerequisites from being met. This results in local oscillator (LO) phase / gain mismatch and baseband filter mismatch.

[0005] The receiver mismatch described above has both frequency-independent and frequency-dependent characteristics, and the IQ mismatch of the receiver described above acts by adding a conjugate image to the original signal to be received, thereby causing the signal-to-noise ratio (SNR) of the received signal to deteriorate and the bit error rate (BER) to increase, just like an interferer.

[0006] One of the problems to be solved by the present technology is to provide a technology that can compensate for the two mismatches that are difficult in the prior art.

[0007] Solutions for solving technical problems

[0008] This embodiment relates to a receiver, comprising: an I branch, including an I-branch mixer that outputs an I component by down-converting a radio frequency (RF) signal, and an I-branch channelization filter that separates a baseband signal from the output signal of the I-branch mixer; a Q branch, including a Q-branch mixer that outputs a Q component by down-converting the RF signal, and a Q-branch channelization filter that separates the baseband signal from the output signal of the Q-branch mixer; a frequency-dependent mismatch estimator that calculates a frequency-dependent mismatch between the I branch and the Q branch; and a frequency-dependent mismatch compensation unit that compensates for the frequency-dependent mismatch between the I branch and the Q branch based on the calculation result of the frequency-dependent mismatch estimator.

[0009] In one aspect of this embodiment, the frequency-dependent mismatch compensator further includes, on the I path, a compensation filter having a transfer function corresponding to a result of dividing a transfer function of the I-branch channelization filter by a transfer function of the Q-branch channelization filter.

[0010] In one aspect of this embodiment, the frequency-dependent mismatch compensating unit further includes a delay line corresponding to the delay of the compensation filter on the Q path.

[0011] In one aspect of this embodiment, the frequency-dependent mismatch estimating unit calculates a transfer function of a compensation filter for compensating for the frequency-dependent mismatch from a value obtained by normalizing the time average of the tone-image correlation of the signal output from the channelized filter by the average power of the signal output from the channelized filter.

[0012] In one aspect of this embodiment, when the value is m(k), the frequency-dependent mismatch estimation unit satisfies the mathematical formula The relationship between the two is obtained by extrapolating the argument and absolute value of Γ(k) to obtain the H D (k), by D (k) Perform a normalized N-point inverse discrete Fourier transform (IDFT) and apply a window to find the impulse response h of the compensation filter. D (n).

[0013] In one aspect of this embodiment, the frequency-dependent mismatch estimation unit uses a pilot signal including single-sideband (SSB) multi-tone to obtain a value obtained by normalizing the time average of the tone-image correlation by the average power of the signal output from the channelization filter.

[0014] This embodiment relates to a receiver, comprising: an I branch, including an I-branch mixer that outputs an I component by down-converting a radio frequency (RF) signal, and an I-branch channelization filter that separates a baseband signal from the output signal of the I-branch mixer; a Q branch, including a Q-branch mixer that outputs a Q component by down-converting the RF signal, and a Q-branch channelization filter that separates the baseband signal from the output signal of the Q-branch mixer; a frequency-independent mismatch estimator that calculates a frequency-independent mismatch between the I branch and the Q branch; and a frequency-independent mismatch compensator that compensates for the frequency-independent mismatch between the I branch and the Q branch based on the calculation result of the frequency-independent mismatch estimator.

[0015] In one aspect of this embodiment, the independent mismatch estimator calculates the gain mismatch and phase mismatch of the I-branch mixer and the Q-branch mixer from a value obtained by normalizing the time average of the power of the output signals of the I-branch mixer and the Q-branch mixer by the time average of the squares of the output signals of the I-branch mixer and the Q-branch mixer.

[0016] In one aspect of this embodiment, the independent mismatch estimation unit is provided by the mathematical formula The gain mismatch (g RX ) and phase mismatch (θ RX ). (x n : The power of the mixer in the time domain)

[0017] In one aspect of this embodiment, the frequency-independent mismatch compensation unit includes: a first multiplier connected to the I branch; a second multiplier connected to the Q branch; a third multiplier for amplifying the output signal of the first multiplier; and an adder for adding the output of the second multiplier and the output of the third multiplier.

[0018] In one aspect of this embodiment, the gain of the first multiplier corresponds to g RX , the gain of the second multiplier corresponds to cos(1 / θ RX), and the gain of the third multiplier corresponds to tan(θ RX ).

[0019] Effects of the Invention

[0020] Through this embodiment, the advantage of being able to compensate for both frequency-dependent mismatch and frequency-independent mismatch of the receiver can be achieved. BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Figure 1 is a schematic diagram illustrating an overview of a practical receiver with mismatch.

[0022] Figure 2 This is a schematic diagram illustrating an outline of a frequency-independent mismatch compensation unit.

[0023] Figure 3 This is a schematic diagram illustrating an outline of a frequency-dependent mismatch compensation unit.

[0024] Description of Reference Numerals

[0025] 10: Receiver

[0026] 100: I branch

[0027] 110: I branch mixer

[0028] 120: I branch channelization filter

[0029] 123: Delay Line

[0030] 200: Q branch

[0031] 210: Q branch mixer

[0032] 220: Q branch channelization filter

[0033] 223: Difference Filter

[0034] 310: Frequency-dependent mismatch estimation unit

[0035] 320: Frequency-dependent mismatch compensation section

[0036] 410: Frequency-independent mismatch estimation unit

[0037] 420: Frequency-independent mismatch compensation unit DETAILED DESCRIPTION

[0038] Next, this embodiment will be described with reference to the accompanying drawings. Figure 1 is a schematic diagram illustrating an overview of a practical receiver 10 with mismatch. Figure 1According to the receiver 10 of this embodiment, it includes: an I branch 100, including an I branch mixer 110 that outputs an I component by down-converting a radio frequency (RF) signal r(t); and a Q branch 200, including a Q branch mixer 210 that outputs a Q component by down-converting the radio frequency (RF) signal r(t).

[0039] In one embodiment, the I branch 100 may further include an I branch channelization filter 120 for separating a baseband signal from the output signal of the I branch mixer 110, and the Q branch 200 may further include a Q branch channelization filter 220 for separating a baseband signal from the output signal of the Q branch mixer 210.

[0040] The receiver 10 may include: a frequency-dependent mismatch estimating unit (FD est) 310 for calculating the frequency-dependent mismatch between the I branch 100 and the Q branch 200; and a frequency-dependent mismatch compensating unit 320 for compensating for the frequency-dependent mismatch between the I branch 100 and the Q branch 200 according to the calculation result of the frequency-dependent mismatch estimating unit 310.

[0041] In addition, in one embodiment of the receiver 10, the receiver 10 may include: a frequency-independent mismatch estimation unit 410, which calculates the frequency-independent mismatch between the I branch 100 and the Q branch 200; and a frequency-independent mismatch compensation unit 420, which compensates for the frequency-independent mismatch between the I branch 100 and the Q branch 200 based on the calculation result of the frequency-independent mismatch estimation unit 410.

[0042] In the illustrated embodiment, an example diagram is provided for a case where the receiver includes both a frequency-dependent mismatch estimation unit 310 and a frequency-dependent mismatch compensation unit 320 as well as a frequency-independent mismatch estimation unit 410 and a frequency-independent mismatch compensation unit 420. However, in an embodiment of the receiver not illustrated, only the frequency-dependent mismatch estimation unit and the frequency-dependent mismatch compensation unit or only one of the frequency-independent mismatch estimation unit and the frequency-independent mismatch compensation unit may be included.

[0043] See Figure 1 The received signal r(t) is input to the I branch 100 and the Q branch 200. The signal r(t) is mixed with the down-converted signal cosω in the mixer 110 of the I branch 100. RX The signal r(t) is down-converted by mixing with the down-conversion signal in the mixer 210 of the Q branch 200.

[0044] The signal x is down-converted in the mixer 110 of the I branch.I (t) is input to the I-branch channelization filter 120, and the I-branch channelization filter 120 outputs a baseband signal y I (t). The impulse response of the I-branch channelization filter 120 in the time domain can be expressed as h I RX (t) The down-converted signal x in the mixer 210 of the Q branch Q (t) is input to the Q branch channelization filter 220. The Q branch channelization filter 220 outputs the baseband signal y Q (t). The impulse response of the Q branch channelization filter 220 in the time domain can be expressed as h Q RX (t).

[0045] As one embodiment, the I-branch channelization filter 120 and the Q-branch channelization filter 220 may be low-pass filters (LPFs). In one embodiment, the outputs of the I-branch channelization filter 120 and the Q-branch channelization filter 220 may be provided to an analog-to-digital converter (ADC). In one embodiment, the I-branch 100 may further include a delay line 123 that is delayed by a corresponding delay time to match the delay of the Q-branch 200. The outputs of the I-branch channelization filter 120 and the Q-branch channelization filter 220 may be provided to an analog-to-digital converter (ADC).

[0046] Ideally, the signals provided by a local oscillator (not shown) to the mixers for down-conversion have the same amplitude. Furthermore, ideally, the signals provided to the I-branch mixer 110 and the Q-branch mixer 210 have a 90-degree phase difference.

[0047] However, in reality, the signal sizes provided by the local oscillator (LO) vary, and the phase difference of the provided signals does not exactly form 90 degrees. In the above case, for the mismatch in signal size, the gain mismatch is expressed as g RX and denote the phase mismatch as θ RX However, the mixer's gain and phase mismatch during down-conversion cannot be distinguished from the local oscillator (LO) mismatch and may not be displayed separately. As will be described later, gain mismatch and phase mismatch are time (and frequency) independent.

[0048] The signal with mismatch L is provided to the mixer. RX (t) can be expressed by the following mathematical formula.

[0049]

Mathematical formula 3

[0050]

[0051] (L RX (t): local oscillator (LO) signal, g RX : gain mismatch, θ RX : Phase mismatch, ω RX : Down-conversion signal frequency)

[0052] The output signal x(t) down-converted using the local oscillator (LO) signal including mismatch can be expressed by the following mathematical formula.

[0053]

Mathematical formula 4

[0054]

[0055] (x(t): output signal of the mixer, r BB (t): received signal converted to baseband, r * BB (t): complex conjugate of the received signal converted to baseband

[0056] Furthermore, by reorganizing Equation 4 in the frequency domain, the following Equation 5 can be obtained.

[0057]

Mathematical formula 5

[0058]

[0059] (α, β: constants)

[0060] In the ideal case without mismatch effect, g RX is 1, and θ RX is 0, so the resulting β value is 0, and because the transmission frequency ω TX and receiving frequency ω RX The same, so the baseband signal r can be obtained BB However, when gain mismatch and phase mismatch occur, the β value is not 0, and it can be confirmed that an unexpected r is generated due to IQ mismatch. BB (t) The conjugate image of the signal is r * BB (t) Ingredients.

[0061] The conjugate image generated as described above is r * BB The (t) component will interfere with the target signal, thus degrading the signal-to-noise ratio and reception performance of the receiver. In addition, as shown in Mathematical Formula 5, from the perspective of the frequency domain, the target signal rBB The coefficients of (t) and its conjugate image, α and β, are not functions of frequency but are constants. In this sense, they can be regarded as frequency-independent IQ mismatch.

[0062] The filters h of the baseband I branch 100 and Q branch 200 RX I (t) and h RX Q The mismatch of (t) includes the pole and zero differences caused by the circuit elements in the semiconductor engineering of the anti-aliasing low-pass filter of the analog baseband and the propagation delay differences until the analog-to-digital converter (ADC) samples and holds.

[0063] When the baseband filter has IQ mismatch, the input and output of the filter stage satisfy the following mathematical formula 6.

[0064]

Mathematical formula 6

[0065]

[0066] (x i (t): Output signal of the I-branch mixer, x Q (t): Output signal of the Q-branch mixer, h i RX (t): impulse response of the I branch channelization filter, hQRX(t): impulse response of the Q branch channelization filter, Convolution operation)

[0067] In equation 6, it can be confirmed that the target signal, x(t), is not transmitted directly, but is attached with a conjugate image, x * (t). From the perspective of frequency domain, the coefficient H of the target signal and the conjugate image e RX (f) and H o RX (f) is a function of frequency, and the interference caused by the image is also a function of frequency. From this perspective, it can be called frequency-dependent IQ mismatch.

[0068] Therefore, when the IQ mismatch of the local oscillator (LO), the mixer, and the baseband branch is determined, the demodulated signal y(t) can be expressed as shown in the following equation 7. Due to the mismatch of the I branch 100 and the Q branch 200 at the receiving end 10, a complex conjugate term as shown in equation 6 is formed in the output signal, thereby forming an image in the frequency domain.

[0069]

Mathematical formula 7

[0070]

[0071] In Mathematical Formula 7, when H I RX (f) can be expressed as H when extracted as a common factor D RX (f) = H Q RX (f) / H I RX (f), which is the ratio of the transfer function of the Q branch channelized filter to the transfer function of the I branch channelized filter. Therefore, the channelized filter 220 of the Q branch is as follows: Figure 1 As shown, in the time domain, it can be represented as an I-branch channelization filter 120 and a filter (h D RX (t)) 223 cascade. The mathematical formula 7 can be organized as follows.

[0072]

Mathematical formula 8

[0073]

[0074] You can consider H I RX (f) as a common factor of the I branch and the Q branch, and including the frequency-independent mismatch compensation unit 410 (including the H D RX (f) = H Q RX (f) / H I RX (f) Frequency dependence of the transfer function of the mismatch compensation filter) of the system model. This can be achieved using the discrete time domain h D RX (n) Figure 2 The method shown is shown. D RX (t) can be organized as the following mathematical formula 9, which can express the mismatch between the baseband filters of the IQ branches.

[0075]

Mathematical formula 9

[0076]

[0077] The mismatch between the IQ branch baseband filters is expressed as H of the Q branch. D RX (f), and in order to compensate for it, H is added to the I branch D RX(f) To achieve this, after performing analog-to-digital (AD) sampling, by setting h in the I branch D RX (t) is the discrete-time version of h D RX (n) and compensate. Because h D RX (n) is an infinite impulse response (IIR), so for practical implementation, a finite impulse response (FIR) approximation is used. In this case, the Q branch includes a tapped-delay-line 325 and is connected to h D RX The integer set delay D of (n) is kept balanced. Thus, the reception frequency-dependent IQ mismatch can be compensated.

[0078] As shown in Equation 10, the conversion equation for the complex envelope of the gain mismatch and phase mismatch induced by Equations 4 and 5 can be obtained, and further, as shown in Equation 11, the conversion equation for the matrix form of the IQ vector can be obtained. (z(t) = rBB(t))

[0079]

Mathematical formula 10

[0080]

[0081]

Mathematical formula 11

[0082]

[0083] By arranging Equation 10 into a matrix and finding its inverse matrix, the following Equation 11 can be obtained, which can compensate for the gain mismatch and phase mismatch of the local oscillator (LO) on the receiving side.

[0084] Figure 3 This is a schematic diagram illustrating the outline of the frequency-independent mismatch compensation unit 420. Figure 3 The frequency-independent mismatch compensation unit 410 includes: a first multiplier 421, connected to the I branch; a second multiplier 422, connected to the Q branch; a third multiplier 423, multiplying the output signal of the first multiplier; and an adder 424, adding the output of the second multiplier 422 and the output of the third multiplier 423.

[0085] Figure 3 The frequency-independent mismatch compensation unit 410 shown in FIG. 1 realizes the equation 11. The gain of the first multiplier 421 included in the frequency-independent mismatch compensation unit 420 corresponds to g RXThe gain of the second multiplier 422 corresponds to cos(1 / θ RX As an embodiment, Taylor series is used to convert cos(1 / θ RX ) is approximately 1+(θ RX 2 ) / 2, and use it as the gain of the second multiplier 422.

[0086] The gain of the third multiplier corresponds to tan(θ RX As an embodiment, tan(θ RX ) is approximately θ RX +(θ RX 2 ) / 3, and use it as the gain of the third multiplier 423. By compensating for the independent mismatch in the manner described above, a mismatch-compensated received signal r can be obtained. I and r Q .

[0087] In order to compensate for the frequency-independent IQ mismatch, it is necessary to estimate the IQ gain mismatch of the local oscillator (LO) and the mixer circuit, i.e., g RX and IQ phase mismatch, i.e., θ RX In the above-mentioned mathematical formula 4, when it is assumed that the N-point normalized discrete Fourier transform (DFT) of the discrete-time signal x(n) sampled with the signal x(t) is X(k) (-N / 2≤k≤N / 2-1), the k-th frequency component X(k) includes the conjugate image, i.e., the complex conjugate image R(k) of the -k-th frequency component, in addition to the k-th frequency component R(k) of the original signal. * (-k).

[0088] Because X(k)=αR(k)+βR * (-k), X(-k) = αR(-k) + βR * (k), therefore, the time average of the product of X(k) and X(-k) can be obtained as the following mathematical formula 12.

[0089]

Mathematical formula 12

[0090] E[X(k)X(-k)]=α 2 E[R(k)R(-k)]+β 2 E[R * (-k)R * (k)]+αβE[R(k)R * (k)]+αβE[R(-k)R * (-k)]

[0091] In equation 12, for thermal noise and general signals, R(k) and R(−k) are uncorrelated and E[R(k)]=0. Therefore, equation 12 can be transformed into equation 13.

[0092]

Mathematical formula 13

[0093]

[0094] The relationship between the expected values ​​of signal power before and after being affected by IQ mismatch is expressed in the following equation 14.

[0095]

Mathematical formula 14

[0096] E[|X(k)| 2 ]=E[X(k)X * (k)] = E[{αR(k) + βR * (-k)}{α * R * (k)+β * R(-k)}]

[0097] (αα * +ββ * )E[|R(k)| 2 ]

[0098] By using the above two formulas, we can obtain the following equation 15 for estimating the IQ signal gain mismatch and phase mismatch of the receiver local oscillator (LO) and mixer, i.e., g RX ,θ RX indicators.

[0099]

Mathematical formula 15

[0100]

[0101] That is, the gain mismatch component of the mixer is g RX and the phase mismatch component, θ RX The time average E[xn 2 ] is the time average of the power of the mixer output signal E[|xn| 2 The estimation index can be applied to any received signal r without any special conditions. BB (t), and because no separate pilot signal is required, it can be called blind estimation.

[0102] When X(k) is an ergodic process, the time average E[X(k)X(-k)] should be the same as the ensemble average. Furthermore, as shown in Equation 16 below, the ensemble average of X(k)X(-k) is the same as the time average, indicating that X(k) is an ergodic process. Therefore, the ensemble average can be used as an estimation metric for the time average.

[0103]

Mathematical formula 16

[0104]

[0105] The estimation index is the system average of the audio-visual correlation. However, the system average of the audio-visual correlation can be organized into the following mathematical formula.

[0106]

Mathematical formula 17

[0107]

[0108] The average of the signal products provides an estimation metric as shown in the following mathematical formula 18.

[0109]

Mathematical formula 18

[0110]

[0111] The signal power before and after being contaminated due to IQ mismatch satisfies the following mathematical formula 19.

[0112]

Mathematical formula 19

[0113]

[0114] Through the two formulas, only the time domain signal x that is contaminated due to IQ mismatch can be used. n , to obtain the value used to estimate g RX ,θ RX (Indicator for estimating IQ gain mismatch and phase mismatch).

[0115]

Mathematical formula 20

[0116]

[0117] As mentioned above, in order to compensate for the frequency-dependent IQ mismatch, it is necessary to estimate h D RX The discrete signal version of (t) is h D RX (n). In the channelization filter, h is estimated from the output signal y(t) after being affected by the conjugate image. D RX(n). Assuming that the discrete signal samples of signal y(t) are y(n) and the N-point normalized discrete Fourier transform (DFT) of y(n) is Y(k), Y(k) satisfies the relationship of Mathematical Formula 21 according to the mathematical formula of the continuous-time version as described above.

[0118]

Mathematical formula 21

[0119] Y(k)=G1 RX (k)R BB (k)+G2 RX (k)R BB * (-k)

[0120] Y(-k)=G1 RX (-k)R BB (-k)+G2 RX (-k)R BB * (k)

[0121] When single-sideband (SSB) multi-tones filling the positive half of the frequency bin of the N-point discrete Fourier transform (DFT) are input as pilot signals, R BB (-k)=R BB * (-k)=0, and the formula can be organized into mathematical formula 22.

[0122]

Mathematical formula 22

[0123]

[0124] The result of normalizing the time average of the audio-visual correlation Y(k)Y(-k) by the average power of Y(k) is defined as the following m(k).

[0125]

Mathematical formula 23

[0126]

[0127] In Equation 23, because h D (n) is a real filter, so H is sorted by applying conjugate symmetry. D (-k)=H D * (k), H can be calculated as shown in the following mathematical formula 24 D (k)g RX e -jθRX .

[0128]

Mathematical formula 24

[0129]

[0130]

[0131] Since positive single-sideband (SSB) multitones are input, the calculation of m(k) is only valid in the interval 1≤k≤(N / 2-1) (E[|Y(k)| 2 ]=0for-N / 2≤k≤0). However, according to the conjugate symmetry arrangement H D (-k)=H D * (k), because H D The argument (arg) of (0) is 0, so θ RX = -arg{Γ(0)}. However, since Γ(0) cannot be calculated directly, it is estimated by extrapolating arg{Γ(k)}.

[0132] In addition, the local oscillator (LO) gain mismatch g RX Can be included in H D (k) does not need to be calculated separately, but because of the low-pass filter mismatch |H D (0)|=1, so we can use g RX =|Γ(0)|. Similarly, estimate by extrapolating |Γ(k)|. In the interval -N / 2≤k≤0, H D (k) Filling can be performed using conjugate symmetry arrangement.

[0133] As mentioned above, since H is estimated for all intervals -N / 2≤k≤N / 2-1 D (k), the compensated finite impulse response filter (FIR filter) hD(n) can be obtained by calculating the N-point normalized inverse discrete Fourier transform (IDFT) and applying an appropriate window, and the result of expressing it as a mathematical formula is shown in the following mathematical formula 24.

[0134]

Mathematical formula 24

[0135] h D (n) = NIDFT{H D (k)}·w(n)

[0136] That is, by calculating the ratio of the time average of the correlation operation between the signal and the image to the average power of the signal at each discrete frequency position, it is possible to grasp the mismatch and phase mismatch in the filter at each frequency position.

Claims

1. A receiver, comprising: An I branch includes an I branch mixer that down-converts a radio frequency (RF) signal to output an I component, and an I branch channelization filter that separates a baseband signal from the output signal of the I branch mixer. a Q branch including a Q branch mixer that outputs a Q component by down-converting the radio frequency (RF) signal and a Q branch channelization filter that separates a baseband signal from an output signal of the Q branch mixer; a frequency-dependent mismatch estimating unit, configured to calculate a frequency-dependent mismatch between the I branch and the Q branch; as well as, The frequency-dependent mismatch compensating unit compensates for the frequency-dependent mismatch between the I branch and the Q branch according to the calculation result of the frequency-dependent mismatch estimating unit.

2. The receiver according to claim 1, the frequency-dependent mismatch compensation section, Also included on the I path is a compensation filter having a transfer function corresponding to a result of dividing the transfer function of the I-branch channelization filter by the transfer function of the Q-branch channelization filter.

3. The receiver according to claim 2, the frequency-dependent mismatch compensation section, The Q path further includes a delay line corresponding to the delay of the compensation filter.

4. The receiver according to claim 1, the frequency-dependent mismatch estimation section, A transfer function of a compensation filter for compensating for frequency-dependent mismatch is obtained from a value obtained by normalizing a time average of tone-image correlation of a signal output from the channelization filter by an average power of the signal output from the channelization filter.

5. The receiver according to claim 4, When the value is m(k), Frequency dependent mismatch estimation unit, Satisfy the mathematical formula relationship, The H is obtained by extrapolating the argument and absolute value of Γ(k) D (k), The impulse response h of the compensation filter is obtained by performing a normalized N-point inverse discrete Fourier transform (IDFT) on the HD(k) and applying a window D (n).

6. The receiver according to claim 4, the frequency-dependent mismatch estimation section, A value obtained by normalizing a time average of tone-image correlation by an average power of a signal output from the channelization filter using a pilot signal including single sideband (SSB) multi-tone is obtained.

7. A receiver comprising: An I branch includes an I branch mixer that down-converts a radio frequency (RF) signal to output an I component, and an I branch channelization filter that separates a baseband signal from the output signal of the I branch mixer. a Q branch including a Q branch mixer that outputs a Q component by down-converting the radio frequency (RF) signal and a Q branch channelization filter that separates a baseband signal from an output signal of the Q branch mixer; a frequency-independent mismatch estimating unit, configured to calculate a frequency-independent mismatch between the I branch and the Q branch; as well as, The frequency-independent mismatch compensating unit compensates for the frequency-independent mismatch between the I branch and the Q branch according to the calculation result of the frequency-independent mismatch estimating unit.

8. The receiver according to claim 7, the independent mismatch estimation unit, The gain mismatch and phase mismatch of the I-branch mixer and the Q-branch mixer are calculated from a value obtained by normalizing a time average of the power of the output signals of the I-branch mixer and the Q-branch mixer to a time average of the squares of the output signals of the I-branch mixer and the Q-branch mixer.

9. The receiver according to claim 7, the independent mismatch estimation unit, Through mathematical formula Calculation, Estimated gain mismatch (g RX ) and phase mismatch (θ RX ): X(k): output of the mixer in the discrete time domain, R(k): input of the mixer in the discrete time domain.

10. The receiver according to claim 9, The frequency-independent mismatch compensation unit includes: a first multiplier connected to the I branch; a second multiplier connected to the Q branch; a third multiplier, configured to amplify an output signal of the first multiplier; as well as, An adder adds the output of the second multiplier and the output of the third multiplier.

11. The receiver according to claim 10, The gain of the first evaporator corresponds to g RX , The gain of the second multiplier corresponds to cos(1 / θ RX ), The gain of the third multiplier corresponds to tan(θ RX ).

Images