Drive circuit of power transistor, load switch circuit, and power supply module

Abstract

Disclosed in the present invention are a drive circuit of a power transistor, a load switch circuit, and a power supply module. The drive circuit comprises: a critical saturation drive voltage generation module, used for generating a critical saturation drive voltage; a buffer module, used for generating, on the basis of the critical saturation drive voltage, a drive signal applied to a control end of the power transistor, so as to cause the power transistor to work in a critical saturation on state between a saturation on state and a current limiting state; and an over-current protection module, used for controlling the power transistor to be off when over-current occurs in the power transistor. When short circuit occurs at a voltage output end, the power transistor, due to being in the critical saturation on state, first enters the current limiting state to limit the change rate of the current, preventing excessive switching stress from being generated during a subsequent protective disconnection operation, thereby preventing generation of voltage overshoot in the power transistor.

Description

Power transistor drive circuit, load switch circuit and power supply module

[0001] This application claims priority to a Chinese invention application filed on October 19, 2023, with application number 202311352706X and titled “Power transistor drive circuit, load switch circuit and power supply module”, and incorporates the entire specification, claims, drawings and abstract of the above-mentioned Chinese invention application into this application by reference. Technical Field

[0002] The present invention relates to the technical field of power transistor driving, and more particularly to a power transistor driving circuit, a load switch circuit, and a power supply module. Background Art

[0003] With the continuous development of high-power electronic devices, the demand for reliable driving circuits under high current conditions is gradually increasing. Especially in fields such as electric vehicles and power management systems, the demand for efficient and reliable short-circuit protection technology is becoming increasingly urgent.

[0004] Existing high-power electronic devices typically use MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors, or simply power transistors) to control current flow. When using MOSFETs as switches connecting power sources to loads, it is generally desirable for the MOSFET to have as low an on-resistance as possible to ensure efficient circuit operation. However, if the load shorts, a large current will flow through the power transistor if the MOSFET's on-resistance is too low, increasing stress on the circuit and device.

[0005] In addition, there is a new problem of how to disconnect the MOSFET when the load is short-circuited. If the disconnection speed is too slow, a large current will continue to flow for a relatively long time, increasing the stress on the MOSFET and the load, and may even cause damage to the MOSFET. However, if the disconnection speed is too fast, the rapid rate of change of current and the parasitic inductance in the current path will generate a kickback voltage in the circuit, which will have an adverse effect on the MOSFET and the load.

[0006] Therefore, current MOSFET switch technology presents some challenges in short-circuit protection and control during disconnection. In high-power applications and high-frequency operation, a new solution is needed that can ensure efficient conduction, provide effective protection in short-circuit conditions, and control the current change rate during disconnection to reduce the generation of kickback voltage. Summary of the Invention

[0007] In view of the above problems, the object of the present invention is to provide a driving circuit, a load switch circuit and a power supply module for a power transistor. The driving circuit can not only provide effective protection for the power transistor in the event of overcurrent, but also control the current change rate when the power transistor is disconnected to reduce the generation of kickback voltage.

[0008] According to one aspect of the present invention, a drive circuit for a power transistor is provided, wherein the power transistor is connected between a voltage input terminal and a voltage output terminal, and the drive circuit comprises: a critical saturation drive voltage generating module for generating a critical saturation drive voltage; a buffer module for outputting a drive signal applied to a control terminal of the power transistor based on the critical saturation drive voltage, so that the power transistor is in a critical saturation conduction state, wherein in the critical saturation conduction state, when the current of the power transistor is less than a rated current, the power transistor operates in a saturation conduction state, and when the current of the power transistor is greater than the rated current, the power transistor operates in a current limiting state due to the limitation of the drive signal; and an overcurrent protection module for detecting the current of the power transistor to determine whether an overcurrent occurs in the power transistor, and controlling the power transistor to be turned off when an overcurrent is detected in the power transistor; wherein, when a short circuit occurs at the voltage output terminal, the power transistor first enters a current limiting state due to being in the critical saturation conduction state, and then the overcurrent protection module turns the power transistor off.

[0009] Optionally, the critical saturation drive voltage generating module includes: a first transistor, whose first end is connected to the voltage input end; a state control module, wherein the state control module is configured to control the voltage between the first end and the second end of the first transistor to be a preset voltage value, and at the same time control the current of the first transistor to be a preset current value, so that the first transistor is in a critical saturation conduction state; wherein the voltage on the control end of the first transistor is the critical saturation drive voltage.

[0010] Optionally, the state control module includes: a constant current source connected to the second end of the first transistor; an operational amplifier, whose positive input end is connected to a reference voltage source, whose negative input end is connected to the common connection node of the first transistor and the constant current source, and whose output end is connected to the control end of the first transistor, wherein the operational amplifier regulates the first transistor through negative feedback to obtain the critical saturation drive voltage.

[0011] Optionally, the overcurrent protection module includes: a second transistor and a controlled current source connected in series between the voltage input terminal and ground, the control terminal of the second transistor being connected to the drive signal; a sampling current acquisition unit, configured to adjust the current in the controlled current source so that the second terminal voltage of the second transistor is equal to the second terminal voltage of the power transistor, and obtain a sampling current based on the current of the controlled current source at this time, wherein the sampling current has a second current ratio with the current in the power transistor; a voltage comparison unit, configured to obtain a voltage difference between the input voltage of the voltage input terminal and the output voltage of the voltage output terminal; and a pull-down control unit, whose input terminal is connected to the outputs of the sampling current acquisition unit and the voltage comparison unit, and whose output is connected to the control terminal of the power transistor, and configured to determine whether to pull the control terminal of the power transistor down to ground based on the sampling current acquisition unit and the voltage comparison unit.

[0012] Optionally, the power transistor includes a plurality of third transistors connected in parallel between the voltage input terminal and the voltage output terminal, the control terminals of the plurality of third transistors are connected to the drive signal, and the first transistor and the second transistor have the same transistor size as the third transistors.

[0013] Optionally, the first transistor, the second transistor and the power transistor are formed on the same semiconductor die.

[0014] Optionally, the driving circuit further includes: a coupling capacitor connected between the control terminal and the second terminal of the power transistor.

[0015] Optionally, the buffer module includes a driver, the input end of the driver is used to receive a switch control signal, the voltage bias end of the high potential side or low potential side of the driver is used to receive the critical saturation drive voltage, and the output end of the driver is used to output the drive signal.

[0016] Optionally, the buffer module includes an isolation resistor, a first end of the isolation resistor is connected to the critical saturation driving voltage, and a second end of the isolation resistor is used to output the driving signal.

[0017] According to another aspect of the present invention, a load switch circuit is provided, comprising the above-mentioned power transistor and a driving circuit for driving the power transistor.

[0018] According to another aspect of the present invention, a power supply module is provided, comprising the above-mentioned power transistor and a driving circuit for driving the power transistor.

[0019] Optionally, the power supply module includes a switching power converter or a low voltage dropout linear regulator.

[0020] In summary, the present invention provides a driving circuit for a power transistor, the driving circuit comprising a critical saturation driving voltage generating module and an overcurrent protection module, wherein the critical saturation driving voltage generating module extracts a portion of the transistor branches from the plurality of transistor branches in the power transistor, and uses a negative feedback regulation method to make these portion of the transistor branches behave as a constant current source, so that the voltage driving these portion of the transistor branches is the critical saturation driving voltage, and then the critical saturation driving voltage is used to make the plurality of transistor branches in the power transistor be in a critical saturation conduction state. If the current in the power transistor is less than the rated current, the power transistor is in a saturated conduction state, with a small on-resistance and low loss, and can provide a strong current transmission capability. Once the current in the power transistor exceeds its rated current due to a short circuit, since the MOSFET structure is an intrinsic characteristic of the transconductance device, the power transistor naturally enters a current limiting state, and then when the protective disconnection action is performed by the overcurrent protection module, no excessive switching stress will occur. Therefore, the driving circuit of the present invention can not only provide effective protection in the event of a short circuit, but also control the current change rate when disconnecting to reduce the generation of a kickback voltage. BRIEF DESCRIPTION OF THE DRAWINGS

[0021] The above and other objects, features and advantages of the present invention will become more apparent through the following description of the embodiments of the present invention with reference to the accompanying drawings, in which:

[0022] FIG1 shows a schematic structural diagram of a power transistor driving circuit according to a first embodiment of the present invention.

[0023] FIG. 2 a shows a schematic structural diagram of a buffer module of a driving circuit according to a first embodiment of the present invention.

[0024] FIG. 2 b shows another schematic structural diagram of the buffer module of the driving circuit according to the first embodiment of the present invention.

[0025] FIG3 shows a schematic structural diagram of an overcurrent protection module of a driving circuit according to a first embodiment of the present invention.

[0026] FIG4 shows a schematic circuit diagram of a power module according to a second embodiment of the present invention.

[0027] FIG5 shows a schematic circuit diagram of another power module according to a third embodiment of the present invention.

[0028] FIG6 shows a schematic circuit diagram of yet another power module according to a fourth embodiment of the present invention. DETAILED DESCRIPTION

[0029] Reference will now be made in detail to the exemplary embodiments of the present disclosure, examples of which are illustrated in the accompanying drawings.Where possible, the same reference numerals will be used throughout the drawings to refer to the same or like parts.

[0030] In the specification, it should be noted that similar reference numerals that have been used to represent similar components in other drawings are used for these elements as much as possible. In the following description, when functions and configurations known to those skilled in the art are not related to the basic configuration of the present disclosure, their detailed description will be omitted. The terms described in the specification should be understood as follows.

[0031] The advantages and features of the present disclosure and their implementation methods will be described through the following embodiments described with reference to the accompanying drawings. However, the present disclosure can be implemented in different forms and should not be construed as limited to the embodiments described herein. Rather, these embodiments are provided to make the present disclosure comprehensive and complete, so as to fully convey the scope of the present disclosure to those skilled in the art. Furthermore, the present disclosure is limited only by the scope of the claims.

[0032] The shapes, sizes, ratios, angles, and quantities disclosed in the drawings used to describe the embodiments of the present disclosure are merely examples, and the present disclosure is therefore not limited to the details illustrated. Like reference numerals denote like elements throughout. In the following description, when it is determined that a detailed description of related known functions or structures would inevitably obscure the key points of the present disclosure, the detailed description will be omitted.

[0033] In case the terms “including,” “having,” and “comprising” described in this specification are used, another part may be added unless “only to” is used. Terms in the singular form may include plural forms unless otherwise specified.

[0034] When describing a positional relationship, for example, when the positional relationship between two parts is described as, for example, "on," "over," "below," or "next to," one or more other parts may be provided between the two parts unless "only" or "directly" is used. When describing a temporal relationship, for example, when the temporal sequence is described as "after," "following," "next to," or "before," discontinuous cases may be included unless "just" or "directly" is used.

[0035] It should be understood that although the terms "first," "second," etc. may be used herein to describe various components, these components should not be limited by these terms. These terms are only used to distinguish one component from another. For example, a first component can be referred to as a second component, and similarly, a second component can be referred to as a first component without departing from the scope of this disclosure.

[0036] The term "at least one" should be understood to include any and all combinations of one or more of the corresponding listed items. For example, "at least one of a first item, a second item, and a third item" means all combinations of two or more of the first item, the second item, and the third item, as well as the first item, the second item, or the third item.

[0037] As will be fully understood by those skilled in the art, the features of the various embodiments of the present disclosure may be combined or combined in part or in whole, and may interoperate and technically drive each other in various ways. The embodiments of the present disclosure may be performed independently of each other, or may be performed together in a mutually dependent relationship.

[0038] Hereinafter, exemplary embodiments of the present disclosure will be described in detail with reference to the accompanying drawings.

[0039] FIG1 is a schematic circuit diagram of a power transistor drive circuit according to a first embodiment of the present invention. As shown in FIG1 , the circuit of this embodiment includes a power transistor 101 and a drive circuit 100 thereof. The power transistor 101 is connected to a voltage input terminal Vin and a voltage output terminal Vout, and is used to control the current transmission between the voltage input terminal and the voltage output terminal. The drive circuit 100 is connected to the power transistor 101 and is used to drive the power transistor 101. When a load at the voltage output terminal is short-circuited, the power transistor 101 is promptly turned off to prevent the power transistor 101 from being damaged by high current.

[0040] Furthermore, power transistor 101 may be a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor). To improve the current transfer capability of power transistor 101 and reduce its on-resistance, thereby obtaining a device capable of handling high currents and enabling high-power applications, this embodiment also includes implementing power transistor 101 using multiple low-power MOSFETs (hereinafter referred to as transistors) connected in parallel and / or cascade. Specifically, as shown in FIG1 , power transistor 101 may include multiple transistor branches connected in parallel, such as transistor branch M31, transistor branch M32, transistor branch M33, and transistor branch M3n, where n is an integer greater than 1. Each transistor branch may include one or more low-power transistors. Connecting the sources and drains of the multiple transistor branches together can enhance the current handling capability of the overall device. When each transistor branch includes a low-power transistor, the source and drain of the low-power transistor serve as the source and drain of the corresponding transistor branch, respectively. When each transistor branch includes multiple low-power transistors, the multiple low-power transistors are connected in a cascade structure, and the source or drain of the first low-power transistor in the cascade structure is used as the source or drain of the entire transistor branch, and the drain or source of the last low-power transistor in the cascade structure is used as the drain or source of the entire transistor branch.

[0041] To ensure efficient conduction and provide effective protection in the event of a short circuit, while also controlling the current change rate during disconnection to reduce the generation of a kickback voltage, this embodiment proposes a critical saturation drive scheme. The drive circuit 100 operates the power transistor 101 in a critical saturation conduction state, i.e., the drive voltage of the power transistor 101 is just sufficient to pass its rated current. If the current in the power transistor 101 is less than the rated current, the power transistor 101 is in a saturated conduction state, with low on-resistance and low loss, thus providing strong current transmission capability. Once the current in the power transistor 101 exceeds its rated current, the power transistor 101 naturally enters a current limiting state due to the intrinsic characteristics of the MOSFET structure as a transconductance device. Subsequently, when the protective disconnection action is performed, excessive switching stress will not occur.

[0042] Saturation conduction refers to full conduction. When used as a switch, a MOSFET is designed to conduct fully when turned on, exhibiting a low on-resistance. Because it's unclear at what gate drive voltage the MOSFET is fully conducting, the gate drive voltage is often very high. For high-current switches, a high gate voltage can cause a significant short-circuit surge current to flow through the MOSFET in the event of a short circuit. This can cause the current density to exceed the tolerance of the connected system, potentially damaging the switch. Alternatively, during protective short-circuit current disconnection, the excessive current and the high rate of change during disconnection can lead to excessive surge voltage in the switch, potentially causing breakdown and burnout. Critical saturation conduction attempts to increase the gate voltage to a level just high enough to conduct the rated current. If a short circuit occurs, the gate voltage is just high enough to conduct the rated current. If the short-circuit current exceeds the rated current, the MOSFET enters current limiting due to insufficient drive voltage. When the current in the MOSFET is below the rated current, the gate drive voltage is sufficiently high, causing the MOSFET to exhibit oversaturation conduction, passing current with a low on-resistance. Both critical saturation conduction and oversaturation conduction are defined for specific currents. Because saturation levels vary due to process and temperature variations, the critical saturation drive voltage varies for each device. A branch fabricated simultaneously with the main switch and distributed within the main switch branch requires a drive voltage close to the voltage required to pass the corresponding current through the remaining branches. Driving the main switch at this voltage, the main switch branch can only output the specified current at most. This voltage is the saturation drive voltage corresponding to this specified current. The critical saturation on-current of the main switch is determined by the ratio of the number of switches in the branch to the number of remaining branches. If the branch ratio is 10 million, then when the specified current of the branch is 1μA, the saturation on-current of the main switch is 10 million × 1μA, or 10A. This current limit is determined by the gate drive voltage and does not require external circuitry. In the event of a short circuit, this mechanism primarily limits the current surge, allowing other protection mechanisms time to react.

[0043] To achieve critical saturation drive, the driving circuit 100 of this embodiment further includes a critical saturation drive voltage generating module 110, an overcurrent protection module 120, and a buffer module 130. The critical saturation drive voltage generating module 110 is configured to generate a critical saturation drive voltage Vcb, and the buffer module 130 is configured to generate a drive signal Vg applied to the control terminal (e.g., gate) of the power transistor 101 based on the critical saturation drive voltage Vcb, so that the power transistor 101 operates in the critical saturation conduction state.

[0044] Furthermore, the critical saturation drive voltage generating module 110 includes a transistor M1 and a state control module. The state control module is configured to control the voltage between the first terminal (e.g., the drain) and the second terminal (e.g., the source) of the transistor M1 to be a preset voltage value, and simultaneously control the current of the transistor M1 to be a preset current value, so that the transistor M1 is in a critical saturation conduction state, wherein the voltage on the control terminal (e.g., the gate) of the transistor M1 is the critical saturation drive voltage Vcb.

[0045] For example, the state control module includes an operational amplifier 111, a constant current source 112, and a reference voltage source 113. A first terminal of the constant current source 112 is connected to the second terminal (e.g., source) of the transistor M1, a first terminal (e.g., drain) of the transistor M1 is connected to the voltage input terminal Vin, and a second terminal of the constant current source 112 is connected to a reference ground. A positive input terminal of the operational amplifier 111 is connected to the second terminal of the reference voltage source 113, a first terminal of the reference voltage source 113 is connected to the voltage input terminal Vin, and the reference voltage source 113 is used to provide a reference voltage. A negative input terminal of the operational amplifier 111 is connected to a common connection node between the transistor M1 and the constant current source 112. An output terminal of the operational amplifier 111 is connected to a control terminal (e.g., gate) of the transistor M1. The output voltage of the operational amplifier 111 drives the gate of the transistor M1, so that the voltage difference between the source and drain of the transistor M1 is regulated by negative feedback and stabilized at the reference voltage provided by the reference voltage source 113. At this point, by matching the size of the constant current source 112 with that of the transistor M1, the gate voltage of the transistor M1 can be raised to a level just high enough to achieve saturation conduction and pass the rated current. This means that the transistor M1 is in a critical saturation conduction state. When the transistor M1 is in the constant current source state, the output voltage of the operational amplifier 111 is the critical saturation drive voltage Vcb.

[0046] Furthermore, the transistor M1 has the same size and characteristics as the multiple transistor branches M31, M32, M33, and M3n in the power transistor 101. For example, the transistor M1 and the power transistor 101 are fabricated on the same wafer. A portion of the transistor branches is selected from a MOSFET semiconductor die (also known as a die or chip) obtained from the wafer through a series of processing steps to serve as the transistor M1 (for example, the ratio of the number of transistor M1 to the number of transistor branches of the power transistor 101 is 1 / 100,000; of course, another number of branches may also be selected to serve as the transistor M1). A preset and known sampling resistor is then added to the transistor M1. Negative feedback regulation is used to make the transistor M1 behave as a constant current source. The gate drive voltage of the transistor M1 is then equal to the critical saturation drive voltage Vcb required to generate this constant current. As mentioned above, these transistor branches on the same wafer are manufactured in the same batch using the same process and have the same transistor size and characteristics. Therefore, when the critical saturation drive voltage Vcb is applied to the multiple transistor branches of the power transistor 101, these transistor branches of the power transistor 101 will also exhibit a constant current state under the drive voltage.

[0047] Taking the ratio of the number of branches of transistor M1 to that of power transistor 101 (i.e., the transistor size ratio) as 1 / 100,000 as an example, when the critical saturation drive voltage Vcb can produce a rated current of 100μA in transistor M1, then when this critical saturation drive voltage Vcb is applied to all transistor branches of power transistor 101, a rated current of 100μA × 100,000 = 10A can be obtained in the entire power transistor 101. When the current flowing through power transistor 101 is less than 10A, because the current limit has not been reached, the multiple transistor branches in power transistor 101 are all in a saturated conduction state. When the current flowing through power transistor 101 is greater than 10A, for example, when a short circuit occurs at the voltage output terminal, the multiple transistor branches in power transistor 101 enter a current limiting state, limiting the current in power transistor 101 to the rated current of 10A.

[0048] The overcurrent protection module 120 is used to detect the current in the power transistor 101 and turn off the power transistor 101 when the current in the power transistor 101 is equal to the rated current (that is, the power transistor 101 is operating in a current limiting state at this time) and the voltage difference between the input voltage Vin and the output voltage Vout is greater than a protection threshold, so as to prevent the power transistor 101 from being damaged due to a large current flowing through it for a long time when the load at the voltage output end is short-circuited.

[0049] FIG2a shows a schematic structural diagram of a buffer module of a driving circuit according to a first embodiment of the present invention. In an exemplary embodiment, the power transistor 101 and the driving circuit 100 for driving the power transistor 101 can be applied in a static power switch scenario, in which case the buffer module 130 can be implemented by an isolation resistor R1. As shown in FIG2a, the first end of the isolation resistor R1 is connected to the critical saturation drive voltage Vcb, and the second end of the isolation resistor R1 is connected to the control end of the power transistor 101 to provide the drive signal Vg. The isolation resistor R1 can achieve isolation between the gate of the power transistor 101 and the gate of the transistor M1 in the critical saturation drive voltage generating module 110, thereby preventing the transistor M1 in the critical saturation drive voltage generating module 110 from being affected and changing state when the overcurrent protection module 120 pulls down the gate voltage of the power transistor 101 due to a short circuit.

[0050] Figure 2b shows another schematic structural diagram of a buffer module of a driver circuit according to the first embodiment of the present invention. In another exemplary embodiment, the power transistor 101 and the driver circuit 100 driving the power transistor 101 can be used in a high-speed dynamic power switching scenario. In this case, the buffer module 130 can be implemented by a driver DRV1. As shown in Figure 2b, the input terminal of the driver DRV1 is used to input a switching control signal Vpwm. The high-voltage or low-voltage bias terminal of the driver DRV1 is connected to the critical saturation drive voltage generation module 110 to receive the critical saturation drive voltage Vcb. The output terminal of the driver DRV1 is used to output the drive signal Vg. The driver DRV1 is used to generate the drive signal Vg based on the switching control signal Vpwm using the critical saturation drive voltage Vcb as a bias. Therefore, the swing amplitude (or voltage level) of the drive signal Vg is limited by the critical saturation drive voltage Vcb. Therefore, when the power transistor 101 is turned on, it operates in the critical saturation conduction state due to the limitation of the critical saturation drive voltage Vcb. The switch control signal Vpwm may be various types of PWM (Pulse Width Modulation) signals, which is not limited in the present invention.

[0051] FIG3 shows a schematic structural diagram of an overcurrent protection module of a driving circuit according to a first embodiment of the present invention. As shown in FIG3 , in an exemplary embodiment, the overcurrent protection module 120 includes a transistor M2, a controlled current source 121, a sampling current acquisition unit 122, a voltage comparison unit 123, and a pull-down control unit 124. The overcurrent protection module 120 is configured to use the current path of the transistor M2 as a reference path. The overcurrent protection module 120 detects the current flowing through the transistor M2 to obtain a sampling current that is proportional to the overall current of the power transistor 101, and performs current detection on the power transistor 101 based on the sampling current. In addition, the overcurrent protection module 120 is further configured to determine whether a short circuit event has occurred at the voltage output terminal of the power transistor 101 based on the voltage difference between the sampling current and the voltage input / output terminal, and to shut down the power transistor 101 if a short circuit is triggered.

[0052] In an exemplary embodiment, the transistor M2 has the same transistor size as each of the multiple transistor branches (e.g., transistor branches M31, M32, M33, and M3n) in the power transistor 101. For example, the transistor M2 and the power transistor 101 are fabricated on the same wafer, and a portion of the transistor branches are selected from a MOSFET semiconductor die (also known as a die or chip) obtained from the wafer through a series of processing steps to serve as the transistor M2 (e.g., the ratio of the number of transistor branches of the transistor M2 to the number of transistor branches of the power transistor 101 is 1 / 100,000; of course, another number of branches may also be selected to serve as the transistor M2).

[0053] The transistor M2 has a first terminal (e.g., drain) connected to the voltage input terminal Vin, a control terminal (e.g., gate) connected to the drive signal Vg, and a second terminal connected to the first terminal of the controlled current source 121. The second terminal of the controlled current source 121 is grounded. The sampling current acquisition unit 122 adjusts the current of the controlled current source 121 so that the source voltage (or second terminal voltage) of the transistor M2 is consistent with the source voltage (or second terminal voltage) of the power transistor 101. Since the power transistor 101 also has a first terminal (e.g., drain) connected to the voltage input terminal Vin, a second terminal (e.g., source) connected to the voltage output terminal Vout, and a control terminal (e.g., gate) connected to the drive signal Vg, when the source voltages of the transistor M2 and the power transistor 101 are consistent, the transistor M2 and the power transistor 101 can be considered to be in the same state. At this time, the current ratio between the transistor M2 and the power transistor 101 is determined by the transistor size ratio between the two.

[0054] Taking the transistor size ratio between the transistor M2 and the power transistor 101 as 1 / 100000 as an example, when the controlled current source 121 makes the source voltage of the transistor M2 consistent with the source voltage of the power transistor 101, assuming that the current of the controlled current source 121 is equal to 1μA at this time, it means that the current in the power transistor 101 is equal to 1μA×100000=0.1A.

[0055] The voltage comparison unit 123 is configured to obtain a voltage difference between the input voltage Vin and the output voltage Vout, compare the voltage difference with a protection threshold, and output a comparison result.

[0056] The input of the pull-down control unit 124 is connected to the output of the sampling current acquisition unit 122 and the voltage comparison unit 123, and the output of the pull-down control unit 124 is connected to the control terminals (e.g., gates) of the power transistor 101 and the transistor M2. The pull-down control unit 124 is configured to determine whether to pull the gate voltage of the power transistor 101 down to ground based on the sampled current output by the sampling current acquisition unit 122 and the comparison result of the voltage comparison unit 123. When the sampled current indicates that the current in the power transistor 101 has reached the current limit value, and the voltage difference between the input voltage Vin and the output voltage Vout is greater than the protection threshold, the pull-down control unit 124 is triggered to turn on, pulling the gate voltages of the power transistor 101 and the transistor M2 down to ground.

[0057] In summary, the drive circuit 100 of this embodiment uses a transistor M1 having the same transistor size as each transistor branch in the power transistor 101 to obtain a critical saturation drive voltage Vcb through the critical saturation drive voltage generation module 110. The drive signal Vg for driving the multiple transistor branches in the power transistor 101 is then derived based on the critical saturation drive voltage Vcb, so that the power transistor 101 can operate in a critical saturation conduction state when it is turned on. When a short circuit occurs at the voltage output terminal of the power transistor 101, the current in the power transistor 101 will increase. However, since the multiple transistor branches in the power transistor 101 all operate in the critical saturation conduction state, if the increased current exceeds their rated current, the multiple transistor branches in the power transistor 101 enter a current limiting state, limiting the current in the power transistor 101 to the rated current (e.g., 10A). The overcurrent protection module 120 then uses a transistor M2 having the same transistor size as each transistor branch in the power transistor 101 to obtain a sampled current proportional to the overall current of the power transistor 101. Among them, when the sampling current in the transistor M2 indicates that the current of the entire power transistor 101 reaches the current limit value, and the voltage difference between the voltage input / output terminals is higher than the protection threshold, the overcurrent protection module 120 determines that the power transistor 101 triggers a short circuit event at this time, and pulls down the gate voltage of the power transistor 101 according to the judgment result to turn off the power transistor 101.

[0058] Furthermore, the drive circuit 100 of this embodiment further includes a coupling capacitor C1 connected between the control terminals (e.g., gates) and second terminals (e.g., sources) of the multiple transistor branches of the power transistor 101. Because parasitic capacitances Cgd and Cgs exist between the gate-drain and gate-source of a MOSFET, respectively (as shown in transistor branch M3n in FIG1 and FIG3 ), when the source of the power transistor 101 is rapidly discharged instantaneously due to a short-circuit event, a large voltage difference is generated between the parasitic capacitances Cgd and Cgs. Because the parasitic capacitances Cgd and Cgs are of equal size under existing process technologies, the voltage divider effect of the parasitic capacitances Cgd and Cgs increases the gate voltages of the multiple transistor branches in the power transistor 101, causing the transistor branches M31 to M3n to exit the critical saturation conduction state. Therefore, the coupling capacitor C1 of this embodiment can be used to increase the coupling between the gate and source of the multiple transistor branches M31, M32, M33 and M3n in the power transistor 101, thereby preventing the gate voltage of the transistor branches M31, M32, M33 and M3n from being raised too much due to the voltage division effect of the parasitic capacitance during instantaneous discharge, and ensuring that the multiple transistor branches in the power transistor 101 can enter the current limiting state after a short circuit event occurs.

[0059] Furthermore, the driving circuit 100 of this embodiment can be integrated with the power transistor 101 in the same chip for use in various scenarios, such as a high-current static load switch or a high-speed dynamic load switch, a high-current switching power supply, a high-current LDO (low dropout regulator), and a high-current high-side switch / low-side switch, etc.

[0060] Figure 4 shows a schematic circuit diagram of a power module according to a second embodiment of the present invention. As shown in Figure 4, the power module 200 of this embodiment is, for example, a load switch circuit. The load switch circuit 200 includes a power switch connected between a power source 210 and a load 220, and a drive circuit 100. The drive circuit 100 is configured to provide a drive signal Vg to a control terminal (e.g., a gate) of the power switch based on a switch control signal Vpwm to control the on and off state of the power switch, thereby enabling current transfer between the power source 210 and the load 220. The power switch can be implemented using the power transistor 101 described in the first embodiment. The drive circuit 100 includes a critical saturation drive voltage generation module 110, an overcurrent protection module 120, and a driver DRV1. The critical saturation drive voltage generation module 110 is configured to obtain a critical saturation drive voltage Vcb and provide the critical saturation drive voltage Vcb to a voltage bias terminal on a high potential side or a low potential side of the driver DRV1. The driver DRV1 is configured to generate the drive signal Vg based on the switch control signal Vpwm using the critical saturation drive voltage Vcb as a bias, such that the swing amplitude (or level voltage) of the drive signal Vg is limited by the critical saturation drive voltage Vcb. Therefore, when the power transistor 101 is turned on, it operates in the critical saturation conduction state due to the limitation of the critical saturation drive voltage Vcb, thereby enabling the multiple transistor branches in the power transistor 101 to enter a current limiting state when a short circuit occurs at the load end. The specific process has been described in detail in the above embodiments and will not be repeated here. The switch control signal Vpwm can be various types of PWM (Pulse Width Modulation) signals, which are not limited in the present invention.

[0061] The overcurrent protection module 120 is used to detect the current in the power transistor 101 and, when the current in the power transistor 101 is equal to the rated current (i.e., the power transistor 101 is operating in a current-limiting state at this time) and the voltage difference between the input voltage Vin and the output voltage Vout is greater than a protection threshold, lower the gate voltage of the power transistor 101 to prevent damage to the power transistor 101 caused by a large current flowing through the power transistor 101 for a long time when a load at the voltage output end is short-circuited. The specific process has been described in detail in the above embodiments and will not be repeated here.

[0062] FIG5 shows a schematic circuit diagram of another power module according to a third embodiment of the present invention. As shown in FIG5 , the power module 300 of this embodiment is, for example, a switching power converter, which includes a power circuit having one or more switching elements and filter elements (e.g., inductors and / or capacitors). The one or more switching elements and filter elements are configured to regulate the transfer of power from the input to the output of the switching converter in response to a switch drive signal, thereby converting an input voltage Vin into a stable and continuous output voltage Vout.

[0063] In some embodiments, the switching power converter 300 can be divided into a buck converter, a boost converter, a flyback converter, and a buck-boost converter according to the topology of the power circuit.

[0064] In an exemplary embodiment, the power circuit is implemented using a buck topology and includes a high-side switch MD1, a low-side switch MD2, and an inductor Lx. The high-side switch MD1 and the low-side switch MD2 are connected between a voltage input terminal Vin and ground. A first end of the inductor Lx is connected to a switching node SW between the high-side switch MD1 and the low-side switch MD2, and a second end of the inductor Lx is connected to a voltage output terminal Vout. The high-side switch MD1 and the low-side switch MD2 can be any controllable semiconductor switching device, such as a metal oxide semiconductor field effect transistor (MOSFET) or an insulated gate bipolar transistor (IGBT), for example, an N-channel MOSFET. These devices are controlled by gate drive signals HSD and LSD, respectively, to alternate between a first state and a second state (e.g., an on state and an off state). Furthermore, the power circuit of this embodiment includes an output capacitor Cout connected between the voltage output terminal Vout and the ground node.

[0065] Furthermore, the switching power converter 300 of this embodiment further includes a high-side driver 310 and a low-side driver 320. The high-side driver 310 is configured to provide a gate drive signal HSD to the high-side switch MD1 based on the high-side control signal HSON to control the on / off state of the high-side switch MD1. The low-side driver 320 is configured to provide a gate drive signal LSD to the low-side switch MD2 based on the low-side control signal LSON to control the on / off state of the low-side switch MD2. The high-side control signal HSON and the low-side control signal LSON can be various types of PWM (Pulse Width Modulation) signals.

[0066] It should be noted that although MOSFETs are used as switching elements in this embodiment, any other suitable type of switching element may be used without departing from the principles of the present invention. Furthermore, although this embodiment is described using a synchronous buck converter, the present invention is not limited thereto and is equally applicable to non-synchronous buck converters. Those skilled in the art may also use a rectifier diode in place of the low-side switch MD2 in the above embodiment.

[0067] Furthermore, the high-side switch MD1 of this embodiment can be implemented by the power transistor 101 of the first embodiment, and the high-side driver 310 can be implemented by the driving circuit 100 of the first embodiment. As previously described, the driving circuit 100 includes a critical saturation driving voltage generating module 110, an overcurrent protection module 120, and a driver DRV1. The critical saturation drive voltage generation module 110 is used to obtain a critical saturation drive voltage Vcb and provide the critical saturation drive voltage Vcb to the voltage bias terminal on the high potential side or the low potential side of the driver DRV1. The driver DRV1 is used to generate the gate drive signal HSD based on the upper tube control signal HSON with the critical saturation drive voltage Vcb as the bias. Therefore, the swing amplitude (or level voltage) of the gate drive signal HSD is limited by the critical saturation drive voltage Vcb. Therefore, when the high-end switch MD1 is turned on, it will operate in the critical saturation conduction state due to the limitation of the critical saturation drive voltage Vcb, so that the multiple transistor branches in the high-end switch MD1 can enter the current limiting state when a short circuit event occurs at the voltage output terminal Vout. The specific process has been described in detail in the above embodiment and will not be repeated here.

[0068] It should be noted that although the high-side switch MD1 is used as an example in this embodiment, the power transistor and its driving circuit 100 of the first embodiment of the present invention can also be applied to the low-side switch MD2 in the switching power converter 300. The specific process is similar to that of the high-side switch MD1 and will not be repeated here.

[0069] Figure 6 shows a schematic circuit diagram of another power module according to a fourth embodiment of the present invention. The power module 400 of this embodiment is, for example, a low-dropout linear regulator (LDO), comprising a power transistor MP, an error amplifier EA, and a driver circuit 100 connected between a voltage input terminal Vin and a load 401. The power transistor MP is the chip's primary output transistor, with a first terminal (e.g., drain) receiving an input voltage Vin and a second terminal providing an output voltage Vout to the load 401. The error amplifier EA compares feedback of the output voltage Vout with a reference voltage Vref and outputs an error signal representing the deviation between the two. The input of the driver circuit 100 is connected to the output of the error amplifier EA, and the output of the driver circuit 100 is connected to the control terminal (e.g., gate) of the power transistor MP. The driver circuit 100 drives the power transistor MP based on the output of the error amplifier EA, thereby controlling current flow between the voltage input terminal and the load.

[0070] Furthermore, the power transistor MP of this embodiment can be implemented by the power transistor 101 of the first embodiment. The driving circuit 100 is, for example, the driving circuit of the first embodiment described above. The driving circuit 100 includes a critical saturation driving voltage generating module 110, an overcurrent protection module 120, and a driver DRV1. The critical saturation driving voltage generating module 110 is configured to obtain a critical saturation driving voltage Vcb and provide the critical saturation driving voltage Vcb to a voltage bias terminal on a high potential side or a low potential side of the driver DRV1. The driver DRV1 is configured to generate the driving signal Vg based on the error signal output by the error amplifier EA using the critical saturation driving voltage Vcb as a bias. Therefore, the swing amplitude (or level voltage) of the driving signal Vg is limited by the critical saturation driving voltage Vcb. Therefore, when the power transistor 101 is turned on, it operates in the critical saturation conduction state due to the limitation of the critical saturation driving voltage Vcb. This allows the multiple transistor branches in the power transistor 101 to enter a current limiting state when a short circuit occurs at the load end. The specific process has been described in detail in the above embodiment and will not be repeated here.

[0071] The overcurrent protection module 120 is used to detect the current in the power transistor 101 and turn off the power transistor 101 when the current in the power transistor 101 is equal to the rated current (that is, the power transistor 101 is operating in a current-limiting state at this time) and the voltage difference between the input voltage Vin and the output voltage Vout is greater than a protection threshold, so as to prevent the power transistor 101 from being damaged due to a large current flowing through it for a long time when a load at the voltage output end is short-circuited. The specific process has been described in detail in the above embodiment and will not be repeated here.

[0072] It should be noted that the combination of the power transistor and its driving circuit in this embodiment can also be used in other scenarios requiring high-current MOSFETs. The present invention is not limited to this. Those skilled in the art can set the power transistor and its driving circuit of the present invention in high-power applications according to specific circumstances, so that the MOSFET in the high-power application can be turned off in time in the event of a rapid short circuit, and the current overshoot in the circuit can be reduced.

[0073] In summary, the present invention provides a driving circuit for a power transistor, the driving circuit comprising a critical saturation driving voltage generating module and an overcurrent protection module, wherein the critical saturation driving voltage generating module extracts a portion of the transistor branches from the plurality of transistor branches in the power transistor, and uses a negative feedback regulation method to make these portion of the transistor branches behave as a constant current source, so that the voltage driving these portion of the transistor branches is the critical saturation driving voltage, and then the critical saturation driving voltage is used to make the plurality of transistor branches in the power transistor be in a critical saturation conduction state. If the current in the power transistor is less than the rated current, the power transistor is in a saturated conduction state, with a small on-resistance and low loss, and can provide a strong current transmission capability. Once the current in the power transistor exceeds its rated current due to a short circuit, since the MOSFET structure is an intrinsic characteristic of the transconductance device, the power transistor naturally enters a current limiting state, and then when the protective disconnection action is performed by the overcurrent protection module, no excessive switching stress will occur. Therefore, the driving circuit of the present invention can not only provide effective protection in the event of a short circuit, but also control the current change rate when disconnecting to reduce the generation of a kickback voltage.

[0074] The embodiments of the present invention are described above, but these embodiments do not describe all details in detail, nor do they limit the invention to specific embodiments. Obviously, many modifications and variations can be made based on the above description. This specification selects and specifically describes these embodiments in order to better explain the principles and practical applications of the present invention, so that those skilled in the art can make good use of the present invention and modify and use it based on the present invention. The scope of protection of the present invention shall be based on the scope defined by the claims of the present invention.

Claims

1. A driving circuit of a power transistor, wherein the power transistor is connected between a voltage input terminal and a voltage output terminal, and the driving circuit comprises: A critical saturation driving voltage generating module, used for generating a critical saturation driving voltage; a buffer module, configured to output a driving signal applied to the control terminal of the power transistor based on the critical saturation driving voltage, so that the power transistor is in a critical saturation conduction state, wherein in the critical saturation conduction state, when the current of the power transistor is less than the rated current, the power transistor operates in a saturation conduction state, and when the current of the power transistor is greater than the rated current, the power transistor operates in a current limiting state due to the limitation of the driving signal; and An overcurrent protection module, used for detecting the current of the power transistor to determine whether the power transistor has an overcurrent, and controlling the power transistor to shut down when an overcurrent is detected in the power transistor; Wherein, when a short circuit occurs at the voltage output terminal, the power transistor first enters a current limiting state due to being in a critical saturation conduction state, and then the power transistor is turned off by the overcurrent protection module.

2. The driving circuit according to claim 1, wherein: The critical saturation driving voltage generating module comprises: A first transistor, a first terminal of which is connected to the voltage input terminal; a state control module, wherein the state control module is configured to control the voltage between the first terminal and the second terminal of the first transistor to be a preset voltage value and the current of the first transistor to be a preset current value, so that the first transistor is in a critical saturation conduction state; Wherein, the voltage on the control terminal of the first transistor is the critical saturation driving voltage.

3. The driving circuit according to claim 2, wherein: The state control module comprises: a constant current source connected to the second end of the first transistor; an operational amplifier, wherein the positive input terminal thereof is connected to a reference voltage source, the negative input terminal thereof is connected to a common connection node of the first transistor and the constant current source, and the output terminal thereof is connected to the control terminal of the first transistor, The operational amplifier adjusts the first transistor through negative feedback to obtain the critical saturation driving voltage.

4. The driving circuit according to claim 2, wherein: The overcurrent protection module comprises: A second transistor and a controlled current source are connected in series between the voltage input terminal and ground, wherein a control terminal of the second transistor is connected to the driving signal; a sampling current acquisition unit, configured to adjust the current in the controlled current source so that the second terminal voltage of the second transistor is equal to the second terminal voltage of the power transistor, and obtain a sampling current according to the current of the controlled current source at this time, wherein the sampling current has a second current ratio with the current in the power transistor; a voltage comparison unit, configured to obtain a voltage difference between an input voltage of the voltage input terminal and an output voltage of the voltage output terminal; and A pull-down control unit, whose input end is connected to the output of the sampling current acquisition unit and the voltage comparison unit, and whose output is connected to the control end of the power transistor, is used to determine whether to pull the control end of the power transistor down to ground based on the sampling current acquisition unit and the voltage comparison unit.

5. The driving circuit according to claim 4, wherein: The power transistor includes a plurality of third transistors connected in parallel between the voltage input terminal and the voltage output terminal, and control terminals of the plurality of third transistors are connected to the driving signal. The first transistor, the second transistor and the third transistor have the same transistor size.

6. The driving circuit according to claim 4, wherein: The first transistor, the second transistor, and the power transistor are formed on a same semiconductor die.

7. The driving circuit according to claim 1, further comprising: A coupling capacitor is connected between the control terminal and the second terminal of the power transistor.

8. The driving circuit according to claim 1, wherein: The buffer module includes a driver, the input end of the driver is used to receive a switch control signal, the voltage bias end of the high potential side or the low potential side of the driver is used to receive the critical saturation drive voltage, and the output end of the driver is used to output the drive signal.

9. The driving circuit according to claim 1, wherein: The buffer module comprises an isolation resistor, a first end of the isolation resistor is connected to the critical saturation driving voltage, and a second end of the isolation resistor is used to output the driving signal.

10. A load switch circuit, comprising the power transistor according to any one of claims 1 to 9 and a driving circuit for driving the power transistor.

11. A power module, comprising the power transistor according to any one of claims 1 to 8 and a driving circuit for driving the power transistor.

12. The power module according to claim 11, wherein: The power supply module includes a switching power converter or a low voltage drop linear regulator.

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