Double-rotor magnetic flux switching permanent magnet electric motor, double-rotor magnetic flux switching permanent magnet electric motor optimization method and optimization device, medium and product

Abstract

Disclosed in the present application are a double-rotor magnetic flux switching permanent magnet electric motor, a double-rotor magnetic flux switching permanent magnet electric motor optimization method and optimization device, a medium and a product. The double-rotor magnetic flux switching permanent magnet electric motor comprises an inner rotor, an outer rotor and an intermediate stator, wherein the intermediate stator comprises a stator core; permanent magnets are arranged inside the stator core on two sides thereof; the stator core comprises a plurality of toothed portions; and an excitation winding is arranged between every two adjacent toothed portions.

Description

Dual-rotor flux-switching permanent magnet motor, optimization method, device, medium and product Cross-reference to Related Applications

[0001] This application claims priority to Chinese Patent Application No. 2024104283554, filed on April 10, 2024, the entire contents of which are incorporated herein by reference. TECHNICAL FIELD

[0002] The present disclosure relates to the technical field of mechanical rotors, and in particular to a dual-rotor flux-switching permanent magnet motor, an optimization method, a device, a medium and a product. BACKGROUND

[0003] With the increasing depletion of oil energy and the sudden increase of air pollutants, fuel vehicles with large amounts of pollutant emissions have gradually been replaced by new energy vehicles, among which new energy vehicles relying on hybrid power and pure electric technology are receiving widespread attention. Although fuel cells and pure electric vehicles have advantages such as cleanliness and quietness that traditional fuel vehicles do not have, the short battery range and high price of hydrogen fuel seriously limit their application and popularization. Therefore, hybrid electric vehicles have the advantages of both fuel and pure electric vehicles, and can achieve the driving goals of reducing vehicle fuel consumption, prolonging battery life, and minimizing exhaust emissions by combining advanced energy management strategies. However, the current mainstream electric structure is a combination of a generator P1 and a drive motor P3, which has high requirements for the layout space of the vehicle, which is not conducive to the spatial arrangement of new energy vehicles. SUMMARY

[0004] The present disclosure provides a dual-rotor flux-switching permanent magnet motor, an optimization method, a device, a medium and a product, aiming to solve the technical problem that the vehicle cannot meet the tight space in new energy vehicles due to the large space waste of the motor of the hybrid electric vehicle.

[0005] According to a first aspect of the present disclosure, a dual-rotor flux-switching permanent magnet motor is provided, which includes an inner rotor, an outer rotor and an intermediate stator, the intermediate stator includes a stator core, permanent magnets are built-in on both sides of the stator core, the stator core includes a plurality of tooth portions, and an excitation winding is arranged between adjacent tooth portions.

[0006] According to a second aspect of the present disclosure, a dual-rotor flux-switching permanent magnet motor optimization method is provided, which is applied to the dual-rotor flux-switching permanent magnet motor as described in the first aspect. The dual-rotor flux-switching motor optimization method comprises: determining the number of phase group modules of the dual-rotor flux-switching permanent magnet motor, and obtaining power grid harmonics according to the number of phase group modules; determining the inner torque of the inner rotor and the outer torque of the outer rotor; obtaining the motor fundamental frequency according to the rotor pole number and the rotating speed, and obtaining the electrical period according to the motor fundamental frequency; performing Fourier analysis on the electrical period of the power grid harmonics in the inner torque and the outer torque to obtain suppression harmonics; analyzing the suppression harmonics to obtain target suppression harmonics; performing simulation analysis on the target suppression harmonics to obtain an effective shift angle, and decoupling the effective shift angle to obtain a target shift angle; and optimizing the dual-rotor flux-switching permanent magnet motor according to the target shift angle.

[0007] According to a third aspect of the present disclosure, a dual-rotor flux-switching permanent magnet motor optimization device is provided, which is applied to the dual-rotor flux-switching permanent magnet motor as described above. The dual-rotor flux-switching motor optimization device comprises: a first determination module configured to determine the number of phase group modules of the dual-rotor flux-switching permanent magnet motor, and obtain power grid harmonics according to the number of phase group modules; a second determination module configured to determine the inner torque of the inner rotor and the outer torque of the outer rotor; an electrical period acquisition module configured to obtain the motor fundamental frequency according to the rotor pole number and the rotating speed, and obtain the electrical period according to the motor fundamental frequency; a first analysis module configured to perform Fourier analysis on the electrical period of the power grid harmonics in the inner torque and the outer torque to obtain suppression harmonics; a second analysis module configured to analyze the suppression harmonics to obtain target suppression harmonics; a third analysis module configured to perform simulation analysis on the target suppression harmonics to obtain an effective shift angle, and decouple the effective shift angle to obtain a target shift angle; and a processing module configured to optimize the dual-rotor flux-switching permanent magnet motor according to the target shift angle.

[0008] According to a fourth aspect of the present disclosure, a computer readable storage medium is provided, which stores a computer program. The program is executed by a processor to implement the steps of the dual-rotor flux-switching permanent magnet motor optimization method as described above.

[0009] According to a fifth aspect of the present disclosure, a computer program product is provided, which comprises a computer program. The computer program is executed by a processor to load and execute the steps of the dual-rotor flux-switching permanent magnet motor optimization method as described above. BRIEF DESCRIPTION OF DRAWINGS

[0010] FIG. 1 is a schematic diagram of a dual-motor parallel shaft configuration in a hybrid vehicle according to some embodiments of the present disclosure;

[0011] FIG. 2 is a schematic diagram of a dual-motor concentric shaft configuration in a hybrid vehicle according to some embodiments of the present disclosure;

[0012] FIG. 3 is a schematic diagram of a structure of a stator permanent magnet type motor according to some embodiments of the present disclosure;

[0013] FIG. 4 is a schematic diagram of a structure of a stator permanent magnet type dual-rotor motor according to some embodiments of the present disclosure;

[0014] FIG. 5 is a schematic diagram of a structure of an FSPM motor with improved core according to some embodiments of the present disclosure;

[0015] FIG. 6 is a topology diagram of a dual-rotor flux switching motor with a shift angle of 0° according to some embodiments of the present disclosure;

[0016] FIG. 7 is a topology diagram of a dual-rotor flux switching motor with a shift angle of 90° according to some embodiments of the present disclosure;

[0017] FIG. 8 is a schematic diagram of an E-shaped core and an I-shaped permanent magnet according to some embodiments of the present disclosure;

[0018] FIG. 9 is a schematic diagram of a structure of a dual-rotor flux switching motor according to some embodiments of the present disclosure;

[0019] FIG. 10 is a flowchart of an optimization method of a dual-rotor flux switching motor according to some embodiments of the present disclosure;

[0020] FIG. 11 is a frequency spectrum diagram of a cogging torque of a dual-rotor flux switching motor according to some embodiments of the present disclosure; and

[0021] FIG. 12 is a schematic diagram of a relationship between an inner-outer coupling degree and a shift angle of a dual-rotor motor according to some embodiments of the present disclosure. DETAILED DESCRIPTION

[0022] It should be understood that the specific embodiments described herein are merely illustrative of the present disclosure and do not limit the scope of the present disclosure.

[0023] With the increasing capacity of batteries loaded on hybrid electric vehicles, there is a higher requirement for miniaturization, integration and light weight of the hybrid powertrain. The current hybrid electric vehicle is composed of an engine, a generator P1 and a drive motor P3 in a hybrid manner, and various driving modes such as pure electric driving, engine direct driving, parallel driving, series power generation and energy recovery are formed through mode switching, which can effectively avoid the range anxiety of pure electric vehicles and the emission anxiety of fuel vehicles. The current mainstream hybrid mode is generally a P1+P3 dual-motor structure, which realizes the integration of the generator, the drive motor and the reduction gear set through parallel shafts or concentric shafts. Referring to FIG. 1 and FIG. 2, FIG. 1 is a dual-motor parallel shaft configuration according to some embodiments of the present disclosure, and FIG. 2 is a dual-motor concentric shaft configuration according to some embodiments of the present disclosure.

[0024] However, the above schemes all have high requirements for the layout space of the hybrid powertrain in the vehicle, and the parallel shaft configuration increases the X and Z direction space of the whole machine, and the concentric shaft configuration has higher requirements for the Y direction space, which is contradictory to the tight layout space of new energy vehicles.

[0025] High-performance electric machines as power sources and moving parts of new energy vehicles are the core and key to their development towards high-end. Current electric machines are mainly divided into three categories according to the excitation source: induction motor, switched reluctance motor and permanent magnet motor. Induction motors have simple structure, high reliability and mature control technology, but have large eddy current loss, low power factor and low efficiency. Switched reluctance motors have mature manufacturing process, but have large torque ripple, large system vibration and noise, and poor low-speed high-torque performance. Permanent magnet motors use permanent magnets as excitation sources, have high power factor, and have higher efficiency, power and torque density than the first two types of motors, and have become the mainstream. Current permanent magnet synchronous motors are mainly rotor-type permanent magnet motors, including surface-mounted and embedded structures.

[0026] Surface-mounted: Since the surface-mounted magnet directly contacts the air gap magnetic field, under the action of the harmonic magnetic field, the permanent magnet has large eddy current loss, which easily worsens the temperature rise of the permanent magnet. Surface-mounted permanent magnet motors need to take protective measures for the permanent magnet, such as using non-magnetic alloy sheath or carbon fiber binding technology, which will cause heat dissipation difficulties, even irreversible demagnetization, and reduce operation reliability. At the same time, the permanent magnet protection sleeve will increase the equivalent air gap length, reducing the output torque and power density of the motor.

[0027] Embedded: The permanent magnet is embedded in the rotor, and the rotating centrifugal stress will have a certain impact on the strength of the rotor. The complex magnetic isolation structure also brings about magnetic leakage and saturation problems, limiting the improvement of power and torque density, and the structure design is relatively complex. Current traditional permanent magnet motors mostly use distributed windings, which have long ends, large winding resistance and high mutual inductance, resulting in large copper loss, low efficiency, complex winding and other problems, and it is difficult to realize modularization and fault-tolerant design of the motor.

[0028] Currently, the stator permanent magnet type motor mainly includes a doubly salient permanent magnet motor, a flux reversal permanent magnet motor and a flux switching permanent magnet motor. Referring to FIG. 3, FIG. 3 is a structural schematic diagram of a stator permanent magnet type motor according to some embodiments of the present disclosure. The flux switching permanent magnet motor (FSPM motor) has the following advantages: the stator presents a modular arrangement, the mutual inductance is relatively small and has a high fault tolerance; there is a magnetic aggregation effect between the tangential relative permanent magnet and the permanent magnet, so the torque density is high; the magnetic motive force of the permanent magnet is not directly connected in series with the armature main magnetic flux, and the anti-demagnetization capability is strong; the induced back electromotive force is very high in sine degree, and the harmonic control degree is good.

[0029] For the low-speed large-torque working condition requirement, the permanent magnet motor has a large split ratio (the ratio of the inner diameter to the outer diameter of the motor stator), so the internal space of the motor cannot be effectively utilized. Referring to FIG. 4, FIG. 4 is a structural schematic diagram of a stator permanent magnet type double-rotor motor according to some embodiments of the present disclosure. The double-rotor permanent magnet motor with a double-gap structure can flexibly utilize the excessive remaining space due to the addition of a layer of air gap. Moreover, the inner and outer motor designs are also flexible, which can be used as an auxiliary generator or a driving motor, and the overall torque density of the motor is large and the anti-saturation performance is strong.

[0030] Due to the large split ratio of the traditional motor, the number of teeth and slots of the stator is relatively large. The stator core with a split tooth structure design enlarges the number of tooth slot torque cycles, so that the amplitude of the tooth slot torque harmonic is relatively reduced. Referring to FIG. 5, FIG. 5 is a structural schematic diagram of an improved FSPM motor with a core according to some embodiments of the present disclosure. From the fault tolerance aspect, the stator split tooth structure and the E-type core structure have a high degree of modularization in the magnetic circuit, so they have a higher self-inductance and a lower mutual inductance than the traditional FSPM motor, thereby effectively limiting the amplitude of the short-circuit current.

[0031] The traditional double-port FSPM motor structure is to design the inner and outer motors as two independent motors, add a magnetic separation layer to the stator yoke after separating the permanent magnets, and try to avoid the interference of the inner and outer magnetic field coupling on the driving control.

[0032] In combination with the above description, the high-performance double-electron topology structure is proposed in the embodiments of the present disclosure. Referring to FIG. 6 and FIG. 7, FIG. 6 is a topology diagram of a double-rotor flux switching motor with a shift angle of 0° according to some embodiments of the present disclosure, and FIG. 7 is a topology diagram of a double-rotor flux switching motor with a shift angle of 90° according to some embodiments of the present disclosure. The flux switching principle is combined with the double-rotor and stator structure. The motor can be understood as a combination of an outer motor and an inner motor. The intermediate stator provides a magnetic flux loop for the inner and outer permanent magnets to realize the flux switching principle, thereby improving the integration of the motor. Similar to the traditional flux switching permanent magnet motor, in the double-rotor flux switching motor topology structure proposed in the embodiments of the present disclosure, the N-pole and S-pole permanent magnets are radially built-in on both sides of the iron core to realize the magnetic concentration effect. To realize the alternating concentration of magnetic flux and form a balanced three-phase counter electromotive force, the pole-slot matching relationship needs to be limited, and the circumferential geometric size is also constrained. In the present disclosure, the stator core includes a plurality of tooth portions, and the arc width of the center line of adjacent tooth portions is greater than or equal to 0 and less than or equal to π electrical angle. In some embodiments, to obtain a unit winding factor, the arc width of the center line of the adjacent two tooth portions in the same iron core module can be set to π electrical angle (rad), and at the same time, the stator tooth width and the rotor pole width, and the phase inner slot width are all the same, that is, the unit angle θ = π / 2. Thus, the phase angle difference of the adjacent two coils is 2π electrical angle. In the stator structure in the double-rotor flux switching motor topology diagram, the double-rotor structure shares the common stator core and permanent magnet. For example, an E-type core and an I-type permanent magnet are used for phase-to-phase and phase-to-phase, referring to FIG. 8, which is a schematic diagram of an E-type core and an I-type permanent magnet according to some embodiments of the present disclosure.

[0033] In the double-rotor flux switching motor, the two rotors need to maintain the same speed and direction of movement to realize the alternating magnetic concentration effect, so they need to be fixedly connected to the same output shaft through mechanical tooling (flange plate), referring to FIG. 9, which is a structural diagram of a double-rotor flux switching motor according to some embodiments of the present disclosure. The inner rotor and the outer rotor are fixedly connected to the same motor shaft through mechanical tooling, and the intermediate stator is fixed to the motor housing. The excitation winding is also included between the inner and outer rotors.

[0034] The double-rotor flux switching permanent magnet motor proposed in the present disclosure is a concentric double-rotor double-electricity structure, which can greatly reduce the axial size and longitudinal height of the hybrid system, reserve more space for vehicle layout, design the permanent magnet and winding in the stator at the same time, make the motor system heat path shorter and easier to dissipate heat, effectively improve the torque density, and provide high winding factor and short end length for the phase group concentrated winding, reduce the mutual inductance ratio, improve the fault tolerance, realize the topology modularization, and realize the magnetic flux alternating concentration through the shift angle design, which can effectively improve the utilization rate of permanent magnets and effectively suppress the cogging torque.

[0035] The embodiment of the present disclosure provides a double-rotor flux switching motor optimization method, referring to FIG. 10, which is a flowchart of a double-rotor flux switching motor optimization method according to some embodiments of the present disclosure, including the following steps S10 to S70.

[0036] In step S10, the number of phase group modules of the double-rotor flux switching permanent magnet motor is determined, and the power grid harmonic is obtained according to the number of phase group modules.

[0037] It should be noted that the double-rotor flux switching motor optimization method provided by the present disclosure can be applied to a double-rotor flux switching motor optimization device. The double-rotor flux switching motor optimization device has functions of data processing, data communication, program running, etc. The double-rotor flux switching motor optimization device can be an integrated controller, a control computer, etc. Of course, it can also be other devices with similar functions, and the present embodiment does not limit this.

[0038] It can be understood that the number of phase group modules of the double-rotor flux switching permanent magnet motor refers to the number of internal coils in the double-rotor flux switching permanent magnet motor. When the number of phase group modules of the double-rotor flux switching permanent magnet motor is three, that is, the double-rotor flux switching permanent magnet motor is a three-phase motor, the step angle of the three-phase motor is 0.75° / 1.5°. The power grid harmonic is the component greater than the fundamental frequency obtained by Fourier series decomposition of the periodic sinusoidal alternating current. The power grid harmonic will cause the power factor to decrease, the power loss, etc., that is, it will cause the waste of electric energy.

[0039] In some embodiments, according to the double-rotor flux switching permanent magnet motor, the number of coils n1 in one phase group and the corresponding number of phase groups n2 can be determined. The number of coils in one phase group is also the number of I-type permanent magnets. According to the characteristic that the stator core structure presents a modular phase group, the number of stator slots can be corrected to the number of phase group modules, that is, 3n2. Combined with the pole-slot matching formula, the following can be obtained:

[0040]

[0041] In the above formula, N s is the number of stator slots, N r is the number of rotor poles, GCD(N s ,N r ) is the greatest common divisor of the number of stator slots N s and the number of rotor poles N r , and k is a natural number. Compared with the modular design structure of the centralized winding in the phase group, the motor cogging torque is mainly the 6kth harmonic, that is, there are 6k wave peaks in one electric period.

[0042] In step S20, the inner torque of the inner rotor and the outer torque of the outer rotor are determined.

[0043] It should be noted that the inner torque refers to the torque of the inner rotor, and the outer torque refers to the torque of the outer rotor. When the torque of the rotor is determined, the torque can be obtained by dividing the mechanical power output by the mechanical angular velocity of the rotor.

[0044] In some embodiments, the determination of the inner torque of the inner rotor and the outer torque of the outer rotor comprises: determining the number of I-type permanent magnets and the number of phase groups; obtaining the mechanical angle according to the electrical angle and the number of rotor poles; obtaining the inner torque of the inner rotor according to the mechanical angle and the number of rotor poles; and obtaining the outer torque of the outer rotor according to the mechanical angle, the number of rotor poles and the phase difference.

[0045] In some embodiments, in order to obtain balanced three-phase no-load counter electromotive force, the phase angle difference between the adjacent two coils in the phase should be 2kπ+4π / 3 (k is a natural number), and a displacement angle needs to be designed for the inner and outer rotors of the double-rotor motor to realize this scheme. For the relationship between the unit angle θ and the number of phase coils and the number of phase groups, the following relationship can be obtained:

[0046]

[0047] At the same time, in order to meet the three-phase, the number of stator slots N s on either side needs to satisfy: N s =3n2. In the n1=6, n2=2 pole-slot matched double-rotor flux switching permanent magnet motor, the mechanical angle θ e can be obtained according to the electrical angle θ r and the number of rotor poles N r , and the calculation formula can be:

[0048]

[0049] Since the rotor pole pitch is equal to 4θ, the number of rotor poles N

[0050] The inner torque of the inner rotor can be obtained according to the mechanical angle θ r and the number of rotor poles N r :

[0051]

[0052] The outer torque of the outer rotor can be obtained according to the mechanical angle, the number of rotor poles and the phase difference:

[0053]

[0054] In the above formula, T2, T5, T6 and T7 respectively represent the tooth slot torques corresponding to the 2nd, 5th, 6th and 7th harmonics of the motor, respectively represent initial phase angles corresponding to the m-th harmonic of the motor 2, 5, 6, 7, and Δθ represents a phase difference caused by a shift angle.

[0055] In step S30: obtain the motor fundamental frequency according to the rotor pole number and the rotating speed, and obtain the electrical period according to the motor fundamental frequency.

[0056] In some embodiments, obtaining the motor fundamental frequency according to the rotor pole number and the rotating speed, and obtaining the electrical period according to the motor fundamental frequency can be obtained by the relationship between the motor fundamental frequency f and the rotor pole number N r and the rotating speed n, and the relationship expression is:

[0057]

[0058] Through the above steps, the motor fundamental frequency at this time can be obtained, and the current electrical period is obtained based on the motor fundamental frequency, and the electrical period is the reciprocal of the product of the motor fundamental frequency and the pole pair number.

[0059] In step S40: Fourier analysis is performed on the power grid harmonics in the electrical period of the inner and outer torques to obtain the suppression harmonics.

[0060] It should be noted that the suppression harmonics refer to the relative angle mechanical energy adjustment of the inner and outer rotors, so that the harmonics with larger amplitude in the torques received by the two rotors can be peak-valley counteracted and superimposed, and the harmonics with greater influence on the rotors are obtained.

[0061] In some embodiments, the harmonic component of the power grid harmonics is determined, the greatest common divisor of the stator slot number and the rotor pole number is obtained according to the stator slot number and the rotor pole number, the order of the harmonic component is obtained according to the stator slot number, the greatest common divisor of the stator slot number and the rotor pole number, the motor cogging torque is obtained according to the harmonic component, the order of the harmonic component, the rotor pole number and the mechanical angle, and the suppression harmonics are obtained according to the motor cogging torque according to the power grid harmonics. Fourier analysis is performed on the electrical period of the inner and outer torques, Fourier expansion is performed on the cogging torque of the FSPM motor, and the Fourier expansion expression is:

[0062]

[0063]

[0064] In the above formula, T m is the m-th harmonic component. GCD(N s , N r ) represents the greatest common divisor of the stator slot number N s and the rotor pole number N r , θ rrepresents the rotor position angle, i.e., the mechanical angle, and k is a natural number. The topology provided by the present disclosure adopts a phase group centralized winding, and the stator core structure presents a modular phase group feature. Therefore, the number of stator slots in the above formula is modified to the number of phase group modules, i.e., 3n2. In combination with the pole-slot matching formula, the following formula is obtained:

[0065]

[0066] By adjusting the relative angle of the inner rotor and the outer rotor, the peak-to-valley reverse superposition of the torque with a larger amplitude can be achieved, and the maximum inhibition of the cogging torque is realized.

[0067] For the n1=6, n2=2 pole-slot matching double-rotor motor, Fourier analysis of the inner rotor and the outer rotor cogging torque in an electrical period can be performed, and it can be concluded that the inner rotor and the outer rotor cogging torque mainly has a large proportion of 2, 5, 6, and 7th harmonics. Referring to FIG. 11, which is a double-rotor flux switching motor cogging torque spectrum diagram according to some embodiments of the present disclosure.

[0068] In step S50, the suppression harmonics are analyzed to obtain a target suppression harmonic.

[0069] It should be noted that the target suppression harmonic refers to the suppression harmonic that plays a dominant role among the many suppression harmonics. According to the spectrum in FIG. 11, the outer rotor cogging torque mainly has a large proportion of 2, 5, 6, and 7th harmonics, and the 6th harmonic is the most prominent. Therefore, the target suppression harmonic of the embodiments of the present disclosure can be the 6th harmonic.

[0070] In step S60, the target suppression harmonic is simulated and analyzed to obtain an effective shift angle, and the effective shift angle is decoupled to obtain a target shift angle.

[0071] In some embodiments, simulation analysis of target harmonic suppression can obtain that the synthesized cogging torque varies greatly with the double-rotor displacement angle and has periodicity with 180° as the period. When there is no displacement, the phase angles of the harmonics of the inner and outer rotor cogging torques are the same, and the synthesized cogging torque is in a completely positive superposition state and reaches a maximum value. When the displacement angle is 90°, the peak-to-peak value of the synthesized cogging torque reaches a minimum value, and when the displacement angle is 270°, the value is close to the minimum value. This is because the second and sixth harmonics with the largest amplitude and the corresponding harmonics of the outer rotor are in an opposite phase superposition state and are offset. When the displacement angle is 30°, 150°, 270°, or 360°, the sixth harmonics are in a reverse superposition state, and thus the synthesized cogging torque is also small. Therefore, when the displacement angle is 30°, 90°, 150°, 270°, or 360°, the influence of harmonic suppression on the cogging torque can be effectively reduced. However, this is based on the fact that the amplitudes of the harmonics of the two-side cogging torques are equal and remain unchanged with the displacement angle. In fact, the change of the displacement angle will change the magnetic circuit of the motor, thereby affecting the coupling degree of the inner and outer air gap magnetic fields, so that the air gap flux density and the amplitude of the cogging torque change.

[0072] In the embodiments of the present disclosure, the target displacement angle can be obtained by: analyzing the effective displacement angles, determining the magnetic flux harmonic content corresponding to each of the effective displacement angles, obtaining the total harmonic distortion value of the no-load back electromotive force according to the effective displacement angles and the magnetic flux harmonic content, and screening the total harmonic distortion value of the no-load back electromotive force to obtain the target displacement angle.

[0073] When the displacement angle is 0°, the I-type permanent magnet is uniformly utilized by the two-side air gaps, which means that the adjacent two permanent magnets generate almost completely symmetrical and identical magnetic flux loops, which are jointly superimposed by the permanent magnets participating in the magnetic aggregation. At this time, the part of the permanent magnet in the stator yoke region only provides 50% of the effective permanent magnet usage for the air gap on either side.

[0074] When the displacement angle is 180°, the magnetic flux passing through the inner air gap is a series type magnetic flux. The series type magnetic flux simultaneously interlinks the inner and outer windings. In this case, the utilization rate of the permanent magnet is the highest, but the series type magnetic flux passes through four air gap reluctances, so the generated fundamental magnetic flux is not much larger than that when the displacement angle is 90°. Even the magnetic lines of force have a large leakage due to the large magnetic resistance on the path, resulting in an increase in the harmonic content of the magnetic linkage.

[0075] When the shift angle is 90°, the part of the I-type permanent magnet in the stator yoke area will act on the inner air gap, and at the same time, the outer part of the permanent magnet yoke area will also participate in providing effective magnetic flux. According to the minimum reluctance principle, increasing the outer rotor tooth height to a certain extent can increase the magnetic reluctance of the leakage magnetic path through the outer part of the I-type permanent magnet, so that more permanent magnets tend to act on the other side air gap. It can be seen that the middle part of the permanent magnet has the effect of "time division multiplexing", that is, at the moment of the maximum flux linkage of each side winding, effective magnetic flux can be provided to form an alternating magnetic aggregation circuit to improve the torque density of the motor. Therefore, when the shift angle is 90°, the main magnetic flux increases by a type of effective magnetic flux compared with 0°, and the magnetic reluctance of the yoke part of the permanent magnet is also reduced, resulting in a significant increase in the synthesized main magnetic flux compared with the non-shifted state. In addition, for the relationship between the shift angle and the total harmonic distortion (THD) of the no-load back electromotive force, it can be found that the THD value of the no-load back electromotive force is the smallest when the shift angle is 90° or 270°.

[0076] In order to evaluate the disturbance effect of the armature magnetic motive force of the outer winding on the inner winding, the back electromotive force coupling rate can be calculated, the no-load back electromotive force of the inner winding and the coupled induced back electromotive force of the inner winding can be determined according to the total harmonic distortion of the no-load back electromotive force, the back electromotive force coupling rate can be obtained according to the no-load back electromotive force of the inner winding and the coupled induced back electromotive force of the inner winding, the magnetic flux density distribution value of the inner rotor can be obtained according to the armature magnetic motive force of the outer winding, the target shift angle can be obtained according to the back electromotive force coupling rate and the magnetic flux density distribution value of the inner rotor. The back electromotive force coupling rate k co can be expressed as:

[0077]

[0078] In the above formula, (EMF i ) is the no-load back electromotive force of the inner winding, (EMF w ) is the coupled induced back electromotive force of the inner winding, and k β is the maximum point of the average value of the magnetic flux density distribution of the inner rotor core.

[0079] k β = max (avg. (B i ))

[0080] In the above formula, B i is the unit magnetic density of the inner rotor core in the electrical period, and k βThe smaller the contribution of the armature magnetomotive force of the outer winding to the magnetic flux density of the inner rotor core is, the more the armature magnetomotive force acts on the outer motor part, and the higher the utilization rate of the armature magnetomotive force is. Referring to FIG. 12, which is a schematic diagram of the inner-outer coupling degree and the shift angle of a double-rotor motor according to some embodiments of the present disclosure. When the back electromotive force coupling rate is less than 180°, it decreases as the shift angle increases, and then increases again when it exceeds 180°. Therefore, when the shift angle is 180°, the armature magnetomotive force of the outer winding has the least interference with the inner winding, and when the shift angle is 90° or 270°, the coupling rate is only 1.94% and 1.99%, which is a low coupling degree. Although the coupling rates of the shift angles of 90° and 270° are roughly the same, due to the influence of the complementary magnetic circuit of the flux switching motor, the k β value of 270° is much larger than that of 90°. For a double-rotor flux switching permanent magnet motor, when the shift angle is set to 90°, the best ideal effect can be achieved, that is, not only the lowest cogging torque can be met (the 6th harmonic content is reduced), but also an alternating magnetic aggregation circuit can be formed to avoid the mutual interference of the inner and outer rotor magnetic fluxes, and the maximum back electromotive force is achieved to ensure a low coupling degree between the double three-phase windings.

[0081] In step S70, the double-rotor flux switching permanent magnet motor is optimized according to the target shift angle.

[0082] In some embodiments, after obtaining the target shift angle, the shift angle of the double-rotor flux switching permanent magnet motor is set to 90° to realize magnetic flux alternation aggregation, reduce the cogging torque, and increase the back electromotive force fundamental wave content, thereby ensuring weak coupling between the double three-phase windings and the double air gaps, and achieving the purpose of optimizing the double-rotor flux switching permanent magnet motor.

[0083] According to the number of phase group modules, the power grid harmonic is obtained, the inner torque of the inner rotor and the outer torque of the outer rotor are determined, the motor fundamental wave frequency is obtained according to the rotor pole number and the rotating speed, the electrical period is obtained according to the motor fundamental wave frequency, the power grid harmonic is subjected to Fourier analysis in the electrical period of the inner torque and the outer torque to obtain a suppression harmonic, the target suppression harmonic is obtained by analyzing the suppression harmonic, the effective shift angle is obtained by simulating and analyzing the target suppression harmonic, the target shift angle is obtained by decoupling the effective shift angle, the double-rotor flux switching permanent magnet motor is optimized according to the target shift angle, the inner and outer motors are combined, the integration degree of the double motor is improved, and the magnetic flux coupling rate of the double motor is also improved.

[0084] Based on the same inventive concept, the embodiment of the disclosure further provides a double-rotor flux-switching permanent magnet motor optimization device, applied to the double-rotor flux-switching permanent magnet motor as described above, comprising: a first determination module configured to determine the number of phase group modules of the double-rotor flux-switching permanent magnet motor, and obtain power grid harmonics according to the number of phase group modules; a second determination module configured to determine the inner torque of the inner rotor and the outer torque of the outer rotor; an electrical period acquisition module configured to obtain the motor fundamental frequency according to the rotor pole number and the rotating speed, and obtain the electrical period according to the motor fundamental frequency; a first analysis module configured to perform Fourier analysis on the electrical period of the power grid harmonics in the inner torque and the outer torque, and obtain suppression harmonics; a second analysis module configured to analyze the suppression harmonics, and obtain target suppression harmonics; a third analysis module configured to perform simulation analysis on the target suppression harmonics, obtain an effective shift angle, and decouple the effective shift angle to obtain a target shift angle; and a processing module configured to optimize the double-rotor flux-switching permanent magnet motor according to the target shift angle.

[0085] In some embodiments, the second determination module is further configured to: determine the number of I-type permanent magnets and the number of phase groups; obtain a mechanical angle according to the electrical angle and the rotor pole number; obtain the inner torque of the inner rotor according to the mechanical angle and the rotor pole number; and obtain the outer torque of the outer rotor according to the mechanical angle, the rotor pole number and the phase difference.

[0086] In some embodiments, the double-rotor flux-switching permanent magnet motor optimization device further comprises a rotor pole number acquisition module configured to obtain the rotor pole number according to the number of I-type permanent magnets and the number of phase groups.

[0087] In some embodiments, the first analysis module is further configured to: determine the harmonic component of the power grid harmonics; obtain the greatest common divisor of the stator slot number and the rotor pole number according to the stator slot number and the rotor pole number; obtain the number of times of the harmonic component according to the stator slot number and the greatest common divisor of the stator slot number and the rotor pole number; obtain the motor cogging torque according to the harmonic component, the number of times of the harmonic component, the rotor pole number and the mechanical angle; and obtain the suppression harmonics according to the motor cogging torque.

[0088] In some embodiments, the third analysis module is further configured to: analyze the effective shift angle to determine the flux harmonic content corresponding to each effective shift angle; obtain the total harmonic distortion value of the no-load back electromotive force according to the effective shift angle and the flux harmonic content; and screen the total harmonic distortion value of the no-load back electromotive force to obtain the target shift angle.

[0089] In some embodiments, the third analysis module is further configured to: determine an inner winding no-load back EMF and an inner winding coupled induced back EMF according to the total harmonic distortion of the no-load back EMF; obtain a back EMF coupling ratio according to the inner winding no-load back EMF and the inner winding coupled induced back EMF; determine a magnetic flux density distribution value of the inner rotor according to the total harmonic distortion of the no-load back EMF; and obtain a target shift angle according to the back EMF coupling ratio and the magnetic flux density distribution value of the inner rotor.

[0090] In some embodiments, the third analysis module is further configured to: determine a cell magnetic flux of the inner rotor in the electric period according to the total harmonic distortion of the no-load back EMF; and obtain an armature magnetic motive force of the outer winding according to the cell magnetic flux of the inner rotor to obtain the magnetic flux density distribution value of the inner rotor.

[0091] Based on the same inventive concept, the disclosure further provides a computer readable storage medium having a computer program stored thereon, the program being executed by a processor to implement the steps of the above-mentioned double-rotor flux-switching permanent magnet motor optimization method.

[0092] Based on the same inventive concept, the computer program product comprises a computer program, which, when executed by a processor, is configured to load and execute the steps of the above-mentioned double-rotor flux-switching permanent magnet motor optimization method.

[0093] It should be understood that the above is only an example and does not constitute any limitation on the technical solutions of the disclosure. In specific applications, those skilled in the art can make settings according to needs, and the disclosure does not limit this.

[0094] It should be understood that, although the steps in the flowchart in the embodiments of the disclosure are displayed in sequence according to the arrows, these steps are not necessarily executed in sequence according to the arrows. Unless otherwise specified herein, the execution of these steps has no strict sequence limitation, and they can be executed in other orders. Moreover, at least part of the steps in the figure can include multiple sub-steps or multiple stages, which are not necessarily executed at the same time, but can be executed at different times, and the execution order is not necessarily sequential, but can be alternately executed with other steps or sub-steps or stages of other steps.

[0095] It should be noted that the above-described workflow is only illustrative and does not limit the scope of protection of the disclosure. In actual applications, those skilled in the art can select part or all of them to achieve the purpose of the embodiments of the disclosure according to actual needs, which is not limited here.

[0096] It should be noted that in the present disclosure, the terms "comprising", "containing" or any other variants thereof are intended to cover non-exclusive inclusion, so that a process, method, article or system including a series of elements not only includes those elements, but also includes other elements not explicitly listed or inherent to such a process, method, article or system. Without more limitations, the element defined by the statement "comprising a" does not exclude the presence of other identical elements in the process, method, article or system including the element.

[0097] The above sequence numbers of the embodiments of the present disclosure are only for description, and do not represent the advantages and disadvantages of the embodiments.

[0098] From the above description of the embodiments, those skilled in the art can clearly understand that the above-mentioned embodiment methods can be realized by means of software and necessary general hardware platforms, and of course can also be realized by hardware. Based on such understanding, the technical solutions of the present disclosure can be embodied in the form of a software product, which is stored in a storage medium (such as a read only memory (ROM) / RAM, a magnetic disk, an optical disk), and includes a plurality of instructions for making a terminal device (which can be a mobile phone, a computer, a server, or a network device, etc.) execute the methods described in various embodiments of the present disclosure.

[0099] The above is only the preferred embodiment of the present disclosure, and does not limit the patent scope of the present disclosure, and any equivalent structure or equivalent flow transformation made by using the content of the present disclosure specification and drawings, or directly or indirectly applied to other related technical fields, are also included in the patent protection scope of the present disclosure.

Claims

1. A dual-rotor flux switching permanent magnet motor, comprising an inner rotor, an outer rotor and an intermediate stator, wherein the intermediate stator comprises a stator core, permanent magnets are disposed on both sides of the stator core, the stator core comprises a plurality of teeth, and an excitation winding is disposed between adjacent teeth.

2. The dual-rotor flux switching permanent magnet motor according to claim 1, wherein: The electrical angle of the centerline arc width of adjacent teeth is greater than or equal to 0 and less than or equal to π.

3. The dual-rotor flux switching permanent magnet motor according to claim 2, wherein: The electrical angle of the centerline arc width of the adjacent teeth is π.

4. The inner rotor and the outer rotor according to any one of claims 1 to 3 are fixedly connected to the same motor shaft through mechanical tooling, and the intermediate stator is fixed to the motor housing.

5. The dual-rotor flux switching permanent magnet motor according to any one of claims 1 to 3, wherein: The electronic core of the intermediate stator is an E-type stator core, and the permanent magnet is an I-type permanent magnet.

6. A method for optimizing a dual-rotor flux-switching permanent magnet motor, applied to the dual-rotor flux-switching permanent magnet motor according to any one of claims 1 to 5, comprising: Determining the number of phase group modules of the dual-rotor flux-switching permanent magnet motor, and obtaining grid harmonics according to the number of phase group modules; Determine the internal torque of the inner rotor and the external torque of the outer rotor; Obtaining a motor fundamental frequency according to the number of rotor poles and the rotational speed, and obtaining an electrical period according to the motor fundamental frequency; Performing Fourier analysis on the grid harmonics during the electrical cycles of the internal torque and the external torque to obtain suppressed harmonics; Analyzing the suppressed harmonics to obtain target suppressed harmonics; Performing simulation analysis on the target suppressed harmonics to obtain an effective shift angle, and decoupling the effective shift angle to obtain a target shift angle; as well as The dual-rotor flux-switching permanent magnet motor is optimized according to the target shift angle.

7. The method according to claim 6, wherein: Determining the internal torque of the inner rotor and the external torque of the outer rotor includes: Determine the number of I-type permanent magnets and the number of phase groups; Obtain the mechanical angle based on the electrical angle and the number of rotor poles; Obtaining the internal torque of the inner rotor according to the mechanical angle and the number of rotor poles; and The external torque of the outer rotor is obtained according to the mechanical angle, the number of rotor poles and the phase difference.

8. The method of claim 6, further comprising: The number of rotor poles is obtained according to the number of the I-type permanent magnets and the number of phase groups.

9. The method of claim 6, wherein: The performing Fourier analysis on the grid harmonics during the electrical cycles of the internal torque and the external torque to obtain suppressed harmonics includes: determining harmonic components of the grid harmonics; Obtaining a greatest common divisor of the number of stator slots and the number of rotor poles according to the number of stator slots and the number of rotor poles; Obtaining the order of the harmonic component according to the number of stator slots and the greatest common divisor of the number of stator slots and the number of rotor poles; Obtaining the motor cogging torque according to the harmonic component, the order of the harmonic component, the number of rotor poles and the mechanical angle; and Harmonics are suppressed based on the motor cogging torque.

10. The method according to claim 6, wherein: Decoupling the effective shift angle to obtain a target shift angle includes: Analyzing the effective shift angles to determine the magnetic flux harmonic content corresponding to each effective shift angle; Obtaining a no-load back electromotive force total harmonic distortion value according to the effective shift angle and the magnetic flux harmonic content; and The total harmonic distortion value of the no-load back electromotive force is screened to obtain a target shift angle.

11. The method according to claim 10, wherein: The screening of the total harmonic distortion value of the no-load back electromotive force to obtain a target shift angle includes: Determining the inner winding no-load back electromotive force and the inner winding coupled induced back electromotive force according to the no-load back electromotive force total harmonic distortion value; Obtaining a back electromotive force according to the no-load back electromotive force of the inner winding and the coupled induced back electromotive force of the inner winding; Determining the magnetic flux density distribution value of the inner rotor according to the no-load back electromotive force total harmonic distortion value; and A target shift angle is obtained according to the back electromotive coupling ratio and the magnetic flux density distribution value of the inner rotor.

12. The method of claim 11, wherein: Determining the magnetic flux density distribution value of the inner rotor according to the total harmonic distortion value of the no-load back electromotive force includes: determining the unit magnetic flux density of the inner rotor within the electrical cycle according to the total harmonic distortion value of the no-load back electromotive force; and The armature magnetomotive force of the outer winding is obtained according to the unit magnetic density of the inner rotor, and the magnetic flux density distribution value of the inner rotor is obtained.

13. A dual-rotor flux switching permanent magnet motor optimization device, applied to the dual-rotor flux switching permanent magnet motor according to any one of claims 1 to 5, comprising: a first determining module, configured to determine the number of phase group modules of the dual-rotor flux switching permanent magnet motor, and obtain grid harmonics according to the number of phase group modules; a second determining module, configured to determine an internal torque of the inner rotor and an external torque of the outer rotor; An electrical cycle acquisition module is used to obtain the motor fundamental frequency according to the number of rotor poles and the rotational speed, and obtain the electrical cycle according to the motor fundamental frequency; A first analysis module is configured to perform Fourier analysis on the grid harmonics during the electrical cycles of the internal torque and the external torque to obtain suppressed harmonics; A second analysis module is used to analyze the suppressed harmonics to obtain target suppressed harmonics; a third analysis module, configured to perform simulation analysis on the target suppressed harmonic to obtain an effective shift angle, and decouple the effective shift angle to obtain a target shift angle; as well as A processing module is used to optimize the dual-rotor flux switching permanent magnet motor according to the target shift angle.

14. The apparatus of claim 13, wherein: The second determining module is further configured to: Determine the number of I-type permanent magnets and the number of phase groups; Obtain the mechanical angle based on the electrical angle and the number of rotor poles; Obtaining the internal torque of the inner rotor according to the mechanical angle and the number of rotor poles; as well as The external torque of the outer rotor is obtained according to the mechanical angle, the number of rotor poles and the phase difference.

15. The apparatus of claim 13, further comprising: The rotor pole number acquisition module is used to obtain the rotor pole number according to the number of the I-type permanent magnets and the number of phase groups.

16. The apparatus of claim 13, wherein: The first analysis module is further configured to: determining harmonic components of the grid harmonics; Obtaining a greatest common divisor of the number of stator slots and the number of rotor poles according to the number of stator slots and the number of rotor poles; Obtaining the order of the harmonic component according to the number of stator slots and the greatest common divisor of the number of stator slots and the number of rotor poles; Obtaining the motor cogging torque according to the harmonic component, the order of the harmonic component, the number of rotor poles and the mechanical angle; and Harmonics are suppressed based on the motor cogging torque.

17. The apparatus of claim 13, wherein: The third analysis module is further configured to: Analyzing the effective shift angles to determine the magnetic flux harmonic content corresponding to each effective shift angle; Obtaining a no-load back electromotive force total harmonic distortion value according to the effective shift angle and the magnetic flux harmonic content; and The total harmonic distortion value of the no-load back electromotive force is screened to obtain a target shift angle.

18. The apparatus of claim 17, wherein: The third analysis module is further configured to: Determining the inner winding no-load back electromotive force and the inner winding coupled induced back electromotive force according to the no-load back electromotive force total harmonic distortion value; Obtaining a back electromotive force according to the no-load back electromotive force of the inner winding and the coupled induced back electromotive force of the inner winding; Determining the magnetic flux density distribution value of the inner rotor according to the no-load back electromotive force total harmonic distortion value; and A target shift angle is obtained according to the back electromotive coupling ratio and the magnetic flux density distribution value of the inner rotor.

19. A computer-readable storage medium comprising a computer program stored thereon, wherein when the computer program is executed by a processor, the steps of the method according to any one of claims 6 to 12 are implemented.

20. A computer program product, comprising a computer program, wherein when the computer program is executed by a processor, the computer program is configured to load and execute the method according to any one of claims 6 to 12.

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