A parameter synthesis design method for interleaved parallel boost converters

Abstract

The present invention discloses a parameter comprehensive design method for an interleaved parallel Boost converter, comprising the following steps: (1) determining the relationship between parameter values ​​and ripple requirements; (2) constructing a volume model for passive components, including input capacitor volume modeling, output capacitor volume modeling and inductor volume modeling; (3) constructing a loss model, wherein the loss of the converter includes switch tube loss and inductor loss, and the switch tube loss is related to input and output voltages, switching frequency, power and current ripple; the inductor loss includes copper loss and iron loss; (4) comprehensive optimization of the converter parameters; the loss and volume of the converter are ternary functions, and the corresponding loss and volume results are calculated within the respective value ranges of three independent variables; in combination with the converter efficiency requirements, the volume of the passive components and the minimum independent variable value result are selected from the independent variable values ​​whose loss does not exceed the limit value, and the corresponding inductor and capacitor values ​​are calculated.

Description

Technical Field

[0001] The invention belongs to the field of switch power supply design, and in particular relates to a parameter design method of an interleaved parallel Boost converter taking into account both efficiency and power density. Background Art

[0002] The interleaved parallel Boost converter has the advantages of simple structure, small ripple, and high energy conversion efficiency, and has been widely used in fuel cells, photovoltaics and other fields. In the traditional interleaved parallel converter parameter design, only the passive component parameter selection is considered to meet the ripple requirements, and the basis for selecting the switching frequency cannot be given, so the optimal parameter design scheme that takes into account efficiency and power density cannot be obtained theoretically. Therefore, it is of great engineering value to propose a comprehensive converter parameter design method that takes into account efficiency and power density. Summary of the invention

[0003] The purpose of the present invention is to overcome the deficiencies in the prior art, solve the problem that the efficiency and power density cannot be taken into account in the design process of the staggered parallel Boost converter, and provide a parameter comprehensive design method for the staggered parallel Boost converter.

[0004] The objective of the present invention is achieved through the following technical solutions:

[0005] A parameter comprehensive design method for an interleaved parallel Boost converter comprises the following steps:

[0006] (1) Determine the relationship between parameter values ​​and ripple requirements;

[0007] For the interleaved parallel converter, the total ripple during interleaved operation does not exceed the ripple when the branch is operated independently; if the inductor current ripple meets the requirement when the branch is operated independently, the converter as a whole meets the ripple requirement; thus, the relationship between the inductance, input and output capacitance values ​​and the converter ripple requirement is obtained;

[0008] (2) Constructing volume models for passive components, including input capacitor volume modeling, output capacitor volume modeling, and inductor volume modeling;

[0009] (3) Construct a loss model. The converter loss includes switch loss and inductor loss. The switch loss is related to the input and output voltage, switching frequency, power, and current ripple. The inductor loss includes copper loss and iron loss.

[0010] (4) Comprehensive optimization of converter parameters; the converter loss and volume are three-variable functions. The corresponding loss and volume results are calculated within the respective value ranges of the three independent variables. In combination with the converter efficiency requirements, the volume of passive components and the smallest independent variable value result are selected from the independent variable values ​​whose loss does not exceed the limit, and the corresponding inductance and capacitance values ​​are calculated.

[0011] Furthermore,

[0012] In step (1), for the determined electrical parameters and ripple requirements, based on the N-way staggered parallel Boost converter, in the mode of continuous inductor current, the parameter values ​​of the passive components are about the inductor current ripple ΔI L and the switching frequency f s Function of

[0013]

[0014]

[0015]

[0016] Where L is the inductance value, C o is the output capacitor value, C in is the value of the input capacitance, λ u is the output voltage ripple rate, U i and U o is the input and output voltage, P is the rated power, ΔU i is the input voltage ripple, and N is the number of interleaved parallel branches.

[0017] Furthermore,

[0018] In step (2):

[0019] (201) The input capacitor volume modeling and output capacitor volume modeling are as follows

[0020] The relationship between the value and volume of the same series of capacitors is a discrete point, showing a linear relationship. The capacitor volume model is built by fitting; the input capacitance V Cin and output capacitor volume V Co The volume fitting result is

[0021]

[0022] In the formula, a 1 , b 1 、a 2 , b 2 is the fitting coefficient; C o is the output capacitor value, C in Select the value for the input capacitor.

[0023] Combining equations (2) to (4), the input capacitor volume V Cin and output capacitor volume V Co is about the switching frequency f s and the inductor current ripple ΔI L The function of

[0024]

[0025] (202) Inductor volume modeling; for a given core shape, the size of the inductor depends on the winding size (length and width are denoted by d w and h w ), winding turns n w , core cross-sectional area A e And the core window k, that is

[0026] V L =V L (n w ,A e ,d w ,h w ,k) (6)

[0027] Similarly, the core volume V Fe It can be expressed as

[0028] V Fe =V Fe (n w ,A e ,d w ,h w ,k) (7)

[0029] Average magnetic circuit l of the inductor core e It is related to the inductor size and can be expressed as

[0030] l e = l e (n w ,A e ,d w ,h w ,k) (8)

[0031] For a given winding size and core shape, d w 、h w And the core opening k is a constant. At this time, the inductor volume V L and magnetic circuit e All are related to the number of winding turns n w and the core cross-sectional area A e The function of

[0032]

[0033] Considering the theoretical calculation formula of inductance L is

[0034]

[0035] In the formula, μ r is the relative magnetic permeability; μ0 is the vacuum magnetic permeability, which is 0.4π×10 -8 H / m; l e is the equivalent magnetic circuit; α is the magnetic permeability attenuation coefficient, which is related to the magnetic field change frequency and the magnetic field intensity H and is taken as a constant.

[0036] Combining equations (1), (8) and (10), we get the number of turns n w The functional relationship is

[0037] n w =n w [L(ΔI L ,f s ),A e ] (11)

[0038] Combining equations (6) to (11), we can see that the inductor volume V L and the core volume V Fe It is about the core cross-sectional area A e , switching frequency f s And the inductor current ripple ΔI L The ternary function of

[0039]

[0040] It can be obtained that the volume of passive components and V t Expressed as

[0041] V t (A e ,ΔI L ,f s )=N·V L (A e ,ΔI L ,f s )+V Cin (f s ,ΔI L )+V Co (f s ) (13)

[0042] Furthermore, in step (3), in order to analyze the loss of the converter, it is first necessary to determine the current peak and valley values ​​I during the switching process. max and I min . Assume that the effective value of the inductor current is I rms , whose value is the input current I in 1 / N of.

[0043]

[0044] In high-power converters, MOS are usually connected in parallel to increase the current carrying capacity. Let the number of parallel MOS be n.d , its value is related to the number of branches N. For an N-branch parallel Boost converter, the power device loss of a single branch is

[0045]

[0046] Where P con is the conduction loss, P on and P off is the turn-on and turn-off loss, P oss is the output capacitor loss. R ds is the MOS on-state resistance, Q gs is the gate-source charge, Q gd is the gate-drain charge, V th is the threshold voltage, V mp is the Miller voltage, R on To turn on the drive resistor, R off is the turn-off drive resistor, C oss is the equivalent output capacitance.

[0047] Under the maximum voltage and current stress, the switch loss P sw is the current ripple ΔI L and the switching frequency f s Function

[0048] P sw =P sw (ΔI L ,f s ) (16)

[0049] Inductor loss P L Including copper loss P cu and iron loss P Fe Two parts. For a certain core shape, after the inductance is determined, the core cross-sectional area A e is a variable, then the number of winding turns n w It can be solved. For copper loss P cu , for a given winding width and thickness d w and h w , under the maximum current stress, the copper loss P cu is about the current ripple ΔI L , switching frequency f s and the core cross-sectional area A e The function of

[0050] P cu =P cu {[L(ΔI L ,f s ),A e ],A e =Pcu (ΔI L ,f s ,A e ) (17)

[0051] Iron loss P Fe The inductor core material and core volume V Fe , switching frequency f s And the current ripple ΔI L For a certain core material, the iron loss calculation formula is

[0052] P Fe =P Fe [V Fe (A e ,ΔI L ,f s ),ΔI L ,f s ]=P Fe (A e ,ΔI L ,f s ) (18)

[0053] From equations (16) to (18), we can see that the converter loss P loss Expressed as

[0054] P loss (ΔI L ,f s ,A e )=P sw (ΔI L ,f s )+P cu (ΔI L ,f s ,A e )+P Fe (ΔI L ,f s ,A e ) (19)

[0055] It can be obtained that the converter efficiency η can be expressed as

[0056]

[0057] Equations (13) and (20) establish a comprehensive model of converter efficiency and passive power component volume. The model includes three independent variables, namely, current ripple ΔI L , switching frequency f s and the core cross-sectional area A e Within the range of independent variable transformation, the converter efficiency and power density can be optimized through iterative calculation.

[0058] Compared with the prior art, the technical solution of the present invention has the following beneficial effects:

[0059] 1. A comprehensive converter efficiency-volume model was established, and the theoretical optimal solution for parameter design was given.

[0060] In traditional parameter methods, the parameter values ​​of staggered parallel converters often depend on the experience of R&D personnel, so the quality of parameter design results is difficult to evaluate. The technical solution of the present invention establishes a converter volume-efficiency comprehensive model, which contains the same independent variable parameters and can simultaneously evaluate the efficiency and power density of the converter. By iteratively calculating the independent variables, the optimal design result that meets the converter efficiency and power density requirements can be obtained, achieving dual optimization of efficiency-power density.

[0061] 2. During the parameter design process, not only the electrical characteristics of the converter are considered, but also the impact of the parameter values ​​on the volume.

[0062] Traditional parameter design methods mainly consider whether ripple and efficiency requirements are met, but do not quantitatively consider the volume of the converter. In the technical solution of the present invention, after determining the inductance and capacitance values, the influence of the core radius and capacitor selection on the inductance volume and the capacitance volume is also quantitatively analyzed, thereby realizing the optimized design of converter parameters.

[0063] 3. The technical solution of the present invention has good feasibility and universality.

[0064] The volume of the converter is affected by many factors such as passive power devices, heat sinks, and device layout, so it is difficult to quantify comprehensively. In high-power converters, the volume of passive power devices accounts for a relatively high proportion; in addition, as a heat source, the design of the heat sink needs to match the size of the passive power devices. Therefore, using the volume of passive power devices as an indicator to evaluate the power density of the converter makes the technical solution of the present invention have good feasibility and is suitable for the parameter design of other types of converters. BRIEF DESCRIPTION OF THE DRAWINGS

[0065] Attached Figure 1 It is a flow chart of parameter comprehensive design method.

[0066] Attached Figure 2 It is the topology diagram of N-way interleaved parallel Boost converter.

[0067] Attached Figure 3 It is the fitting relationship between the input and output capacitor values ​​and volumes.

[0068] Attached Figure 4 Schematic diagram of the shape of the inductor core in this embodiment.

[0069] Attached Figure 5Schematic diagram of the inductor core size in this embodiment.

[0070] Attached Figure 6 It is the result of the independent variable value that meets the efficiency requirement in this embodiment.

[0071] Attached Figure 7 It is the corresponding volume design result under the independent variable value that meets the efficiency requirement in this embodiment.

[0072] Appendix 1 shows the converter design parameters in this embodiment. DETAILED DESCRIPTION

[0073] The present invention will be further described in detail below in conjunction with the accompanying drawings and specific embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not used to limit the present invention.

[0074] The topology of the interleaved parallel Boost converter used in this embodiment is shown in the attached figure. Figure 2 As shown, the number of branches N is 6. It should be noted that the value of N can be any value. Appendix 1 shows the design parameters of the converter.

[0075] Based on the attached Figure 2 The loss and volume of the converter are included in the same independent variable system as in Appendix 1, and parameter comprehensive design is achieved through parameter optimization method. Since the volume of passive components accounts for the main aspect of the converter volume, in this embodiment, the volume of passive components is approximated as the volume of the converter.

[0076] Step 1: Determine the relationship between parameter values ​​and ripple requirements

[0077] For interleaved parallel converters, the total ripple during interleaved operation does not exceed the ripple when the branch is operating alone. If the ripple when the branch is operating alone does not exceed the ripple requirement, the converter as a whole meets the ripple requirement; thus, the relationship between the values ​​of inductance, input and output capacitance and the converter ripple requirement is as follows. Combined with the voltage range and number of branches in Appendix 1, it can be seen that the parameter values ​​of passive components are about the inductor current ripple ΔI L and the switching frequency f s function.

[0078]

[0079]

[0080]

[0081] Step 2: Passive component volume modeling.

[0082] 1) Input and output capacitor volume modeling

[0083] The relationship between the value and volume of the same series of capacitors is a discrete point, showing a linear relationship, and the capacitor volume modeling is performed by fitting. In this embodiment, the input side capacitor is a Vishay 600V DC-link film capacitor, and the output capacitor is a Vishay 1000V DC-link film capacitor. It can be obtained that the relationship between the input and output capacitor values ​​and the volume is as shown in the attached figure. Figure 3 As shown, the fitting relationship is

[0084]

[0085] 2) Inductor volume modeling

[0086] In this embodiment, the shape of the inductor is as shown in the attached figure. Figure 4 and Figure 5 As shown, the core opening is 50%. When the number of branches N is 6, the current stress of each branch is 50A, so the thickness of the inductor winding h w Take it as 0.2cm, winding width d w is 0.4cm, winding spacing d 0 0.08cm. Core cross-sectional area A e Substituting the core radius r, we can get the inductor volume V L for

[0087] V L (r,n w )=(0.24+4r)·(2r+0.8)·0.14n w (25)

[0088] Average magnetic circuit l e for

[0089] l e (r,n w )=0.32+0.28n w +(π+4)r (26)

[0090] The calculation formula for inductance L is:

[0091]

[0092] Wherein, α is the magnetic permeability attenuation coefficient, which is taken as 0.8 in this embodiment.

[0093] Combining equations (26) and (27), the number of turns n can be obtained: w The calculation formula is

[0094]

[0095] In the formula, σ can be expressed as

[0096]

[0097] Combined with formula (21), the inductance L is the current ripple ΔI L and the switching frequency f s It can be obtained that the inductor volume V L and the core volume V Fe The functional relationship is

[0098]

[0099] It can be obtained that the volume of passive components and V t for

[0100] V t (r,ΔI L ,f s )=6V L (r,ΔI L ,f s )+V Cin (f s ,ΔI L )+V Co (f s ) (30)

[0101] Step 3: Loss Modeling

[0102] In this embodiment, the MOS tube model is Infineon IMZ120R045M1. When the number of branches N is 6, the number of parallel MOS is 2. The effective value of the inductor current I rms is 50A, combined with the current ripple ΔI L It can be seen that the peak and valley values ​​of the inductor current I max and I min for

[0103]

[0104] The on and off resistances are both 3Ω. Combining the data sheet and Appendix 1, we can see that the loss calculation formula is:

[0105]

[0106] It can be obtained that the power device loss P sw for

[0107]

[0108] Inductor loss P L Including copper loss P cu and iron loss P Fe For the winding selected above, the copper loss P cu The expression is

[0109]

[0110] In this example, the inductor core material is POCO NPH-L 60μ, and the iron loss P Fe for

[0111]

[0112] Where, magnetic swing ΔB m The calculation formula is

[0113]

[0114] From equations (28) to (35), we can see that the converter loss P loss Expressed as

[0115] P loss (ΔI L ,f s ,r)=P sw (ΔI L ,f s )+P cu (ΔI L ,f s ,r)+P Fe (ΔI L ,f s ,r) (36)

[0116] It can be obtained that for the 6-branch interleaved parallel Boost converter, its efficiency η can be expressed as

[0117]

[0118] Equations (30) and (37) establish a comprehensive model of converter efficiency and passive power component volume. The model includes three independent variables, namely, current ripple ΔI L , switching frequency f s and the core radius r.

[0119] Table 1

[0120]

[0121] Step 4: Comprehensive optimization of converter parameters

[0122] Based on the attached Figure 1 The comprehensive model established in step 3 has the range of independent variables as f s ∈[20,200kHz], r∈[0,3cm], ΔI L∈[0,87A]. It should be noted that, since the inductor current is in CCM mode, based on Appendix 1, the effective value of the inductor current in each branch is 50A, and the maximum value of the current ripple is the current ripple in CRM mode, that is, 87A.

[0123] As can be seen from Table 1, the design goal in this embodiment is to achieve the optimal power density under the efficiency requirement. Based on the comprehensive model, iterative calculations are performed within the range of independent variables to obtain the values ​​of all independent variables that meet the efficiency requirement as shown in Table 1. Figure 6 As shown. In the independent variable values ​​that meet the efficiency requirements, the corresponding sum of the passive power device volumes is as follows Figure 7 As shown. Figure 6 and attached Figure 7 It is known that among all the design results that meet the loss requirements, when the switching frequency f s =60kHz, current ripple ΔI L =7A, and when the core radius r is 1.2cm, the volume and size of the passive power device are the smallest.

[0124] The present invention is not limited to the embodiments described above. The above description of the specific embodiments is intended to describe and illustrate the technical solution of the present invention. The above specific embodiments are merely illustrative and not restrictive. Without departing from the scope of the present invention and the scope of protection of the claims, a person of ordinary skill in the art can also make many forms of specific changes under the guidance of the present invention, which all fall within the scope of protection of the present invention.

Claims

1. A parameter comprehensive design method for interleaved parallel Boost converter, characterized in that: The following steps are involved: (1) Determine the relationship between parameter values ​​and ripple requirements; For the interleaved parallel converter, the total ripple during interleaved operation does not exceed the ripple when the branch is operated independently; if the inductor current ripple meets the requirement when the branch is operated independently, the converter as a whole meets the ripple requirement; thus, the relationship between the inductance, input and output capacitance values ​​and the converter ripple requirement is obtained; For the determined electrical parameters and ripple requirements, based on the N-way staggered parallel Boost converter, in the mode of continuous inductor current, the parameter values ​​of the passive components are about the inductor current ripple ΔI L and the switching frequency f s Function of Where L is the inductance value, C o is the output capacitor value, C in is the value of the input capacitance, λ u is the output voltage ripple rate, U i and U o is the input and output voltage, P is the rated power, ΔU i is the input voltage ripple, N is the number of interleaved parallel branches; (2) Constructing volume models for passive components, including input capacitor volume modeling, output capacitor volume modeling, and inductor volume modeling; (201) The input capacitor volume modeling and output capacitor volume modeling are as follows The relationship between the value and volume of the same series of capacitors is a discrete point, showing a linear relationship. The capacitor volume model is built by fitting; the input capacitance V Cin and output capacitor volume V Co The volume fitting result is In the formula, a1, b1, a2, b2 are fitting coefficients; C o is the output capacitor value, C in is the value of the input capacitance; N is the number of interleaved parallel branches; f s is the switching frequency, ΔI L is the inductor current ripple; Combining equations (2) to (4), the input capacitor volume V Cin and output capacitor volume V Co is about the switching frequency f s and the inductor current ripple ΔI L The function of (202) Inductor volume modeling; for a given core shape, the size of the inductor depends on the winding size, where the length and width of the winding are denoted by d w and h w 、Number of winding turns n w , core cross-sectional area A e And the core window k, that is V L =V L (n w ,A e ,d w ,h w ,k) (6) Similarly, the core volume V Fe Expressed as V Fe =V Fe (n w ,A e ,d w ,h w ,k) (7) Average magnetic circuit l of the inductor core e It is related to the inductor size and is expressed as l e =l e (n w ,A e ,d w ,h w ,k) (8) For a given winding size and core shape, d w 、h w And the core opening k is a constant. At this time, the core volume V Fe 、Inductor volume V L and magnetic circuit e All are related to the number of winding turns n w and the core cross-sectional area A e The function of Considering the theoretical calculation formula of inductance L is In the formula, μ r is the relative magnetic permeability; μ0 is the vacuum magnetic permeability, which is 0.4π×10 -8 H / m; α is the magnetic permeability attenuation coefficient, which is related to the magnetic field change frequency and the magnetic field intensity H and is taken as a constant; Combining equations (1), (8) and (10), we can get the number of turns n w The functional relationship is n w =n w [L(ΔI L ,f s ),A e ] (11) Combining equations (6) to (11), we can see that the inductor volume V L and the core volume V Fe It is about the core cross-sectional area A e , switching frequency f s And the inductor current ripple ΔI L The ternary function of It can be obtained that the volume of passive components and V t Expressed as V t (A e ,ΔI L ,f s )=N·V L (A e ,ΔI L ,f s )+V Cin (f s ,ΔI L )+V Co (f s ) (13) (3) Construct a loss model. The converter loss includes switch loss and inductor loss. The switch loss is related to the input and output voltage, switching frequency, power, and current ripple. The inductor loss includes copper loss and iron loss. To analyze the losses of the converter, we first need to determine the peak and valley current values ​​I during the switching process. max and I min ;set up The effective value of the inductor current is I rms , whose value is the input current I in 1 / N of In high-power converters, MOS are usually connected in parallel to increase the current carrying capacity. Let the number of parallel MOS be n. d , its value is related to the number of branches N; for an N-branch parallel Boost converter, the power device loss of a single branch is Where P con is the conduction loss, P on and P off is the turn-on and turn-off loss, P oss is the output capacitor loss; R ds is the MOS on-state resistance, Q gs is the gate-source charge, Q gd is the gate-drain charge, V th is the threshold voltage, V mp is the Miller voltage, R on To turn on the drive resistor, R off is the turn-off drive resistor, C oss is the equivalent output capacitance; Under the maximum voltage and current stress, the switch loss P sw is the current ripple ΔI L and the switching frequency f s Function P sw =P sw (ΔI L ,f s ) (16) Inductor loss P L Including copper loss P cu and iron loss P Fe Two parts; for a certain core shape, after the inductance is determined, the core cross-sectional area A e is a variable, then the number of winding turns n w It can be solved; for copper loss P cu , for a given winding width and thickness d w and h w , under the maximum current stress, the copper loss P cu is about the current ripple ΔI L , switching frequency f s and the core cross-sectional area A e The function of P cu =P cu {[L(ΔI L ,f s ),A e ],A e }=P cu (ΔI L ,f s ,A e ) (17) Iron loss P Fe The inductor core material and core volume V Fe , switching frequency f s And the current ripple ΔI L For a certain core material, the iron loss calculation formula is: P Fe =P Fe [V Fe (A e ,ΔI L ,f s ),ΔI L ,f s ]=P Fe (A e ,ΔI L ,f s ) (18) From equations (16) to (18), we can see that the converter loss P loss Expressed as P loss (ΔI L ,f s ,A e )=P sw (ΔI L ,f s )+P cu (ΔI L ,f s ,A e )+P Fe (ΔI L ,f s ,A e ) (19) It can be obtained that the converter efficiency η is expressed as Equations (13) and (20) establish a comprehensive model of converter efficiency and passive power component volume. The comprehensive model includes three independent variables, namely, current ripple ΔI L , switching frequency f s and the core cross-sectional area A e ; Within the range of independent variable variation, through iterative calculation, the dual optimization of converter efficiency and power density can be achieved; (4) Comprehensive optimization of converter parameters; the converter loss and volume are three-variable functions. Within the respective value ranges of the three independent variables, the corresponding loss and volume results are calculated; In combination with the converter efficiency requirements, among the independent variable values ​​whose losses do not exceed the limit, select the passive component volume and the smallest independent variable value result, and calculate the corresponding inductance, input capacitance and output capacitance values.

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