A dual-boost converter and its modulation method

Abstract

The present invention discloses a dual-boost converter and its modulation method, belonging to the technical field of converters. One end of the inductor L1 of the present invention is connected to the input power supply U in at one end. The other end of the inductor L1 is respectively connected to the anode of the diode D1 and the terminal 1 of the power switch tube S1; the cathode of the diode D1 is respectively connected to the terminal 1 of the power switch tube S3 and one end of the capacitor C1; the other end of the capacitor C1 is connected to the terminal 2 of the power switch tube S1 and the other end of the DC side; one end of the inductor L2 is connected to the input power supply U in at one end. The other end of the inductor L2 is respectively connected to the terminal 1 of the power switch tube S2 and one end of the capacitor C2; the other end of the capacitor C2 is connected to the terminal 2 of the power switch tube S4 and the anode of the diode D2; the cathode of the diode D2 is respectively connected to the terminal 2 of the power switch tube S2 and the other end of the DC side. Aiming at the problems of low boost ratio and leakage current existing in the boost inverter of the prior art, the present invention has high integration and a relatively high boost ratio, and can effectively solve the leakage current problem.

Description

Technical Field

[0001] The present invention relates to the technical field of power electronic converters, and in particular to a dual-boost converter and a corresponding modulation method thereof. Background Art

[0002] Single-phase voltage source inverters (VSIs), as the interface between DC output and AC output, play a key role in energy storage systems (ESSs) and distribution systems such as photovoltaic (PV). The traditional voltage-source inverter is a buck-type DC-AC power converter, where the DC power supply voltage is higher than the AC output peak voltage. In the case of a low input voltage, there are generally two methods to achieve voltage boost conversion: one is to use a power frequency transformer. The second is a two-stage photovoltaic boost inverter, which adds a boost-type DC-DC converter before the inverter bridge to obtain a higher AC output voltage. However, the result of the additional power converter in both methods is to increase the volume, weight, and cost of the system, and due to the two-stage conversion affecting the conversion efficiency of the entire inverter, the conversion efficiency is low and the boost ratio is low. On the other hand, due to the different grounding of the input and output, a large leakage current will occur in practical applications, causing safety and electromagnetic interference problems.

[0003] In recent years, more and more scholars have started to turn their research attention to single-stage boost inverters. In "Switched inductor Z-source inverter", a single-stage boost inverter based on the Z-source topology is proposed, and its output AC voltage may be higher than the DC input voltage. However, it requires a Z-source network composed of large inductors and large capacitors, and there is still a leakage current problem. The literature "A Novel Single Stage Zero Leakage Current Transformer-less Inverter for Grid Connected PV Systems" IEEE2015 discloses a single-stage inverter, which is composed of replacing the diode combination of the inverting and non-inverting Cuk inverters. Due to the inherent buck-boost ability of the Cuk converter, the output voltage can be higher or lower than the input voltage, and the input current ripple of this inverter is low. Its disadvantage is that the starting point of this inverter is to reduce the leakage current in the voltage conversion of the photovoltaic array. There are many switching devices, which increase the volume of the circuit, have large losses, and reduce the efficiency of the inverter. Summary of the Invention

[0004] 1. Technical Problems to be Solved by the Invention

[0005] In view of the problems of low boost ratio and leakage current existing in the boost inverters of the prior art, the present invention provides a dual-boost converter and its modulation method. The converter of the present invention has a high integration level, a relatively high boost ratio, and the DC side and the AC side share the same ground, avoiding common-mode interference and having no high-frequency leakage current.

[0006] 2. Technical solution

[0007] To achieve the above object, the technical solution provided by the present invention is as follows:

[0008] A dual-boost converter of the present invention includes power switching tubes S1, S2, S3, and S4, diodes D1, D2, inductors L1, L2, and capacitors C1, C2;

[0009] One end of inductor L1 is connected to the input power supply U in One end, and the other end of inductor L1 is respectively connected to the anode of diode D1 and terminal 1 of power switching tube S1;

[0010] The cathode of diode D1 is respectively connected to terminal 1 of power switching tube S3 and one end of capacitor C1;

[0011] The other end of capacitor C1 is connected to terminal 2 of power switching tube S1 and the other end of the DC side;

[0012] One end of inductor L2 is connected to the input power supply U in One end, and the other end of inductor L2 is respectively connected to terminal 1 of power switching tube S2 and one end of capacitor C2;

[0013] The other end of capacitor C2 is connected to terminal 2 of power switching tube S4 and the anode of diode D2;

[0014] The cathode of diode D2 is respectively connected to terminal 2 of power switching tube S2 and the other end of the DC side;

[0015] Terminal 2 of power switching tubes S1 and S2, one end of capacitor C1, the cathode of diode D2, and the other end of the DC side, and one end of the AC side are connected and all grounded;

[0016] Terminal 2 of power switching tube S3 and terminal 1 of S4 and one end of the AC side are connected to node A;

[0017] Terminal 2 of power switching tubes S1 and S2, one end of capacitor C1, the cathode of diode D2, and the other end of the AC side are connected to node B;

[0018] Node A and B form the output terminal.

[0019] A dual-boost converter of the present invention further includes a filter, the node A and B are connected to the input end of the filter, and the output end of the filter is connected to the power grid U g or load RO 。

[0020] A modulation method of a dual-boost converter according to the above-mentioned converter has the same working principle within the positive and negative half-cycles of the sinusoidal modulation wave;

[0021] When the modulation wave is greater than the carrier wave, control the power switch tubes S1 and S3 to conduct, S2 and S4 to disconnect, the diode D1 to cut off, and the diode D2 to conduct. The input power supply U in charges the inductor L1; the input power supply U in and the inductor L2 charge the capacitor C2; the voltage U C1 across the capacitor C1 and the output voltage u o form a closed loop. At this time, the output voltage amplitude U om = +mU C1 , where m is the modulation ratio;

[0022] When the modulation wave is less than the carrier wave, control the power switch tubes S2 and S4 to conduct, S1 and S3 to disconnect, the diode D1 to conduct, and the diode D2 to cut off. The input power supply U in and the inductor L1 charge the capacitor C1; the input power supply U in charges the inductor L2 through the switch tube S2; the capacitor C2, the power switch tubes S2 and S4 and the output voltage u o form a closed loop; the voltage U C2 across the capacitor C2 supplies power to the power grid or the load R O . At this time, the output voltage amplitude U om = -mU C1 ;

[0023] The inductors L1 and L2 are in the discharging state and the current discontinuous state. The current i L1 or i L2 flowing through the inductor L1 or L2 linearly decreases until it becomes zero, and the diode D1 or D2 cuts off.

[0024] A dual-boost converter of the present invention includes power switch tubes S1, S2, S3, S4 and S5, a diode D1, inductors L1, L2 and capacitors C1, C2;

[0025] One end of the inductor L1 is connected to one end of the input power supply U in , and the other end of the inductor L1 is respectively connected to the anode of the diode D1 and the terminal 1 of the power switch tube S1;

[0026] The cathode of the diode D1 is respectively connected to the terminal 1 of the power switch tube S3 and one end of the capacitor C1;

[0027] The other end of the capacitor C1 is connected to the terminal 2 of the power switch tube S1 and the other end of the DC side;

[0028] One end of the inductor L2 is connected to the input power supply U in One end, the other end of the inductor L2 is respectively connected to the terminal 1 of the power switch tube S2 and one end of the capacitor C2;

[0029] The other end of the capacitor C2 is connected to the terminal 2 of the power switch tube S4 and the terminal 2 of the power switch tube S5;

[0030] The terminal 1 of the power switch tube S5 is respectively connected to the terminal 2 of the power switch tube S2 and the other end of the DC side;

[0031] The terminal 2 of the power switch tubes S1 and S2, one end of the capacitor C1, the terminal 1 of the power switch tube S5, and the other end of the DC side, as well as one end of the AC side are connected and all grounded;

[0032] The terminal 2 of the power switch tube S3, the terminal 1 of the power switch tube S4, and one end of the AC side are connected to the node A;

[0033] The terminal 2 of the power switch tubes S1 and S2, one end of the capacitor C1, the terminal 1 of the power switch tube S5, and the other end of the AC side are connected to the node B;

[0034] The nodes A and B form the output terminal.

[0035] Preferably, when the diode D2 is replaced by the power switch tube S5; the modulation method is: in the positive half cycle of the power frequency, the power switch tubes S1, S3, and S5 always work in the SPWM state, the power switch tube S3 works in the opposite state to the power switch tube S1, the power switch tube S1 is synchronized with S5, and the power switch tubes S2 and S4 are in the off state; in the negative half cycle of the power frequency, the power switch tubes S1, S3, and S5 are all in the off state, the power switch tube S2 works in the SPWM state, and S4 is in the power frequency conduction state.

[0036] A modulation method of a dual-boost converter according to the present invention, according to the above-mentioned converter,

[0037] In the positive half cycle when the output voltage u o is greater than zero, when the modulation wave is greater than the carrier wave, control the power switch tubes S1 and S3 to conduct, S2, S4, and S5 to disconnect, the diode D1 to cut off, and the input power supply U in charges the inductor L1; the voltage U across the capacitor C1 C1 forms a closed loop with the output voltage u o At this time, the amplitude of the output voltage U om = +mU C1 , where m is the modulation ratio;

[0038] When the modulation wave is less than the carrier wave, control the power switch tube S5 to conduct, S1, S2, .S3 and S4 are disconnected, diode D1 conducts, and the input power supply U in and inductor L1 charge capacitor C1. Inductor L1 is in a discharging state and a current discontinuous state; power switch S5 and the anti-parallel diode D of switch S4 S4 provide freewheeling, and the output voltage amplitude U om = 0;

[0039] In the negative half cycle where the output voltage u o is less than zero, when the modulation wave is greater than the carrier wave, control power switches S2 and S4 to conduct, and S1, S3, and S5 to disconnect. Diode D1 is cut off, and the input power supply U in charges inductor L2, and the voltage U across capacitor C2 C2 powers the power grid or load R O , and the output voltage amplitude U om = -mU C1 ;

[0040] When the modulation wave is less than the carrier wave, control power switch S4 to conduct, and S1, S2, S3, and S5 to disconnect. Diode D1 remains cut off, and the input power supply U in and inductor L2 charge capacitor C2. Inductor L2 is in a discharging state and a current discontinuous state; power switch S5 and the anti-parallel diode D of switch S4 S4 provide freewheeling, and the output voltage amplitude U om = 0.

[0041] According to the dual-boost converter described above, it further includes a filter. The nodes A and B are connected to the input end of the filter, and the output end of the filter is connected to the power grid or load R O .

[0042] 3. Beneficial effects

[0043] Adopting the technical solution provided by the present invention, compared with the existing well-known technologies, it has the following remarkable effects:

[0044] (1) For the dual-boost converter of the present invention, the DC side and the AC side share the same ground, avoiding common-mode interference and there is no high-frequency leakage current; and since capacitors C1 and C2 are energy storage elements, playing the role of energy conversion, and the output is AC, the values of capacitors C1 and C2 are flexible and can be non-polar capacitors, making the circuit work reliably and increasing the working life of the circuit.

[0045] (2) A derived topology of the dual boost converter of the present invention, compared with the original structure, only adds 1 power switch. Compared with some circuit structures such as H6 for suppressing high-frequency leakage current, the number of power switches is less, thus reducing the use of power switches, lowering the loss, improving the conversion efficiency of the system, reducing the system cost and increasing the integration degree. The circuit occupies a small space.

[0046] (3) In each working mode of the dual boost converter of the present invention, at most two power switches work, reducing the conduction loss of the switches.

[0047] (4) A modulation method of the dual boost converter of the present invention works in different modes respectively in the positive half cycle where the output voltage is greater than zero and the negative half cycle where the output voltage is less than zero, forming different inverter circuits, which can realize boost inversion, that is, the inverter can normally complete the inversion function and has a stable AC output, and can achieve a relatively high boost ratio. Description of the Drawings

[0048] Figure 1 is a schematic diagram of the circuit structure of an embodiment of the present invention;

[0049] Figure 2 is a modulation strategy diagram of an embodiment of the present invention;

[0050] Figure 3 is a schematic diagram of the first working mode of the circuit of an embodiment of the present invention;

[0051] Figure 4 is a schematic diagram of the second working mode of the circuit of an embodiment of the present invention;

[0052] Figure 5 is a schematic diagram of the third working mode of the circuit of an embodiment of the present invention;

[0053] Figure 6 is a schematic diagram of the fourth working mode of the circuit of an embodiment of the present invention;

[0054] Figure 7 is a schematic diagram of the circuit structure of another dual boost converter of the present invention;

[0055] Figure 8 is a modulation strategy diagram of another dual boost converter of the present invention;

[0056] Figure 9 is a schematic diagram of the first working mode of the circuit of another dual boost converter of the present invention;

[0057] Figure 10 is a schematic diagram of the second working mode of the circuit of another dual boost converter of the present invention;

[0058] Figure 11 Schematic diagram of the third circuit operating mode of another dual-boost converter of the present invention;

[0059] Figure 12 Schematic diagram of the fourth circuit operating mode of another dual-boost converter of the present invention;

[0060] Figure 13 Schematic diagram of the fifth circuit operating mode of another dual-boost converter of the present invention;

[0061] Figure 14 Schematic diagram of the sixth circuit operating mode of another dual-boost converter of the present invention;

[0062] Figure 15 Schematic diagram of the output current, output voltage, and arm voltage of the circuit according to an embodiment of the present invention;

[0063] Figure 16 Schematic diagram of the inductor current of the circuit according to an embodiment of the present invention in the full DCM state;

[0064] Figure 17 THD value of the output voltage when the capacitors C1 = C2 = 200uF;

[0065] Figure 18 Simulation waveform diagram of the inductor current of the circuit of another dual-boost converter of the present invention in the full DCM state;

[0066] Figure 19 Schematic diagram of the arm voltage of the circuit of another dual-boost converter of the present invention under the proposed modulation method. Detailed implementation manner

[0067] To further understand the content of the present invention, the present invention will be described in detail in combination with the accompanying drawings and embodiments.

[0068] In the present invention, terms such as first, second, one end, the other end, etc. are set for the convenience of describing the technical solution of the present invention, and have no specific limiting effect, and are all general references, and do not constitute a limiting effect on the technical solution of the present invention.

[0069] In the present invention, the power switching transistors (including power switching transistors S1, S2, S3, S4, and S5) can use MOSFET devices, or other power switching transistors such as IGBTs or triodes. When using an IGBT or a triode, terminals 1, 3, and 2 of the power switching transistors S1 - S5 respectively correspond to the collector, base, and emitter of the power switching transistors S1 - S5. When using a MOSEFET, terminals 1, 3, and 2 of the power switching transistors S1 - S5 respectively correspond to the drain, gate, and source of the power switching transistors S1 - S5.

[0070] The voltage at the output terminal of the filter, i.e., the power grid voltage U g at both ends or the load R O The voltage across both ends is denoted as the output voltage u o , corresponding to the output voltage amplitude U om ; The output current of the filter, i.e., the current flowing through the load or the power grid, is denoted as the output current i o , corresponding to the output current amplitude of the filter being I om , The input voltage of the filter, i.e., the voltage between circuit nodes A and B, is U AB .

[0071] Embodiment 1

[0072] As Figure 1 shown, a dual-boost converter in this embodiment includes power switch tubes S1, S2, S3, and S4, diodes D1, D2, inductors L1, L2, and capacitors C1, C2; One end of inductor L1 is connected to the input power supply U in at one end, and the other end of inductor L1 is respectively connected to the anode of diode D1 and terminal 1 of power switch tube S1; The cathode of diode D1 is respectively connected to terminal 1 of power switch tube S3 and one end of capacitor C1; The other end of capacitor C1 is connected to terminal 2 of power switch tube S1 and the other end of the DC side; One end of inductor L2 is connected to the input power supply U in at one end, and the other end of inductor L2 is respectively connected to terminal 1 of power switch tube S2 and one end of capacitor C2; The other end of capacitor C2 is connected to terminal 2 of power switch tube S4 and the anode of diode D2; The cathode of diode D2 is respectively connected to terminal 2 of power switch tube S2 and the other end of the DC side; Terminals 2 of power switch tubes S1 and S2, the cathode of capacitor C1 and diode D2, and the other end of the DC side, as well as one end of the AC side, are connected and all grounded; Terminal 2 of power switch tube S3 and terminal 1 of S4 and one end of the AC side are connected to node A; Terminals 2 of power switch tubes S1 and S2, one end of capacitor C1, the cathode of diode D2, and the other end of the AC side are connected to node B; Nodes A and B form the output terminal.

[0073] Furthermore, as Figure 1 shown, the DC side is a DC power supply U in , which is the output voltage of a photovoltaic panel in an actual application scenario. On the AC side, nodes A and B are connected in parallel to the input terminal of the filter, and the output terminal of the filter is connected in parallel to the load or the power grid (feeding the converted electrical energy directly back to the power grid). Among them, the filter filters the voltage U AB to remove harmonic interference, and can be selected according to the actual application scenario, and can be a single-inductor filter, an LC filter, an LCL filter, etc. The output of the filter can be connected to the power grid or the load.

[0074] To achieve the effect of boosting and inverting the DC power supply on the DC side, most of the existing technologies adopt the technical solution of series combination of a DC / DC converter and an inverter. However, there are still problems such as a large number of power switching devices, high losses, and low conversion efficiency. The inventor of this embodiment creatively proposed a converter to convert the DC power supply U on the DC side in for conversion, and the amplitude of the output voltage on the AC side is greater than the DC power supply U in . The number of components is small, especially the number of power switching devices is small, thus reducing the switching loss and the circuit cost. Moreover, the boost ratio is high, the volume is small, and the weight is light, which can be more widely promoted and applied.

[0075] Compared with the comparative literature in the background technology (\"A Novel Single Stage Zero Leakage Current Transformer-less Inverter for Grid Connected PV Systems\"), the number of switching tubes used in this embodiment is small. In that literature, four power switching tubes operate at high frequencies, which will undoubtedly increase the switching loss and reduce the service life of the power switching tubes, thereby reducing the conversion efficiency and service life of the entire boost inverter. In addition, the number of power switching tubes in that literature is one more than that in this embodiment, which will undoubtedly increase the cost of the boost inverter. The circuit structure of the boost inverter in this embodiment creatively solves the above problems. Its working modes include Mode 1, Mode 2, Mode 3, and Mode 4. The details are as follows:

[0076] Mode 1

[0077] Combined with Figure 2 and Figure 3 , switch tubes S1 and S3 are turned on, and switch tubes S2 and S4 are turned off. Diode D1 is reversely cut off, and diode D2 is turned on. The current path is as Figure 3 shown. The output voltage of the inverter is positive. At this time, the output voltage amplitude U om = +mU C1 , where m is the modulation ratio. At this time, inductor L1 is charged by the input power supply U in , and its current increases linearly. And because inductor L2 and the input power supply U in are connected in series to charge capacitor C2, inductor L2 is in the discharging state and the current discontinuous state, and the current of inductor L2 gradually decreases. In this state, the energy stored in capacitor C2 is released to the power grid and inductor L0 through switch tube S3. When the energy stored in the inductor is completely released to capacitor C1, the current of inductor L2 decreases to zero, and diode D2 is reversely biased and cut off. When switch tube S1 is turned off, this mode ends.

[0078] Mode 2

[0079] Combined with Figure 2 and Figure 4 , the switching transistors S1 and S3 are turned off, and the switching transistors S2 and S4 are turned on. The current path is as Figure 4 shown. At this time, the inductor L1 and the input power supply U in are connected in series to charge the capacitor C1. The inductor L1 is in a discharging state and a discontinuous current state, and the current of the inductor L1 gradually decreases. The inductor L2 is charged by the input power supply U in . At the same time, the energy stored in the inductor L0 is released to the load through the anti-parallel diodes of the switching transistors S2 and S4. When the current of the inductor L1 drops to zero, the diode D1 is reversely cut off. When the switching transistors S2 and S4 are turned off, this mode ends. Then, in the positive half-cycle of the sine wave, the cycle repeats from mode one to mode two.

[0080] Mode three

[0081] Combined with Figure 2 and Figure 5 , the switching transistors S1 and S3 are turned on, and the switching transistors S2 and S4 are turned off. Energy is transferred from the DC link to the grid side, as Figure 5 shown. The operating principles of the inductors L1 and L2 are the same as those in mode one and will not be elaborated here. The energy stored in the inductor L0 is still released to the grid through the switching transistor S3. When the current of the inductor L2 decreases to zero, the diode D2 is reversely biased and cut off. When the switches S1 and S3 are turned off, this mode ends.

[0082] Mode four

[0083] Combined with Figure 2 and Figure 6 , the switch S1 and S3 are turned off, and the switching transistors S2 and S4 are turned on. The diode D1 is turned on, and the diode D2 is reversely biased and cut off. The negative output voltage of the inverter is generated by the capacitor C2. The current path is as Figure 6 shown. At this time, the inductor L2 is charged by the input power supply U in , and its current increases linearly. At the same time, the capacitor C1 is charged in series by the inductor L1 and the input power supply U in . In addition, when the inductance current of L1 drops to zero, the diode D1 is cut off. When the switching transistors S2 and S4 are turned off, this mode ends. Then, in the negative half-cycle of the sine wave, the cycle repeats from mode three to mode four.

[0084] The operating principle of the converter is the same in the positive and negative half-waves of the sinusoidal modulation wave. In mode one, by using the unidirectional conduction function of the diode D2, a closed loop is formed, and the electrical energy stored in the input power supply U in is transferred to the inductor L1. At the same time, by controlling the conduction of the power switching transistors S1 and S3, the electrical energy on the capacitor C1 is transferred to the output end, and after the modulation and filtering of the filter, the output voltage u o; In Mode 4, by utilizing the one-way conduction of diode D1, a closed loop is formed, and the input power supply U in and inductor L2 charge capacitor C1 together. By controlling the conduction of power switching transistors S2 and S4, the electrical energy on capacitor C2 is transferred to the output terminal, and the output voltage u is formed through the modulation and filtering of the filter o .

[0085] Boost ratio analysis

[0086] To simplify the analysis of the inverter, the following assumptions are made:

[0087] (1) All semiconductor power devices are ideal devices.

[0088] (2) Capacitors C1 and C2 are large enough to keep their voltages constant within one switching period; the filter inductor L o is large enough so that its current remains constant within one switching period.

[0089] (3) The currents of inductors L 1( and L2 operate in the discontinuous conduction mode (DCM).

[0090] Assume that the turn-on time of power switching transistor S1 is t on . According to the regular symmetric sampling rule, the conduction time can be obtained as:

[0091]

[0092] where m is the modulation ratio. To simplify the analysis, all component losses can be ignored for this converter. Since the proposed circuit is symmetric, the power generated by the input power supply U in is half of the output power, and the formula can be expressed as:

[0093]

[0094] The positive and negative half-cycles of this inverter are symmetric. To simplify the analysis, the positive half-cycle is considered. By applying the volt-second principle of boost inductor L1 within half of the switching period, the following equation can be obtained:

[0095]

[0096] where U om is the amplitude of the inverter output voltage. Since the input average current of the input power supply U in is equal to the average current of inductor L1, it can be concluded that:

[0097]

[0098] The output voltage can be expressed as:

[0099]

[0100] To simplify the analysis, the effective value of sin(wt) is taken as Combining equations (1)-(5), the voltage gain of the proposed inverter can be expressed as:

[0101]

[0102] Compared with the Z-source inverter topology, which has disadvantages such as a more complex structure, a DC bus voltage lower than the capacitor voltage, and the voltage gain being limited by the shoot-through duty ratio and modulation ratio, this embodiment has the advantages of a high voltage gain, improved controllability of the DC bus voltage on the input side of the inverter (referring to the voltage across capacitors C1 and C2 in this embodiment), and low switching losses. In addition:

[0103] 1. The boost inverter of this embodiment significantly reduces the number of components, lowers the system cost, and improves the integration level;

[0104] 2. Compared with the traditional two-stage boost inverter, there is no need to consider the problem of front-back stage matching, and the boost ratio is greatly increased, with a wide voltage adjustable range;

[0105] 3. Only four power switching tubes are used, and only two power switching tubes operate at a high frequency, reducing the switching losses and having a high inverter efficiency;

[0106] 4. Compared with the traditional circuit structure, not all power switching tubes are paralleled with diodes, reducing the number of circuit components, making the circuit structure simple and occupying less space;

[0107] 5. The parameters of the inductor and capacitor are small, and a non-polar capacitor is used for the DC bus capacitor in the circuit, making the circuit work reliably, increasing the working life of the circuit, and reducing the maintenance and management cost;

[0108] 6. Generate an output AC voltage higher than the input voltage;

[0109] 7. The THD of the output voltage / current is low.

[0110] Embodiment 2

[0111] Another dual-boost converter of this embodiment, as Figure 7 shown, includes power switching tubes S1, S2, S3, S4, and S5, diode D1, inductors L1, L2, and capacitors C1, C2; one end of inductor L1 is connected to the input power supply U inOne end of the inductor L1 is connected to the anode of the diode D1 and the terminal 1 of the power switch S1 respectively; the cathode of the diode D1 is connected to the terminal 1 of the power switch S3 and one end of the capacitor C1 respectively; the other end of the capacitor C1 is connected to the terminal 2 of the power switch S1 and the other end of the DC side; one end of the inductor L2 is connected to the input power supply U in One end of the inductor L2 is connected to the terminal 1 of the power switch S2 and one end of the capacitor C2 respectively; the other end of the capacitor C2 is connected to the terminal 2 of the power switch S4 and the terminal 2 of the power switch S5; the terminal 1 of the power switch S5 is connected to the terminal 2 of the power switch S2 and the other end of the DC side respectively; the terminals 2 of the power switches S1 and S2, one end of the capacitor C1, the terminal 1 of the power switch S5 and the other end of the DC side, and one end of the AC side are connected and grounded; the terminal 2 of the power switch S3, the terminal 1 of the power switch S4 and one end of the AC side are connected to the node A; the terminals 2 of the power switches S1 and S2, one end of the capacitor C1, the terminal 1 of the power switch S5, and the other end of the AC side are connected to the node B; the nodes A and B form the output terminals.

[0112] This embodiment also provides a modulation method corresponding to the above converter, as Figure 8 shown: In the positive half cycle of the power frequency, the power switches S1, S3, and S5 always work in the SPWM state. The power switches S3 and S5 are in opposite working states. The power switch S1 is synchronized with S3. The power switches S2 and S4 are in the off state; in the negative half cycle of the power frequency, the power switches S1, S3, and S5 are all in the off state. The power switch S2 works in the SPWM state, and S4 is in the power frequency conduction state.

[0113] Such as Figure 8 , a control method for a single-stage boost inverter realizes the control of the single-stage boost inverter by controlling the conduction sequence of the power switches, including modes a - f, which are described in detail as follows:

[0114] Mode a

[0115] In the positive half cycle where the output voltage u o is greater than zero, when the modulation wave is greater than the carrier wave, control the power switches S1 and S3 to conduct, and S2, S4, and S5 to disconnect. The diode D1 disconnects. The current path is as Figure 9 shown. At this time, the input power supply U in charges the inductor L1, and the current i L1 flowing through the inductor L1 increases linearly; at the same time, the energy stored in the capacitor C1 is released to the load and the inductor L0 through the switch S3. At this time, the output voltage amplitude U om =-mU C1 , where m is the modulation ratio. When the switch S1 is turned off, this mode ends.

[0116] Mode b

[0117] When the modulation wave is less than the carrier wave, such as Figure 8 , control the power switch tubes S1 and S3 to turn off, S5 to turn on, S2 and S4 to remain off, and the diode D1 to turn on. The current path is as Figure 10 shown. At this time, the input power supply U in is in series with the inductor L1 to charge the capacitor C1, and the current i L1 flowing through the inductor L1 decreases linearly. At the same time, the energy stored in the inductor L0 is released to the load through the antiparallel diodes of the switch tubes S5 and S4. When the current i L1 flowing through the inductor L1 drops to zero, this mode ends.

[0118] Mode c; the power switch tubes S1, S2, S3, and S4 remain off, and the power switch tube S5 remains on. The current path is as Figure 11 shown. The current of the inductor L1 is zero, and the energy stored in the inductor L0 is still released to the power grid through the antiparallel diodes of the switch tubes S5 and S4. The power switch tubes S4, S5, and the filter form a closed loop, and the filter supplies power to the power grid or the load R O , and the output voltage amplitude U om = 0; in the positive half cycle where the output voltage u o is greater than zero, it works continuously in the order of modes a, b, and c.

[0119] Mode d

[0120] In the entire negative half-cycle AC output, only the switch tube S2 works in the SPWM, the switch tube S4 is always on, and control the power switch tubes S1, S3, and S5 to remain off. When the switch tubes S2 and S4 are on, the switch tubes S1, S3, and S5 are off. The diode D1 is off. The current path is as Figure 12 shown. At this time, the input power supply U in charges the inductor L2, and the current i L2 flowing through the inductor L2 increases linearly; at the same time, the capacitor C2, the power switch tubes S2, S4, and the filter form a closed loop, and part of the energy stored in the capacitor C2 is released to the load and the inductor L0 through the switch tubes S2 and S4. At this time, the output voltage amplitude U om = -mU C1 , where m is the modulation ratio. When the power switch tube S2 turns off, this mode ends.

[0121] Mode e

[0122] When the modulation wave is less than the carrier wave, control the power switch tube S2 to turn off, the switch tube S4 to remain on, and the switch tubes S1, S3, and S5 to remain off. The current path is as Figure 13As shown. At this time, the input power supply U in is connected in series with the inductor L2 to charge the capacitor C2, and the current i flowing through the inductor L2 L2 decreases linearly. The inductor L2 is in the discharging state and the current discontinuous state. At the same time, the energy stored in the inductor L0 is released to the load through the anti-parallel diodes of the switching transistors S4 and S5. When the current of the inductor L2 drops to zero, this mode ends.

[0123] Mode f

[0124] The switching transistors S1, S2, and S3 remain off, and the switching transistors S4 and S5 remain on. The current path is as Figure 14 shown. The current of the inductor L2 is zero, and the energy stored in the inductor L0 is still released to the load through the anti-parallel diodes of the switching transistors S4 and S5. The amplitude of the output voltage U om = 0. Then, in the negative half-cycle of the sine wave, the operation is continuously repeated in the order of modes d, e, and f.

[0125] In this embodiment, the relationship calculation between the input voltage and the amplitude of the output voltage is the same as that in Embodiment 2. Compared with the traditional boost inverter, the turns ratio is high and the adjustable range is large.

[0126] When the present invention works, it operates in different modes in the positive half-cycle where the output voltage is greater than zero and the negative half-cycle where the output voltage is less than zero, forming different boost inverter circuits. The control signal of the power switching transistor S1 is a high-frequency switching signal. The power switching transistor S2 operates in the high-frequency state in the positive half-cycle where the output voltage u o is greater than zero, and is in the off state in the negative half-cycle; the power switching transistors S3 and S4 adopt the working mode of being normally off (on) in half a cycle, reducing the switching loss; the modulation modes of the power switching transistors S3 and S5 are the same, making the circuit structure simple and the control convenient.

[0127] The control scheme of this circuit is also relatively simple. Adopting the traditional single-closed-loop output voltage control scheme, when the input power supply U in fluctuates greatly, due to the fact that the voltages of the capacitors C1 and C2 cannot change suddenly, the circuit can still obtain a better-quality output voltage u o , with strong anti-interference ability and good stability.

[0128] After comparison, the working principle of a single-stage boost inverter in this embodiment is different from the boost inverter method combining a traditional boost converter and an inverter, mainly including the following points:

[0129] 1. The boost inverter in this embodiment is integrated and single-stage, without the need to consider the mutual matching problem between the output end of the boost converter and the input end of the inverter, and has a high integration level;

[0130] 2. The output terminal of the front-stage boost converter of the traditional two-stage series boost inverter needs to be provided with a filter, and the output terminal of the rear-stage inverter also needs to be provided with a filter. The filter takes up a large amount of space and is cumbersome to design, which will undoubtedly increase the volume of the entire circuit and the circuit design cost. The single-stage boost inverter of this embodiment creatively overcomes the above disadvantages and only requires one filter, taking up less space;

[0131] 3. The traditional two-stage series boost inverter also needs to consider the problem of matching the front-stage output with the rear-stage input, increasing the design cost. The single-stage boost inverter of this embodiment does not have this problem;

[0132] 4. The control circuit of the power switch tube of the single-stage boost inverter of this embodiment is simple, convenient to design, and low in cost. Unlike the traditional two-stage series boost inverter, the front and rear stages need to be controlled separately, resulting in high circuit design and control costs.

[0133] Compared with other single-stage boost inverters:

[0134] 1. The boost inverter of this embodiment significantly reduces the number of components, reduces the system cost, and improves the integration degree;

[0135] 2. Compared with the traditional two-stage boost inverter, there is no need to consider the problem of front and rear stage matching, and the boost ratio is greatly increased, with a wide adjustable range of voltage;

[0136] 3. Only five power switch tubes are used, and only two power switch tubes work in the high-frequency state, reducing the switching loss and having a high inversion efficiency;

[0137] 4. Compared with the traditional circuit structure, the parameters of the inductor and capacitor are small, and the DC bus capacitor in the circuit uses a non-polar capacitor, making the circuit work reliably, increasing the working life of the circuit, and reducing the maintenance and management cost;

[0138] 6. Generate an output AC voltage higher than the input voltage;

[0139] 7. The THD of the output voltage / current is relatively low.

[0140] Figure 15 This is the simulation waveform diagram of the output current, output voltage, and arm voltage of the circuit of the embodiment of the present invention. It can be seen that the output voltage has a high sinusoidal degree and no distortion, and at the same time, the functions of boosting and inversion are realized; Figure 16 This is the simulation waveform diagram of the inductor current L1 and the inductor current L2 of the embodiment of the present invention. It can be seen that the invention can work in the DCM mode; Figure 17 This is the THD value diagram of the output voltage. The THD value is only 1.61%, indicating that the sinusoidal effect is good; Figure 18It is the simulation waveform diagram of the inductor current of another dual-boost converter of the present invention. It can be seen that the present invention can also operate in the DCM mode; Figure 19 It is the simulation waveform diagram of the arm voltage of the circuit of another dual-boost converter of the present invention under the proposed modulation method. It can be seen that it is a switching pulse wave, verifying the feasibility of the present invention.

[0141] The above has schematically described the present invention and its implementation manners. This description is not restrictive. What is shown in the drawings is only one of the implementation manners of the present invention, and the actual structure is not limited thereto. Therefore, if those of ordinary skill in the art are inspired by it and design similar structural manners and embodiments without creative efforts without departing from the gist of the present invention, they shall fall within the protection scope of the present invention.

Claims

1. A dual-boost converter, characterized in that: It includes power switching transistors S1, S2, S3, and S4, diodes D1, D2, inductors L1, L2, and capacitors C1, C2; One end of the inductor L1 is connected to the input power supply U in One end, and the other end of the inductor L1 is respectively connected to the anode of the diode D1 and the terminal 1 of the power switch tube S1; The cathode of diode D1 is respectively connected to terminal 1 of power switching transistor S3 and one end of capacitor C1; The other end of the capacitor C1 is connected to terminal 2 of the power switch tube S1 and the other end of the input power supply U in ; One end of the inductor L2 is connected to the input power supply U in One end, and the other end of the inductor L2 is respectively connected to terminal 1 of the power switch tube S2 and one end of the capacitor C2; The other end of capacitor C2 is connected to terminal 2 of power switching transistor S4 and the anode of diode D2; The cathode of diode D2 is respectively connected to terminal 2 of power switch S2 and the other end of input power supply U in ; Terminal 2 of power switch tube S1, terminal 2 of power switch tube S2, the other end of capacitor C1, the cathode of diode D2 and the other end of input power supply U in are connected to each other and grounded; Terminal 2 of power switching transistor S3, terminal 1 of power switching transistor S4, and the other end of the AC side are connected to node A; Terminal 2 of power switching transistor S1, terminal 2 of power switching transistor S2, the other end of capacitor C1, the cathode of diode D2, and one end of the AC side are connected to node B; Nodes A and B form the output terminals.

2. The dual boost converter according to claim 1, wherein: It further includes a filter. The nodes A and B are connected to the input end of the filter, and the output end of the filter is connected to the power grid U g or the load R O .

3. A modulation method for a dual boost converter, characterized in that, For a dual-boost converter according to claim 1 or 2, the operating principle is the same within the positive and negative half-cycles of the sinusoidal modulation wave; When the modulation wave is greater than the carrier wave, control the power switch tubes S1 and S3 to conduct, and S2 and S4 to disconnect. The diode D1 is cut off, and the diode D2 is conducting. The input power supply U in charges the inductor L1; the input power supply U in and the inductor L2 charge the capacitor C2; the voltage U C1 across the capacitor C1 and the output voltage u o form a closed loop; When the modulation wave is less than the carrier wave, control the power switch tubes S2 and S4 to conduct, S1 and S3 to disconnect, the diode D1 to conduct, and the diode D2 to cut off. The input power supply U in and the inductor L1 charge the capacitor C1; the input power supply U in charges the inductor L2 through the switch tube S2; the capacitor C2, the power switch tubes S2 and S4, and the output voltage u o form a closed loop; the voltage U C2 across the capacitor C2 supplies power to the power grid or the load R O . Inductors L1 and L2 are in the discharge state and the current discontinuous state, and the current i flowing through inductor L1 or L2 L1 or i L2 linearly decreases until it becomes zero, and diodes D1 or D2 are turned off.

4. A dual boost converter, characterized in that: It includes power switching transistors S1, S2, S3, S4, and S5, diode D1, inductors L1, L2, and capacitors C1, C2; One end of the inductor L1 is connected to the input power supply U in One end, the other end of the inductor L1 is respectively connected to the anode of the diode D1 and the terminal 1 of the power switch tube S1; The cathode of diode D1 is respectively connected to terminal 1 of power switching transistor S3 and one end of capacitor C1; The other end of the capacitor C1 is connected to terminal 2 of the power switch tube S1 and the other end of the input power supply U in ; One end of the inductor L2 is connected to the input power supply U in One end, the other end of the inductor L2 is respectively connected to terminal 1 of the power switch tube S2 and one end of the capacitor C2; The other end of capacitor C2 is connected to terminal 2 of power switching transistor S4 and terminal 2 of power switching transistor S5; Terminal 1 of power switch tube S5 is respectively connected to terminal 2 of power switch tube S2 and the other end of input power supply U in ; Terminal 2 of power switch S1, terminal 2 of power switch S2, the other end of capacitor C1, terminal 1 of power switch S5 and the other end of input power supply U in are connected to each other and grounded; Terminal 2 of power switching transistor S3, terminal 1 of power switching transistor S4, and the other end of the AC side are connected to node A; Terminal 2 of power switching transistor S1, terminal 2 of power switching transistor S2, the other end of capacitor C1, terminal 1 of power switching transistor S5, and one end of the AC side are connected to node B; Nodes A and B form the output terminals.

5. A dual-boost converter according to claim 4, characterized in that: It further includes a filter. The nodes A and B are connected to the input end of the filter, and the output end of the filter is connected to the power grid or the load R O .

6. A modulation method for a dual boost converter, characterized in that, For a dual-boost converter according to claim 4 or 5, the modulation method is as follows: within the positive half-cycle of the power frequency, power switching transistors S1, S3, and S5 always operate in the SPWM state, the operating state of power switching transistor S5 is opposite to that of power switching transistor S1, power switching transistors S1 and S3 are synchronous, and power switching transistors S2 and S4 are in the off state; within the negative half-cycle of the power frequency, power switching transistors S1, S3, and S5 are all in the off state, power switching transistor S2 operates in the SPWM state, and S4 is in the power-frequency conduction state.

7. The modulation method of a dual-boost converter according to claim 6, characterized in that: At the positive half-cycle where the output voltage u o is greater than zero, when the modulation wave is greater than the carrier wave, the control power switch tubes S1 and S3 are turned on, and S2, S4, and S5 are turned off. The diode D1 is cut off, and the input power supply U in charges the inductor L1; the voltage U C1 across the capacitor C1 and the output voltage u o form a closed loop; When the modulation wave is less than the carrier wave, control the power switch tube S5 to conduct, S1, S2, S3, and S4 to disconnect, the diode D1 to conduct, and the input power supply U in and the inductor L1 charge the capacitor C1. The inductor L1 is in a discharging state and a current discontinuous state; the energy stored in the inductor L0 is released to the load through the anti-parallel diode of the power switch tube S4 and the power switch tube S5; During the negative half-cycle where the output voltage u o is less than zero, when the modulating wave is greater than the carrier wave, the control power switch tubes S2 and S4 are turned on, while S1, S3, and S5 are turned off. The diode D1 is cut off, and the input power supply U in charges the inductor L2, and the voltage U across the capacitor C2 C2 supplies power to the power grid or the load R O for power supply; When the modulation wave is less than the carrier wave, control the power switch tube S4 to conduct, and S1, S2, S3, and S5 to disconnect. The diode D1 remains cut-off, and the input power supply U in and the inductor L2 charge the capacitor C2. The inductor L2 is in a discharging state and a current discontinuous state; the power switch tube S5 and the freewheeling diode of the switch tube S4 conduct freewheeling.

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