Variable Frequency Drive System and Control Method for Brushless Capacitor Synchronous Reluctance Motor

Abstract

The present invention discloses a variable-frequency drive system for a non-electrolytic-capacitor synchronous reluctance motor, which uses a thin-film capacitor with a small capacitance value at the bus bar end to replace the traditional large-capacity electrolytic capacitor. It includes a single-phase input power supply module, a single-phase rectifier, a thin-film capacitor, a three-phase inverter, an input voltage phase-locked loop, a current control module based on power balance, a power compensation module based on bus voltage control, a first subtractor, a speed PI regulator, a first multiplier, a second subtractor, a third subtractor, a maximum torque current ratio module, a fourth subtractor, a current controller, a fifth subtractor, a sixth subtractor, an IPark transformation module, a Park transformation module, a Clark transformation module, a space vector pulse width modulation module, and a synchronous reluctance motor. The present invention solves the problem of low reliability of the driver caused by the short lifespan of the electrolytic capacitor in the traditional large-capacity electrolytic-capacitor variable-frequency motor, improves the power factor on the input side, and enhances the motor performance.

Description

Technical Field

[0001] The present invention relates to the field of motor control, and specifically to a variable-frequency drive system and control method for a synchronous reluctance motor without electrolytic capacitors. Background Art

[0002] At present, motor variable-frequency drive technology has been widely used in industries, such as industrial robots, building buildings, electric vehicles, and the national defense and military fields, etc. Inverters in motor variable-frequency drive circuits can be divided into voltage-source inverters and current-source inverters according to the nature of the DC-side power supply. The DC side of a current-source inverter is a large inductor, and the current is basically non-pulsating, equivalent to a current source, with strong over-current suppression ability and suitable for starting-type loads that frequently accelerate and decelerate. The DC side of a voltage-source inverter is shunted with a large capacitor, and the voltage is basically non-pulsating, equivalent to a voltage source, with strong surge-voltage suppression ability. Due to the advantages of adjustable frequency and high efficiency of voltage-source inverters, existing motor variable-frequency drive systems generally consist of a voltage-source inverter, a rectifier, a power factor correction (PFC) circuit, and a motor. Its bus terminal is usually a large-capacity electrolytic capacitor, whose main function is to store the energy input from the grid side and stabilize the DC bus voltage to ensure that the motor has sufficient operating voltage. However, electrolytic capacitors have very obvious disadvantages. During use, current ripple and high temperature easily cause the electrolyte to evaporate, resulting in a low lifespan of electrolytic capacitors. For every 10°C increase in temperature, the lifespan is shortened by half, which seriously affects the reliability of the drive circuit. Approximately 60% of drive circuit failures are caused by electrolytic capacitors. At the same time, the disadvantages of large volume and heavy weight of electrolytic capacitors restrict the development of the system towards miniaturization and light weight. In addition, the presence of a large-capacity electrolytic capacitor on the bus will result in a small conduction angle of the diode, serious distortion of the input current, a significant increase in current harmonics, a poor input power factor, and serious pollution of the grid quality. In order to achieve a high power factor and low current harmonics on the grid side, it is usually necessary to add a power factor correction (PFC) circuit to the DC bus. However, introducing a PFC circuit will increase the cost and volume of the system, and the switching losses on the PFC circuit will also reduce the overall efficiency of the motor drive system.

[0003] Therefore, it is necessary to improve the existing technology. Summary of the Invention

[0004] The technical problem to be solved by the present invention is to provide a variable-frequency drive system and control method for a synchronous reluctance motor without electrolytic capacitors, so as to solve the problem of reduced reliability caused by the short lifespan of electrolytic capacitors in traditional variable-frequency drive circuits, and improve the grid-side power factor and motor performance.

[0005] To solve the above technical problems, the present invention provides a variable-frequency drive system for a synchronous reluctance motor without electrolytic capacitors, including a synchronous reluctance motor. The output of the synchronous reluctance motor is respectively connected to the input of a first subtractor and a current control module based on power balance; the output of a single-phase input power supply module is respectively connected to the input of an input voltage phase-locked loop, a current control module based on power balance, a power compensation module based on bus voltage control, and a single-phase rectifier. The output of the single-phase rectifier is connected to the input of a three-phase inverter and a power compensation module based on bus voltage control after being connected in parallel with a thin-film capacitor; the output of the three-phase inverter is respectively connected to the input of the synchronous reluctance motor and a Clark transformation module; the output of the Clark transformation module is connected to the input of a Park transformation module, and the output of the Park transformation module is respectively connected to the input of a third subtractor, a fourth subtractor, a current control module based on power balance, and a power compensation module based on bus voltage control;

[0006] The output of the input voltage phase-locked loop is respectively connected to the input of a current control module based on power balance and a power compensation module based on bus voltage control; the output of the current control module based on power balance is respectively connected to the input of a first multiplier and a second subtractor; the output of the power compensation module based on bus voltage control is respectively connected to the input of a fifth subtractor and a sixth subtractor;

[0007] The output of the first subtractor is connected to the input of a speed PI regulator. The output of the speed PI regulator is respectively connected to the input of a first multiplier and a maximum torque per ampere (MTPA) module. The output of the first multiplier is connected to the input of the second subtractor. The output of the second subtractor is connected to the input of the third subtractor. The output of the third subtractor is connected to the input of a current controller; the output of the maximum torque per ampere (MTPA) module is connected to the input of the fourth subtractor. The output of the fourth subtractor is connected to the input of the current controller; the output of the current controller is respectively connected to the input of the fifth subtractor and the sixth subtractor. The outputs of the fifth subtractor and the sixth subtractor are both connected to the input of an inverse Park transformation module (IPark). The output of the IPark transformation module is connected to the input of a three-phase inverter through a space vector pulse width modulation module (SVPWM).

[0008] The present invention also provides a control method for a synchronous reluctance motor using the variable-frequency drive system for a synchronous reluctance motor without electrolytic capacitors:

[0009] Step 1: Collect the two-phase voltages of the synchronous reluctance motor through two-phase sampling resistors. According to the formula i = u / R and i a +i b +i c = 0, obtain the two-phase actual currents i a 、ib and i c and input it into the Clark transformation module;

[0010] Step 2, the two-phase actual currents i a and i b and i c in the three-phase coordinate system are transformed by Clark transformation to obtain the α-axis current component i α and the β-axis current component i β , and output them to the Park transformation module;

[0011] Step 3, the α-axis current component i α and the β-axis current component i β are transformed by Park transformation to obtain the d-axis current i d and the q-axis current i q , and the q-axis current i q is respectively output to the third subtractor and the power compensation module based on bus voltage control, and the d-axis current i d is respectively output to the fourth subtractor, the current control module based on power balance and the power compensation module based on bus voltage control;

[0012] Step 4, the u in output by the single-phase input power supply module is respectively input into the input voltage phase-locked loop, the current control module based on power balance, the power compensation module based on bus voltage control, and the single-phase rectifier, and the i in output by the single-phase input power supply module is respectively input into the current control module based on power balance;

[0013] Step 5, the single-phase rectifier obtains the DC-side bus voltage value u dc according to Equation (3) and outputs it to the three-phase inverter and the power compensation module based on bus voltage control respectively;

[0014]

[0015] where, U in is the amplitude of the input voltage u in , ω in is the angular frequency of the input voltage u in , θ m is the starting conduction angle of the rectifier diode, and U dcmin is the minimum value of the DC-side bus voltage value u dc ;

[0016] Step 6, the input voltage phase-locked loop calculates and obtains the phase information θ in of the input voltage according to the u in, and output them to the current control module based on power balance and the power compensation module based on bus voltage control respectively;

[0017] Step 7: The current control module based on power balance calculates sin d using the d-axis current i in input in Step 3, the input voltage u in and input current i in input in Step 4, the phase information θ 2 of the input voltage input in Step 6, and the output speed ω of the synchronous reluctance motor. After calculation, sin in ω qc t and i qc are obtained; then i 2 ω in t is output to the first multiplier;

[0018] Step 8: The power compensation module based on bus voltage control calculates Δu d and Δu q using the d-axis current i in and q-axis current i dc input in Step 3, the input voltage u in input in Step 4, the DC bus voltage value u d input in Step 5, and the phase information θ q of the input voltage input in Step 6. Then, Δu q is output to the fifth subtractor, and Δu d is output to the sixth subtractor;

[0019] Step 9: Input the given motor speed ω ref into the first subtractor;

[0020] Step 10: The first subtractor obtains the speed difference Δω according to the following formula and outputs it to the speed PI regulator;

[0021] Δω = ω ref - ω (24)

[0022] where ω is the output speed of the synchronous reluctance motor;

[0023] Step 11: The speed PI regulator obtains T0 according to the following formula and outputs it to the first multiplier and the maximum torque per ampere (MTPA) module respectively: as follows:

[0024]

[0025] where L d is the direct-axis inductance, L q is the quadrature-axis inductance, ω e is the motor speed, pn is the number of pole pairs, U in is the amplitude of the input voltage u in and I in is the amplitude of the input current i in ;

[0026] Step 12: The first multiplier calculates i according to Equation (19) q0 and outputs it to the second subtractor:

[0027]

[0028] Step 13: The second subtractor calculates the q-axis reference current i according to Equation (17) qref and outputs it to the third subtractor:

[0029]

[0030] where C dc is the bus capacitor;

[0031] Step 14: The third subtractor calculates Δi according to Equation (26) q , and outputs Δi q to the current controller:

[0032] Δi q = i qref - i q (26)

[0033] Step 15: The maximum torque per ampere (MTPA) module calculates the d-axis reference current i according to Equation (27) dref and outputs it to the fourth subtractor:

[0034]

[0035] Step 16: The fourth subtractor calculates Δi according to Equation (28) d and outputs it to the current controller;

[0036] Δi d = i dref - i d (28)

[0037] Step 17: The current controller calculates the initial q-axis voltage u q0 and the initial d-axis voltage u d0 respectively according to Equations (29) and (30), and outputs the initial q-axis voltage u q0 to the fifth subtractor, and outputs the d-axis voltage u d0 to the sixth subtractor;

[0038] u q0 = Kp Δi q +K i ∫Δi q dt (29)

[0039] u d0 =K p Δi d +K i ∫Δi d dt (30)

[0040] where K p is the proportional coefficient of the current controller, and K i is the integral coefficient of the current controller;

[0041] Step 18: The fifth subtractor calculates the q-axis voltage u q according to Equation (31) and outputs it to the IPark transformation module;

[0042] u q =u q0 -Δu q (31)

[0043] Step 19: The sixth subtractor calculates the d-axis voltage u d according to Equation (32) and outputs it to the IPark transformation module:

[0044] u d =u d0 -Δu d (32)

[0045] Step 20: The IPark transformation module calculates the α-axis voltage component u α and the β-axis voltage component u β according to Equation (32):

[0046]

[0047] Output u a and u β to the space vector pulse width modulation module (SVPWM);

[0048] Step 21: Based on the u α and u β input in Step 20, the space vector pulse width modulation module (SVPWM) obtains six-way PWM signals by judging the sector where the synthesized vector is located, calculating the action time of each adjacent vector, and the conduction time and duty cycle of the six-bridge arm, and outputs the six-way PWM signals to the three-phase inverter;

[0049] Step 22: The three-phase inverter is based on the six-way PWM signals input in Step 21 and the DC bus voltage value u dcThe phase voltages of each phase of the motor are calculated according to Equation (34) as follows:

[0050]

[0051] Among them, u an is the phase voltage of phase A of the synchronous reluctance motor, u bn is the phase voltage of phase B of the synchronous reluctance motor, u cn is the phase voltage of phase C of the synchronous reluctance motor; the on-off states of the switching elements in the three-phase inverter circuit are defined as follows:

[0052]

[0053] Output the phase voltages u an , u bn , u cn of the three phases A, B, and C to the synchronous reluctance motor to achieve the drive control of the synchronous reluctance motor without electrolytic capacitors.

[0054] As an improvement to the control method of the synchronous reluctance motor in the variable-frequency drive system of the synchronous reluctance motor without electrolytic capacitors of the present invention:

[0055] The specific calculation process of the input voltage phase-locked loop described in Step 6 is as follows: The phase information θ in of the input voltage u in is transformed through trigonometric transformation to obtain cos(θ in ), and then the amplitude U in of the input voltage u in is multiplied by cos(θ in ), and after filtering by a low-pass filter and regulation by a PI regulator, the input voltage angular frequency reference value ω inref is obtained. The difference between the input voltage angular frequency reference value ω inref and the actual input voltage angular frequency ω in is calculated to obtain the input voltage angular frequency error Δω in , and finally the phase information θ in of the input voltage is obtained through an integration link.

[0056] As a further improvement to the control method of the synchronous reluctance motor in the variable-frequency drive system of the synchronous reluctance motor without electrolytic capacitors of the present invention:

[0057] The specific calculation process of the current control module based on power balance described in Step 7 is as follows:

[0058] The expressions of the input voltage u in and the input current i in are respectively as shown in Equations (4) and (5):

[0059] u in =U in sinωin t (4)

[0060]

[0061] where θ is the conduction angle of the rectifier diode; let (π - θ) / 2 = θ m ;

[0062] The power P on the input side in is as shown in Equation (6):

[0063]

[0064] where θ m is the starting conduction angle of the rectifier diode;

[0065] Assuming the input has a unity power factor, the DC bus voltage value is as shown in Equation (7):

[0066]

[0067] where U dcmin is the minimum value of the bus voltage;

[0068] The capacitor current i dc is:

[0069]

[0070] where sgn(x) is the sign function:

[0071]

[0072] The power P on the bus capacitor dc is:

[0073]

[0074] The instantaneous power consumption P of the motor m is as shown in Equation (11):

[0075] P m = P cu + P react + P mech (11)

[0076] where P cu is the copper loss of the motor, P react is the reactive power, P mech is the mechanical output power, and the expressions are as follows:

[0077]

[0078]

[0079] P mech =1.5ω e i q p n (L d -L q )i d (14)

[0080] Where R is the stator resistance; ignore the motor copper loss P cu and reactive power P react The influence of the instantaneous power consumption of the motor P m for:

[0081] P in -P dc =P m (15)

[0082] Combining formulas (6), (11) to (15), we get:

[0083]

[0084] According to equation (16), the q-axis given current i input to the third subtractor is qref The expression is as follows:

[0085]

[0086] Among them, the q-axis current compensation component corresponding to the capacitor power is i qc , the expression is as follows:

[0087]

[0088] The q-axis current corresponding to the input power is i q0 , the expression is as follows:

[0089]

[0090] From equations (17) to (19), the q-axis current based on power balance is given as:

[0091] i qref =i q0 -i qc (20)

[0092] In formula (19), Set as the output T0 of the speed PI regulator:

[0093] i q0 =T0sin 2 ω in t (21)

[0094] As a further improvement to the control method of the synchronous reluctance motor in the electrolytic-capacitor-free synchronous reluctance motor variable-frequency drive system of the present invention:

[0095] The implementation process of the power compensation module based on the bus voltage control described in step 8 is as follows: the input voltage u in The input voltage phase information θ obtained through the input voltage phase-locked loop in , after being processed by trigonometric functions and absolute values, is multiplied by the input voltage amplitude U in to obtain the ideal bus voltage Then, calculate the difference between the ideal bus voltage and the DC-side bus voltage value u dc . The difference ΔP of the inverter output power is obtained through a PI regulator. Divide ΔP by and then multiply by the proportionality coefficient K c to obtain the correction amount Δu of the voltage vector dq , and then calculate Δu d and Δu q according to Equation (22):

[0096]

[0097]

[0098] where θ dq is the tangent angle of the q-axis current i q and the d-axis current i d .

[0099] As a further improvement to the control method of the synchronous reluctance motor in the electrolytic-capacitor-free synchronous reluctance motor variable-frequency drive system of the present invention:

[0100] The Clark transformation formula is:

[0101]

[0102] As a further improvement to the control method of the synchronous reluctance motor in the electrolytic-capacitor-free synchronous reluctance motor variable-frequency drive system of the present invention:

[0103] The Park transformation formula is:

[0104]

[0105] The beneficial effects of the present invention are mainly reflected in:

[0106] The electrolytic-capacitorless variable-frequency drive system of the synchronous reluctance motor of the present invention uses thin-film capacitors to replace electrolytic capacitors. Compared with the traditional variable-frequency motor drive system with large-capacity electrolytic capacitors, it solves the problem of low reliability of the driver caused by the short lifespan of electrolytic capacitors, effectively reducing the volume, weight, and cost of the system; improving the power factor on the input side; enhancing the motor performance, and reducing the motor speed and torque fluctuations. BRIEF DESCRIPTION OF THE DRAWINGS

[0107] The following further elaborates on the specific embodiments of the present invention with reference to the accompanying drawings.

[0108] Figure 1 is the structural block diagram of the electrolytic-capacitorless variable-frequency drive system of the synchronous reluctance motor of the present invention;

[0109] Figure 2 is Figure 1 the principle block diagram of the Clark transformation module in

[0110] Figure 3 is Figure 1 the principle block diagram of the Park transformation module in

[0111] Figure 4 is Figure 1 the implementation flowchart of the input voltage phase-locked loop in

[0112] Figure 5 is Figure 1 the principle block diagram of the current control module based on power balance in

[0113] Figure 6 is Figure 1 the principle block diagram of the power compensation module based on bus voltage control in

[0114] Figure 7 is Figure 1 the implementation flowchart of the power compensation module based on bus voltage control in

[0115] Figure 8 is Figure 1 the principle block diagram of the IPark transformation module in

[0116] Figure 9 is Figure 1 the principle block diagram of the space vector pulse width modulation module in SPECIFIC EMBODIMENTS

[0117] The following further describes the present invention with reference to specific embodiments, but the protection scope of the present invention is not limited thereto:

[0118] Embodiment 1. An electrolytic-capacitorless variable-frequency drive system of a synchronous reluctance motor, as Figure 1As shown in the figure. In a traditional motor variable-frequency drive circuit, large-capacity electrolytic capacitors are usually used as bus capacitors in the bus. The output of a single-phase rectifier is connected in parallel with an electrolytic capacitor and then connected to the input of a three-phase inverter. For example, when the drive power is 1.5 kW, two electrolytic capacitors with a capacitance value of 560 μF and a withstand voltage value of 400 V are usually selected in parallel in the bus. As an energy storage device in the drive system, the electrolytic capacitor can not only play a filtering role at the rectifier end to stabilize the DC bus voltage, but also absorb the energy fed back by the subsequent synchronous reluctance motor to ensure that semiconductor devices are not damaged by the impact voltage. However, the electrolytic capacitor has a short service life, resulting in low reliability of the drive system. In the present invention, a thin-film capacitor with a small capacitance value is used to replace the traditional large-capacity electrolytic capacitor at the bus end, as Figure 1 shown. The output of the single-phase rectifier is connected in parallel with a thin-film capacitor and then connected to the input of the three-phase inverter, thereby constructing a variable-frequency drive system for a synchronous reluctance motor without electrolytic capacitors. For example, when the drive power is 1.5 kW, a thin-film capacitor with a capacitance value of 20 μF and a withstand voltage value of 400 V is used as the bus capacitor. The variable-frequency drive system for a synchronous reluctance motor without electrolytic capacitors constructed by the present invention can solve the problem of low reliability of the drive caused by the short service life of the traditional electrolytic capacitor, and can effectively reduce the volume, weight and cost of the system. At the same time, it can improve the power factor on the input side and improve the motor performance.

[0119] The variable-frequency drive system for a synchronous reluctance motor without electrolytic capacitors includes a synchronous reluctance motor, and the output of the synchronous reluctance motor is respectively connected to the inputs of a first subtractor and a current control module based on power balance;

[0120] The output of the single-phase input power supply module is respectively connected to the inputs of an input voltage phase-locked loop, a current control module based on power balance, and a power compensation module based on bus voltage control. At the same time, the output of the single-phase input power supply module is connected to the input of the single-phase rectifier. The output of the single-phase rectifier is connected in parallel with a thin-film capacitor and then connected to the inputs of the three-phase inverter and the power compensation module based on bus voltage control; the output of the three-phase inverter is respectively connected to the inputs of the synchronous reluctance motor and a Clark transformation module; the output of the Clark transformation module is connected to the input of a Park transformation module, and the output of the Park transformation module is respectively connected to the inputs of a third subtractor, a fourth subtractor, a current control module based on power balance, and a power compensation module based on bus voltage control;

[0121] The output of the input voltage phase-locked loop is respectively connected to the inputs of a current control module based on power balance and a power compensation module based on bus voltage control; the output of the current control module based on power balance is respectively connected to the inputs of a first multiplier and a second subtractor; the output of the power compensation module based on bus voltage control is respectively connected to the inputs of a fifth subtractor and a sixth subtractor;

[0122] The output of the first subtractor is connected to the input of the speed PI regulator. The output of the speed PI regulator is respectively connected to the inputs of the first multiplier and the maximum torque per ampere (MTPA) module. The output of the first multiplier is connected to the input of the second subtractor. The output of the second subtractor is connected to the input of the third subtractor. The output of the third subtractor is connected to the input of the current controller. The output of the maximum torque per ampere (MTPA) module is connected to the input of the fourth subtractor. The output of the fourth subtractor is connected to the input of the current controller. The output of the current controller is respectively connected to the inputs of the fifth subtractor and the sixth subtractor. The outputs of the fifth subtractor and the sixth subtractor are both connected to the input of the IPark transformation module. The output of the IPark transformation module is connected to the input of the three-phase inverter through the space vector pulse width modulation module (SVPWM).

[0123] The method for controlling the synchronous reluctance motor using the above electrolytic-capacitorless variable-frequency drive system for synchronous reluctance motor is specifically as follows:

[0124] Step 1: Sample the two-phase voltage of the synchronous reluctance motor through two-phase sampling resistors, and calculate the two-phase actual currents \(i\) in the three-phase coordinate system according to the formula \(i = u / R\). a and \(i\) b , and the phase current \(i\) c is calculated through the formula \(i\) a +\(i\) b +\(i\) c = 0. Output the two-phase actual currents \(i\) a , \(i\) b and \(i\) c in the three-phase coordinate system to the Clark transformation module.

[0125] Step 2: According to the two-phase actual currents \(i\) a , \(i\) b and \(i\) c in the three-phase coordinate system input in Step 1, as shown in Figure 2 , the Clark transformation is performed to obtain the \(α\)-axis current component \(i\) a , \(β\)-axis current component \(i\) β in the actual stationary two-phase coordinate system. The Clark transformation formula is:

[0126]

[0127] Output the \(α\)-axis current component \(i\) a , \(β\)-axis current component \(i\) β in the actual stationary two-phase coordinate system to the Park transformation module.

[0128] Step 3: According to the \(α\)-axis current component \(i\) a, β-axis current component i β , as Figure 3 shown, the d-axis current i d and q-axis current i q are obtained through Park transformation. The Park transformation formula is:

[0129]

[0130] The q-axis current i q is respectively output to the third subtractor and the power compensation module based on bus voltage control. At the same time, the d-axis current i d is respectively output to the fourth subtractor, the current control module based on power balance, and the power compensation module based on bus voltage control.

[0131] Step 4: The u in output by the single-phase input power supply module is respectively input to the input voltage phase-locked loop, the current control module based on power balance, the power compensation module based on bus voltage control, and the single-phase rectifier. And the i in output by the single-phase input power supply module is respectively input to the current control module based on power balance.

[0132] Step 5: The single-phase rectifier calculates the DC-side bus voltage value u in according to the u dc input in Step 4. The calculation formula is:

[0133]

[0134] where, U in is the amplitude of the input voltage u in , ω in is the angular frequency of the input voltage u in , θ m is the starting conduction angle of the rectifier diode, and U dcmin is the minimum value of the DC-side bus voltage value u dc .

[0135] The DC-side bus voltage value u dc is respectively output to the three-phase inverter and the power compensation module based on bus voltage control.

[0136] Step 6: The input voltage phase-locked loop can obtain the phase information θ in of the input voltage according to the input voltage u in input in Step 4, as Figure 4 shown. The principle is as follows: The phase information θ in of the input voltage u in is obtained through trigonometric transformation to get cos(θ in ), and then the input voltage u inThe amplitude U in is multiplied by cos(θ in ), filtered by a low-pass filter and regulated by a PI regulator to obtain the reference value ω inref of the input voltage angular frequency. Subtracting it from the actual input voltage angular frequency ω in gives the input voltage angular frequency error Δω in . Finally, the phase information θ in of the input voltage is obtained through an integration link. The phase information θ in of the input voltage is respectively output to the current control module based on power balance and the power compensation module based on bus voltage control.

[0137] Step 7: The current control module based on power balance calculates sin d ω in t and i in according to the d-axis current i in input in Step 3, the input voltage u Figure 5 shown in, the input current i 2 ω in t and i qc input in Step 4, the phase information θ

[0138] of the input voltage input in Step 6, and the output speed ω of the synchronous reluctance motor, as in shown. The specific calculation formulas are as follows: The expressions of the input voltage u

[0139] u in and the input current i in sinω in t are respectively as shown in Equation (4) and Equation (5):

[0140]

[0141] where U in is the amplitude of the input voltage u in ω in is the angular frequency of the input voltage u in I in is the amplitude of the input current i in θ is the conduction angle of the rectifier diode. Let (π - θ) / 2 = θ m , then the expression of the input-side power P in is as shown in Equation (6):

[0142]

[0143] In the formula: θ m is the starting conduction angle of the rectifier diode.

[0144] To improve the input power factor, the waveform of the input current i in should be as sinusoidal as possible and in phase with the input voltage u in . Assuming the input has a unity power factor, the DC bus voltage value is shown in Equation (7).

[0145]

[0146] where U dcmin is the minimum value of the bus voltage.

[0147] According to the expression of the bus voltage, the capacitor current i dc is:

[0148]

[0149] where C dc is the bus capacitor, and sgn(x) is the sign function, whose definition is shown in Equation (9):

[0150]

[0151] The power P dc on the bus capacitor is:

[0152]

[0153] The instantaneous power P m consumed by the motor is shown in Equation (11).

[0154] P m = P cu + P react + P mech (11)

[0155] where P cu is the copper loss of the motor, P react is the reactive power, and P mech is the mechanical output power, and the expressions are as follows:

[0156]

[0157]

[0158] P mech = 1.5ω e i q p n (L d - L q )i d (14)

[0159] where R is the stator resistance, L d is the direct-axis inductance, and Lq is the quadrature-axis inductance, ω e is the motor speed, p n is the number of pole pairs. Ignoring the copper loss P cu and the reactive power P react of the motor, according to the power balance relationship, the input-side power P in minus the power P dc on the bus capacitor should be equal to the instantaneous power consumption P m of the motor. Therefore, we have:

[0160] P in -P dc =P m (15)

[0161] Combining equations (6), (11) to (15), we can obtain:

[0162]

[0163] According to the above equation, the given q-axis current i qref input to the third subtractor is expressed as follows:

[0164]

[0165] where the q-axis current compensation component corresponding to the capacitor power is i qc , and the expression is as follows:

[0166]

[0167] The q-axis current corresponding to the input power is i q0 , and the expression is as follows:

[0168]

[0169] From equations (17) to (19), the given q-axis current based on power balance can be obtained as:

[0170] i qref =i q0 -i qc (20)

[0171] Equation (20) gives the given q-axis current i qref based on the power balance relationship. By controlling the motor with this, the goal of high power factor can be achieved. According to the above analysis, the q-axis current of the motor is positively correlated with the motor power. Therefore, in equation (19), can be set as the output T0 of the speed PI regulator, that is:

[0172] i q0 =T0sin 2 ωin t (21)

[0173] Output i qc to the second subtractor, and at the same time output sin 2 ω in t to the first multiplier.

[0174] Step 8: The power compensation module based on bus voltage control, as Figure 6 shown, according to the d-axis current i d and q-axis current i q input in step 3, u in input in step 4, the DC bus voltage value u dc input in step 5, the input voltage phase information θ in input in step 6, calculate to obtain Δu d and Δu q , and output Δu q to the fifth subtractor, and at the same time output Δu d to the sixth subtractor.

[0175] The implementation method of the power compensation module based on bus voltage control is as Figure 7 shown, specifically as follows: The input voltage u in passes through the input voltage phase-locked loop to obtain the input voltage phase information θ in , after passing through trigonometric functions and absolute value processing, multiply by the input voltage amplitude U in to obtain the ideal bus voltage Then calculate the difference between the ideal bus voltage and the DC bus voltage value u dc , obtain the difference ΔP of the inverter output power through the PI regulator, divide it by and then multiply by a certain proportionality coefficient K c to obtain the correction amount Δu dq of the voltage vector.

[0176] There are many compensation methods for the correction amount of the voltage vector. The optimal choice is to make the correction amount of the voltage vector in the same direction as the motor current vector. At this time, the amplitude of the required voltage vector correction amount is the smallest. Decouple the voltage vector correction amount Δu dq to the d and q axis voltages according to the following formula.

[0177]

[0178]

[0179] where, θ dq is the tangent angle of the q-axis current i q and the d-axis current i d .

[0180] Step 9: Input the given motor speed ω ref into the first subtractor; the given motor speed ω ref is a constant directly input externally.

[0181] Step 10: The first subtractor calculates the speed difference Δω based on the given motor speed ω ref input in Step 9 and the output speed ω of the synchronous reluctance motor, and outputs the speed difference Δω to the speed PI regulator;

[0182] Δω = ω ref - ω (24)

[0183] Step 11: The speed PI regulator calculates T0 based on the speed difference Δω input in Step 10, outputs T0 to the first multiplier, and simultaneously outputs T0 to the maximum torque per ampere (MTPA) module. The calculation formula for T0 is as follows:

[0184]

[0185] Step 12: The first multiplier calculates i 2 ω in t input in Step 7 and T0 input in Step 11 through Equation (19), and outputs i q0 , and outputs i q0 to the second subtractor.

[0186] Step 13: The second subtractor calculates the q-axis given current i qc based on i q0 input in Step 7 and i qref input in Step 12 through Equation (17), and outputs i qref to the third subtractor.

[0187] Step 14: The third subtractor calculates Δi q based on i qref input in Step 3 and i q input in Step 13, and outputs Δi q to the current controller;

[0188] Δi q = i qref - i q (26)

[0189] Step 15: The maximum torque per ampere (MTPA) module calculates the d-axis given current i dref based on T0 input in Step 11, and outputs i dref to the fourth subtractor;

[0190]

[0191] Step 16: The fourth subtractor calculates Δi based on the i input in step 3 d and the i input in step 15 dref , and outputs Δi d to the current controller; d

[0192] Δi d = i dref - i d (28)

[0193] Step 17: The current controller calculates the initial q-axis voltage u q and the initial d-axis voltage u d based on the Δi input in step 14 q0 and the Δi input in step 16 d0 , and outputs the initial q-axis voltage u q0 to the fifth subtractor, and at the same time outputs the d-axis voltage u d0 to the sixth subtractor;

[0194] u q0 = K p Δi q + K i ∫Δi q dt (29)

[0195] u d0 = K p Δi d + K i ∫Δi d dt (30)

[0196] where K p is the proportional coefficient of the current controller, and K i is the integral coefficient of the current controller.

[0197] Step 18: The fifth subtractor calculates the q-axis voltage u q based on the Δu input in step 8 q0 and the u input in step 17 q , and outputs u q to the IPark transformation module;

[0198] u q = u q0 - Δu q (31)

[0199] Step 19: The sixth subtractor calculates based on the Δu input in step 8 d and the u input in step 17​d0 , the d-axis voltage u is calculated d , and u d is output to the IPark transformation module;

[0200] u d = u d0 - Δu d (32)

[0201] Step 20: The IPark transformation module calculates the α-axis voltage component u q input in step 18 and the u d input in step 19. As Figure 8 shown, the β-axis voltage component u α and the β-axis voltage component u β are calculated. The calculation formulas are as follows:

[0202]

[0203] u a and u β are output to the space vector pulse width modulation module (SVPWM).

[0204] Step 21: The space vector pulse width modulation module (SVPWM) determines the sector where the synthesized vector is located, calculates the action time of each adjacent vector, and the conduction time and duty cycle of the six-bridge arm based on the u a and u β input in step 20. As Figure 9 shown, six PWM signals are obtained and output to the three-phase inverter.

[0205] Step 22: The three-phase inverter calculates the phase voltages of each phase of the motor based on the six PWM signals input in step 21 and the DC bus voltage value u dc input in step 5. The calculation formulas are as follows:

[0206]

[0207] Among them, u an is the phase voltage of phase A of the synchronous reluctance motor, u bn is the phase voltage of phase B of the synchronous reluctance motor, u cn is the phase voltage of phase C of the synchronous reluctance motor; the on-off states of the switching elements in the three-phase inverter circuit are defined as follows:

[0208]

[0209] The phase voltages of phases A, B, and C, u an , u bn , u cnThe output is sent to the synchronous reluctance motor to achieve the ultimate electrolytic capacitor-free synchronous reluctance motor drive control.

[0210] Experiment 1:

[0211] According to Example 1, under the given rotational speeds of low speed (1rpm), medium speed (1500rpm), and high speed (3000rpm), and the load torques of no-load (0Nm), half-load (1.2Nm), and full-load (2.4Nm), respectively, the synchronous reluctance motor of the electrolytic capacitor-free synchronous reluctance motor variable-frequency drive system of Example 1 is simulated (such as the "introduced" item in the following table), and compared with the traditional small-capacity bus capacitor drive system (such as the "not introduced" item in the following table), and the comparative simulation results are shown in the following table.

[0212]

[0213]

[0214] It can be seen from the table that after using small-capacitance film capacitors to replace traditional large-capacity electrolytic capacitors and introducing the control method of the variable-frequency drive system of the synchronous reluctance motor without electrolytic capacitors, the system can effectively reduce the total distortion rate of the input current and improve the input side power factor, so that it meets the national harmonic standard EN 61000 3-2. At the same time, this method effectively reduces the fluctuation of the motor speed and the torque fluctuation to a certain extent, improves the steady-state error of the motor speed and the system stability, and achieves the purpose of the invention.

[0215] Finally, it should be noted that the above examples are only some specific embodiments of the present invention. Obviously, the present invention is not limited to the above embodiments, and there are many variations. All variations that can be directly derived or associated with the content disclosed by a person skilled in the art should be considered as the protection scope of the present invention.

Claims

1. An electrolytic-capacitorless synchronous reluctance motor variable-frequency drive system, comprising a synchronous reluctance motor, characterized in that: The output of the synchronous reluctance motor is respectively connected to the inputs of the first subtractor and the current control module based on power balance; the output of the single-phase input power supply module is respectively connected to the inputs of the input voltage phase-locked loop, the current control module based on power balance, the power compensation module based on bus voltage control, and the single-phase rectifier. The output of the single-phase rectifier is respectively connected to the inputs of the three-phase inverter and the power compensation module based on bus voltage control after being connected in parallel with a thin-film capacitor; the output of the three-phase inverter is respectively connected to the inputs of the synchronous reluctance motor and the Clark transformation module; the output of the Clark transformation module is connected to the input of the Park transformation module, and the output of the Park transformation module is respectively connected to the inputs of the third subtractor, the fourth subtractor, the current control module based on power balance, and the power compensation module based on bus voltage control; The output of the input voltage phase-locked loop is respectively connected to the inputs of the current control module based on power balance and the power compensation module based on bus voltage control; the output of the current control module based on power balance is respectively connected to the inputs of the first multiplier and the second subtractor; the output of the power compensation module based on bus voltage control is respectively connected to the inputs of the fifth subtractor and the sixth subtractor; The output of the first subtractor is connected to the input of the speed PI regulator. The output of the speed PI regulator is respectively connected to the inputs of the first multiplier and the maximum torque per ampere (MTPA) module. The output of the first multiplier is connected to the input of the second subtractor. The output of the second subtractor is connected to the input of the third subtractor. The output of the third subtractor is connected to the input of the current controller; The output of the maximum torque per ampere (MTPA) module is connected to the input of the fourth subtractor. The output of the fourth subtractor is connected to the input of the current controller; the output of the current controller is respectively connected to the inputs of the fifth subtractor and the sixth subtractor. The outputs of the fifth subtractor and the sixth subtractor are both connected to the input of the IPark transformation module. The output of the IPark transformation module is connected to the input of the three-phase inverter after passing through the space vector pulse width modulation module (SVPWM).

2. A control method for a synchronous reluctance motor using the electrolytic-capacitorless synchronous reluctance motor variable frequency drive system as claimed in claim 1, characterized in that: Step 1: Collect the two-phase voltages of the synchronous reluctance motor through two-phase sampling resistors. According to the formula i = u / R and i a +i b +i c = 0, obtain the two-phase actual currents i a 、i b and i c in the three-phase coordinate system and input them into the Clark transformation module; Step 2: The actual current i of two phases in the three-phase coordinate system a 、i b and i c The α-axis current component i in the actual stationary two-phase coordinate system is obtained by Clark transformation α , β-axis current component i β , and output to the Park transformation module; Step 3, the α-axis current component i a , and the β-axis current component i β are transformed by the Park transformation to obtain the d-axis current i d and the q-axis current i q . The q-axis current i q is respectively output to the third subtractor and the power compensation module based on the bus voltage control, and the d-axis current i d is respectively output to the fourth subtractor, the current control module based on power balance, and the power compensation module based on the bus voltage control; Step 4: Input the u output from the single-phase input power supply module in into the input voltage phase-locked loop, the current control module based on power balance, the power compensation module based on bus voltage control, and the single-phase rectifier respectively, and input the i output from the single-phase input power supply module in into the current control module based on power balance respectively; Step 5: The single-phase rectifier obtains the DC-side bus voltage value u according to Equation (3). dc And outputs them to the three-phase inverter and the power compensation module based on bus voltage control respectively. Among them, U in is the amplitude of the input voltage u in ω in is the angular frequency of the input voltage u in θ m is the starting conduction angle of the rectifier diode, and U dcmin is the minimum value of the DC-side bus voltage value u dc ; Step 6: The input voltage phase-locked loop calculates and obtains the phase information θ of the input voltage according to u input in Step 4 in and outputs them to the current control module based on power balance and the power compensation module based on bus voltage control respectively; in ​ Step 7: The current control module based on power balance calculates sin d according to the d-axis current i in input in Step 3, the input voltage u in and the input current i in input in Step 4, the phase information θ 2 of the input voltage input in Step 6, and the output speed ω of the synchronous reluctance motor. After calculation, sin in ω qc t and i qc are obtained; then i 2 ω in t is output to the first multiplier; Step 8: The power compensation module based on bus voltage control is based on the d-axis current i input in step 3. d and q-axis current i q 、The input voltage u in step 4 in , the DC bus voltage value u entered in step 5 dc 、The phase information of the input voltage in step 6 is θ in , Δu is calculated d and Δu q , and Δu q Output to the fifth subtractor, Δu d Output to the sixth subtractor; Step 9: Input the given motor speed ω ref into the first subtractor; Step 10: The first subtractor obtains the speed difference Δω according to the following formula and outputs it to the speed PI regulator; Δω = ω ref -ω (24) where ω is the output speed of the synchronous reluctance motor; Step 11: The speed PI regulator obtains T0 according to the following formula and outputs it to the first multiplier and the maximum torque per ampere (MTPA) module respectively: as follows: where, L d is the direct-axis inductance, L q is the quadrature-axis inductance, ω e is the motor speed, p n is the number of pole pairs, U in is the magnitude of the input voltage u in and I in is the magnitude of the input current i in ; Step 12. The first multiplier calculates i according to formula (19) q0 and outputs it to the second subtractor: Step 13: The second subtractor calculates the q-axis reference current i according to formula (17) qref and outputs it to the third subtractor: Among them, C dc is the bus capacitor; Step 14: The third subtractor calculates Δi according to Equation (26) q , and outputs Δi q to the current controller: Δi q = i qref - i q (26) Step 15: The maximum torque per ampere (MTPA) module calculates the d-axis reference current i according to Equation (27) dref and outputs it to the fourth subtractor: Step 16: The fourth subtractor calculates Δi according to Equation (28) d and outputs it to the current controller. Δi d = i dref - i d (28) Step 17: The current controller calculates the initial q-axis voltage u q0 and the initial d-axis voltage u d0 respectively according to Equations (29) and (30), outputs the initial q-axis voltage u q0 to the fifth subtractor, and outputs the d-axis voltage u d0 to the sixth subtractor; u q0 = K p Δi q + K i ∫Δi q dt (29) u d0 = K p Δi d + K i ∫Δi d dt (30) Among them, K p is the proportional coefficient of the current controller, and K i is the integral coefficient of the current controller; Step 18: The fifth subtractor calculates the q-axis voltage u according to Equation (31) q and outputs it to the IPark transformation module; u q = u q0 - Δu q (31) Step 19: The sixth subtractor calculates the d-axis voltage u according to Equation (32) d and outputs it to the IPark transformation module: u d = u d0 - Δu d (32) Step 20: The IPark transformation module calculates the α-axis voltage component u α and the β-axis voltage component u β : Output u a and u β to the space vector pulse width modulation module (SVPWM); Step 21: The Space Vector Pulse Width Modulation module (SVPWM) obtains six-channel PWM signals based on u α and u β input in Step 20 by determining the sector where the synthesized vector is located, calculating the action time of each adjacent vector, and the conduction time and duty cycle of the six-bridge arms, and outputs the six-channel PWM signals to the three-phase inverter; Step 22: The three-phase inverter calculates the phase voltages of the motor according to the six-way PWM signals input in Step 21 and the DC bus voltage value u input in Step 5 dc as follows according to Equation (34): where, u an is the phase voltage of phase A of the synchronous reluctance motor, u bn is the phase voltage of phase B of the synchronous reluctance motor, u cn is the phase voltage of phase C of the synchronous reluctance motor; the on / off states of the switching elements in the three-phase inverter circuit are defined as follows: Output the three-phase phase voltages u of A, B, and C an 、u bn 、u cn to the synchronous reluctance motor to achieve drive control of a synchronous reluctance motor without electrolytic capacitors.

3. The electrolytic-capacitorless synchronous reluctance motor variable frequency drive system as claimed in claim 2, characterized in that: The calculation process of the input voltage phase-locked loop described in step 6 is specifically as follows: The phase information θ in of the input voltage u in is obtained through trigonometric transformation to get cos(θ in ). Then, after multiplying the amplitude U in of the input voltage u in by cos(θ in ), it is filtered by a low-pass filter and regulated by a PI regulator to obtain the input voltage angular frequency reference value ω inref . By subtracting the actual input voltage angular frequency ω inref from the input voltage angular frequency reference value ω in , the input voltage angular frequency error Δω in is obtained. Finally, through an integration link, the phase information θ in of the input voltage is obtained.

4. The electrolytic-capacitorless synchronous reluctance motor variable frequency drive system as claimed in claim 3, characterized in that: The specific calculation process of the current control module based on power balance in step 7 is as follows: Input voltage u in and input current i in are expressed by Equations (4) and (5) respectively as follows: u in = U in sinω in t (4) where θ is the conduction angle of the rectifier diode; let (π - θ) / 2 = θ m ; The power P on the input side in As shown in Equation (6): where θ m is the starting conduction angle of the rectifier diode; Assume that the input is a unity power factor, and the DC bus voltage value is as shown in formula (7): where U dcmin is the minimum value of the bus voltage; Capacitive current i dc is as follows: where sgn(x) is the sign function: The power P on the bus capacitor dc is as follows: Instantaneous power consumption P of the motor m As shown in Equation (11): P m = P cu + P react + P mech (11) Among them, P cu is the copper loss of the motor, P react is the reactive power, P mech is the mechanical output power, and the expression is as follows: P mech = 1.5ω e i q p n (L d -L q )i d (14) wherein, R is the stator resistance; ignoring the influence of the copper loss P cu and the reactive power P react of the motor, the instantaneous power consumption P m of the motor is as follows: P in -P dc =P m (15) By simultaneously solving equations (6), (11) to (15), we get: According to Equation (16), the q-axis reference current i input to the third subtractor qref is expressed as follows: Among them, the q-axis current compensation component corresponding to the capacitor power is i qc , and the expression is as follows: The q-axis current corresponding to the input power is i q0 , and the expression is as follows: Based on equations (17) to (19), the reference value of the q-axis current based on power balance is obtained as: i qref = i q0 -i qc (20) In Equation (19), set as the output T0 of the speed PI regulator: i q0 = T0sin 2 ω in t (21).

5. The variable-frequency drive system for an electrolytic-capacitorless synchronous reluctance motor according to claim 4, wherein: The implementation process of the power compensation module based on bus voltage control described in step 8 is as follows: the input voltage u in The input voltage phase information θ obtained by the input voltage phase-locked loop in , after being processed by trigonometric functions and absolute values, is multiplied by the input voltage amplitude U in to obtain the ideal bus voltage Then calculate the ideal bus voltage and the difference from the DC-side bus voltage value u dc . The difference ΔP of the inverter output power is obtained through a PI regulator. Divide ΔP by and then multiply by the proportionality coefficient K c to obtain the correction amount Δu of the voltage vector dq , and then calculate Δu d and Δu q according to Equation (22): where θ dq is the tangent angle of the q-axis current i q and the d-axis current i d .

6. The variable-frequency drive system for an electrolytic-capacitorless synchronous reluctance motor according to claim 5, wherein: The Clark transformation formula is:

7. The variable-frequency drive system for an electrolytic-capacitorless synchronous reluctance motor according to claim 6, wherein: The Park transformation formula is:

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