An ultra-low phase noise microstrip oscillator

Abstract

The present invention provides a microstrip oscillator with ultra-low phase noise, which introduces a quasi-bandpass frequency selection network. The quasi-bandpass frequency selection network is formed by directly coupling the source / load of two quarter-wavelength resonators, and can achieve a quasi-second-order generalized Chebyshev bandpass frequency response. There is a transmission zero on each side of the passband, which can effectively improve the frequency selectivity and help to form a relatively high group delay. In addition, due to the special structure of the feeder, three additional transmission zeros are introduced, greatly improving the out-of-band rejection performance. The test results of the embodiment show that: the oscillation frequency is 1.976 GHz, the output power is 9.61 dBm, the phase noise is -129.85 dBc / Hz@100 kHz, the out-of-band second harmonic rejection reaches 48.21 dBc, and the oscillator quality factor at 100 kHz away from the carrier is -200.66 dBc / Hz.

Description

Technical Field

[0001] The present invention belongs to the field of communication technologies, and particularly relates to a microstrip oscillator with ultra-low phase noise. Background Art

[0002] In recent years, with the rapid development of personal mobile communications, the microwave and wireless markets have attracted much attention. A microwave oscillator is an essential component of a frequency generation source. As a key module in circuits such as phase-locked loops, frequency synthesizers, and clock recovery, it is widely used in electronic systems such as mobile phones, satellite communication terminals, mechanisms, radars, missile guidance systems, military communication systems, digital wireless communications, optical multiplexers, and optical transmitters. As a reference source for various frequency sources and a key device for generating time-frequency benchmarks, the phase noise of a microwave oscillator has increasingly become a key factor restricting the performance of various circuits and systems, having a decisive impact on the performance, size, weight, and cost of electronic systems, and being a difficult point in microwave circuit design and integration. Therefore, researching microwave oscillators with low phase noise is of extremely important significance. Summary of the Invention

[0003] The object of the present invention is to overcome the deficiency of the poor phase noise of existing microwave oscillators, and provide a microstrip oscillator with ultra-low phase noise. This microwave oscillator is based on a microstrip structure and has the advantages of low phase noise, high output power, good harmonic suppression, easy processing, and low cost.

[0004] The microstrip oscillator described in the present invention is as Figure 1As shown in the figure, it is characterized in that: the base terminal of the transistor (BJT) is connected to the first line segment (T1), and the collector terminal is connected to the second line segment (T2); a DC power supply is loaded onto the bias line segment (PT), and a grounding capacitor (SC) is connected to the bias line segment (PT), and it is connected to the first line segment (T1) through a resistor (R), and the bias line segment (PT) is simultaneously connected to the second line segment (T2) to supply power to the transistor (BJT); the left end of the first line segment (T1) is connected to the right end of the first capacitor (C1); the left end of the first capacitor (C1) is connected to the right end of the first terminal short - circuit line segment (ST1), and the first terminal short - circuit line segment (ST1) is grounded through the first via (S1); a first open stub (OT1) is loaded at a point in the middle of the first terminal short - circuit line segment (ST1); one end of the first terminal short - circuit line segment (ST1) is capacitively coupled to the first terminal short - circuit resonator (SR1), and the first terminal short - circuit resonator (SR1) is grounded through the second via (S2); the right end of the second line segment (T2) is connected to the left end of the second capacitor (C2); the right end of the second capacitor (C2) is connected to the left end of the second terminal short - circuit line segment (ST2), and the second terminal short - circuit line segment (ST2) is grounded through the third via (S3); a second open stub (OT2) is loaded at a point in the middle of the second terminal short - circuit line segment (ST2); one end of the second terminal short - circuit line segment (ST2) is capacitively coupled to the second terminal short - circuit resonator (SR2), and the second terminal short - circuit resonator (SR2) is grounded through the fourth via (S4); the first terminal short - circuit line segment (ST1) is coupled to the second terminal short - circuit line segment (ST2) through an interdigital structure (CF); the first terminal short - circuit resonator (SR1) is capacitively coupled to the second terminal short - circuit resonator (SR2); energy is output through an output stub (OUT) loaded at a point in the middle of the second terminal short - circuit line segment (ST2).

[0005] Part of the structure of the microstrip oscillator according to the present invention is as shown in Figure 2 the figure, and it consists of the following: the first terminal short - circuit line segment (ST1) is grounded through the first via (S1); one end of the first terminal short - circuit line segment (ST1) is capacitively coupled to the first terminal short - circuit resonator (SR1), and the first terminal short - circuit resonator (SR1) is grounded through the second via (S2); the second terminal short - circuit line segment (ST2) is grounded through the third via (S3); one end of the second terminal short - circuit line segment (ST2) is capacitively coupled to the second terminal short - circuit resonator (SR2), and the second terminal short - circuit resonator (SR2) is grounded through the fourth via (S4); the first terminal short - circuit line segment (ST1) is coupled to the second terminal short - circuit line segment (ST2) through an interdigital structure (CF); the first terminal short - circuit resonator (SR1) is capacitively coupled to the second terminal short - circuit resonator (SR2); for the sake of simplicity, this part of the structure is called a quasi - band - pass frequency - selective network. This quasi - band - pass frequency - selective network directly determines the performance of the microstrip oscillator according to the present invention.

[0006] The structural parameters of the quasi-bandpass frequency-selective network are labeled as Figure 3 shown. Except for the cross-finger structure (CF), the rest of the quasi-bandpass frequency-selective network is symmetric about the vertical plane. Among them, l1 represents the line length of the gap coupling between the first terminal short-circuit resonator (SR1) and the second terminal short-circuit resonator (SR2); l2 represents the line length of the first terminal short-circuit resonator (SR1), and also represents the line length of the second terminal short-circuit resonator (SR2); l3 represents the line length of the cross-finger structure (CF); l4 represents the line length between the end of the second terminal short-circuit stub (ST2) and the starting position of the coupling with the second terminal short-circuit resonator (SR2); l5 represents the horizontal part line length of the coupling between the second terminal short-circuit stub (ST2) and the second terminal short-circuit resonator (SR2); l6 represents the line length between the access position of the first terminal short-circuit stub (ST1) to the first metallization via hole (S1); w1 represents the line width of the first terminal short-circuit resonator (SR1); w2 represents the line width of the second terminal short-circuit stub (ST2); w3 represents the line width of the cross-finger structure (CF); g1 represents the gap width between the first terminal short-circuit resonator (SR1) and the second terminal short-circuit resonator (SR2); g2 represents the gap width between one end of the first terminal short-circuit stub (ST1) and the first terminal short-circuit resonator (SR1); d represents the diameter of the second metallization via hole (S2).

[0007] The quasi-bandpass frequency-selective network can theoretically achieve a second-order bandpass frequency response, as Figure 4 shown, with one transmission zero on each side of its passband. The corresponding normalized coupling matrix [M] is as follows:

[0008]

[0009] Among them, S represents the first terminal short-circuit stub (ST1), L represents the second terminal short-circuit stub (ST2), 1 represents the first terminal short-circuit resonator (SR1), and 2 represents the second terminal short-circuit stub (ST2). m ij (where i and j respectively take S, L, 1, and 2) represents the normalized coupling coefficient between the first terminal short-circuit stub (ST1), the second terminal short-circuit stub (ST2), the first terminal short-circuit resonator (SR1), and the second terminal short-circuit resonator (SR2).

[0010] Through the following formula, the coupling coefficient k can be determined by the normalized coupling coefficient m ij ij .

[0011]

[0012] External quality factor Q​e It can be determined from the normalized coupling coefficient through the following formula.

[0013]

[0014] where Q eS represents the external quality factor between the first-terminal shorted stub (ST1) and the first-terminal shorted resonator (SR1). Q eL represents the external quality factor between the second-terminal shorted stub (ST2) and the second-terminal shorted resonator (SR2). FBW is the relative bandwidth.

[0015] According to the oscillation frequency requirements of the microstrip oscillator, the initial structural parameter values of the quasi-bandpass frequency-selective network are determined using traditional filter synthesis methods. The initial electrical length values of the first-terminal shorted resonator (SR1) and the second-terminal shorted stub (ST2) are set to a quarter-wavelength corresponding to the oscillation frequency; the initial values of the other parameters g1 and g2 are determined by comparing the ideal and actual coupling coefficients and the ideal and actual external quality factors. Based on these initial values, the structural parameter values are further adjusted through simulation optimization to minimize the insertion loss and maximize the group delay at the oscillation frequency, and the final structural parameter values can be determined.

[0016] The simulation results of the quasi-bandpass frequency-selective network are as Figure 5 shown. There are five transmission zeros at finite frequencies, labeled as f TZ1 , f TZ2 , f TZ3 , f TZ4 and f TZ5 respectively. Among them, the two resonators, the first-terminal shorted resonator (SR1) and the second-terminal shorted stub (ST2), are directly coupled to the source / load to form f TZ2 and f TZ3 . These two transmission zeros are located on each side of the passband, effectively improving the frequency selectivity and at the same time contributing to the formation of a relatively high group delay. The remaining three transmission zeros are located in the stopband, greatly improving the out-of-band rejection ability. These three transmission zeros are related to the special structures of the first-terminal shorted stub (ST1) and the second-terminal shorted stub (ST2), and the schematic diagram of the equivalent circuit is as Figure 6 shown. The input admittance Y in is expressed as

[0017] Y in = -jY1 cotθ1 (4)

[0018] where Y1 and θ1 represent the characteristic admittance and the electrical length respectively. A transmission zero is generated when Y in = ∞.

[0019] The beneficial effects of the oscillator of the present invention are as follows: low phase noise, high power output, good out-of-band harmonic suppression, easy processing, and low cost, etc. BRIEF DESCRIPTION OF THE DRAWINGS

[0020] Figure 1 : Schematic diagram of a microstrip oscillator;

[0021] Figure 2 : Schematic diagram of a quasi-bandpass frequency selection network;

[0022] Figure 3 : Schematic diagram of the structural parameters of a quasi-bandpass frequency selection network;

[0023] Figure 4 : Ideal second-order generalized Chebyshev bandpass frequency response diagram;

[0024] Figure 5 : Simulation result diagram of a quasi-bandpass frequency selection network;

[0025] Figure 6 : Equivalent circuit diagram;

[0026] Figure 7 : Actual coupling coefficient k 12 Curve diagram showing the variation with the gap g1;

[0027] Figure 8 : Actual external quality factor Q e Curve diagram showing the variation with the gap g2;

[0028] Figure 9 : S-parameter simulation result diagram of Example 1;

[0029] Figure 10 : Group delay simulation result diagram of Example 1;

[0030] Figure 11 : Diagram showing the influence of the structural parameter l4 of Example 1 on its performance;

[0031] Figure 12 : Diagram showing the influence of the number of sections N of the cross-finger structure (CF) of Example 1 on its performance;

[0032] Figure 13 : Phase noise test result diagram of Example 2;

[0033] Figure 14 : Output spectrum test result diagram of Example 2. DETAILED DESCRIPTION OF THE INVENTION

[0034] In order to embody the creativity and novelty of the present invention, the following will be described in combination with the drawings and specific embodiments, but the implementation manners of the present invention are not limited thereto.

[0035] In the embodiment, a common microstrip substrate is selected, with a relative dielectric constant of 3.66 and a thickness of 0.508 mm.

[0036] Embodiment 1 is a quasi-bandpass frequency selection network. If the oscillation frequency of the microstrip oscillator is set to 2.0 GHz, Embodiment 1 uses the filter design method to determine the initial values of its structural parameters. The center frequency f0 of Embodiment 1 is 2.0 GHz, and the relative bandwidth FBW is 2%. The normalized coupling coefficient matrix of the quasi-bandpass frequency selection network is

[0037]

[0038] The coupling coefficient matrix can be calculated by formula (2) as

[0039]

[0040] From this, the ideal coupling coefficient k of the first terminal short-circuit resonator (SR1) and the second terminal short-circuit resonator (SR2) can be obtained 12 = 0.03094. The ideal external quality factor Q is calculated by formula (3) as e = Q eS = Q eL = 20.89.

[0041] Figure 7 The curve of the actual coupling coefficient k varying with the gap g1 is given in 12 The curve of the actual external quality factor Q varying with the gap g2 is given in Figure 8 Using the previously calculated ideal coupling coefficient value and ideal external quality factor value, the corresponding g1 and g2 values, namely g1 = 1.3 mm and g2 = 0.1 mm, can be determined from these curves as their initial values. e Based on the initial values of the structural parameters obtained previously, the structural parameter values are adjusted at the center frequency f0 to minimize the insertion loss and maximize the group delay of the quasi-bandpass frequency selection network. Finally, a set of structural parameter values (unit: mm) is determined as: l1 = 5.80, l2 = 23.58, l3 = 2.00, l4 = 11.20, l5 = 4.80, l6 = 31.56, w1 = 0.80, w2 = 0.60, w3 = 0.22, g1 = 1.65, g2 = 0.28, and d = 0.30. The simulation results of Embodiment 1 are as shown in

[0042] The minimum in-band insertion loss is 4.51 dB, and the peak group delay of 11.25 ns is obtained at 2.004 GHz. The five transmission zeros are located at 0.54 GHz, 1.87 GHz, 2.09 GHz, 2.63 GHz, and 3.61 GHz respectively, and the suppression at 2f0 is -28.79 dB. Figure 9 and 10 respectively.

[0043] To show the performance flexibility of Example 1, the influence of the typical structural parameter l4 on the performance of Example 1 is given in Figure 11 . When l4 changes, it mainly affects three transmission zeros, namely f TZ1 , f TZ4 and f TZ5 , and has no influence on the transmission zeros f TZ2 and f TZ3 . This fully shows that the three transmission zeros f TZ1 , f TZ4 and f TZ5 are related to the special structures of the first terminal short - circuit stub (ST1) and the second terminal short - circuit stub (ST2). When l4 increases, f TZ1 , f TZ4 and f TZ5 all move towards the low - frequency direction. Based on the equivalent circuit of Figure 6 , when Y1 = 0.014 S and θ1 = 0.762 rad, the transmission zeros calculated using formula (4) are 2.706 GHz and 5.413 GHz respectively, while the fourth transmission zero f TZ4 = 2.63 GHz and the fifth transmission zero f TZ5 = 3.59 GHz obtained by simulation effectively verify the effectiveness of the equivalent circuit and reveal the generation mechanism of the transmission zeros.

[0044] Figure 12 shows the influence of the number of sections N of the cross - finger structure (CF) on the performance of the quasi - band - pass frequency - selective network. When N = 0, only the three transmission zeros f TZ1 , f TZ2 and f TZ3 exist. When N = 1, four transmission zeros appear, namely f TZ1 , f TZ2 , f TZ3 and f TZ4 . When N continues to increase, the coupling strength between the first terminal short - circuit stub (ST1) and the second terminal short - circuit stub (ST2) also increases, and at this time the fifth transmission zero f TZ5 appears. Therefore, adjusting N can effectively control the performance of the quasi - band - pass frequency - selective network.

[0045] Example 2 is a microstrip oscillator based on Example 1, and the test results are as shown in Figure 13 and 14As shown. When the DC bias voltage is 2.7 V and the power supply current is 12 mA, the measured oscillation frequency is 1.976 GHz, the output power is 9.61 dBm, and the phase noise is measured to be -129.85 dBc / Hz @ 100 kHz. At the same time, due to the good out-of-band performance of the quasi-bandpass frequency selection network, the second harmonic suppression of the second embodiment reaches 48.21 dBc. The oscillator quality factor of the second embodiment at 100 kHz away from the carrier is -200.66 dBc / Hz. The second embodiment exceeds the similar indicators of other S-band microstrip oscillators reported publicly at home and abroad in terms of phase noise, output power, second harmonic suppression, and oscillator quality factor, showing significant technological progress.

[0046] The above-listed embodiments fully illustrate that the microstrip oscillator described in the present invention has the advantages of low phase noise, high output power, good out-of-band harmonic suppression, easy debugging, etc., and shows significant technological progress. Those of ordinary skill in the art will realize that the embodiments described here are for helping readers understand the principles of the present invention, and it should be understood that the protection scope of the present invention is not limited to such specific statements and embodiments. Those of ordinary skill in the art can make various other specific deformations and combinations that do not deviate from the essence of the present invention based on the technical revelations disclosed in the present invention, and these deformations and combinations are still within the protection scope of the present invention.

Claims

1. A microstrip oscillator, characterized in that: The base terminal of the bipolar junction transistor (BJT) is connected to the first line section (T1), and the collector terminal is connected to the second line section (T2); a DC power supply is loaded onto the bias line section (PT), and a ground capacitor (SC) is connected to the bias line section (PT), and is connected to the first line section (T1) through a resistor (R), and the bias line section (PT) is simultaneously connected to the second line section (T2) to supply power to the bipolar junction transistor (BJT); the left end of the first line section (T1) is connected to the right end of the first capacitor (C1); the left end of the first capacitor (C1) is connected to the right end of the first terminal short - circuit line section (ST1), and the first terminal short - circuit line section (ST1) is grounded through the first via (S1); a first open - ended stub (OT1) is loaded at a point in the middle of the first terminal short - circuit line section (ST1); one end of the first terminal short - circuit line section (ST1) is coupled to the first terminal short - circuit resonator (SR1) by a slot; the first terminal short - circuit resonator (SR1) is grounded through the second via (S2); the right end of the second line section (T2) is connected to the left end of the second capacitor (C2); the right end of the second capacitor (C2) is connected to the left end of the second terminal short - circuit line section (ST2), and the second terminal short - circuit line section (ST2) is grounded through the third via (S3); a second open - ended stub (OT2) is loaded at a point in the middle of the second terminal short - circuit line section (ST2); one end of the second terminal short - circuit line section (ST2) is coupled to the second terminal short - circuit resonator (SR2) by a slot; the second terminal short - circuit resonator (SR2) is grounded through the fourth via (S4); the first terminal short - circuit line section (ST1) is coupled to the second terminal short - circuit line section (ST2) through an interdigital structure (CF); the first terminal short - circuit resonator (SR1) is coupled to the second terminal short - circuit resonator (SR2) by a slot; the energy is output through an output stub (OUT) loaded at a point in the middle of the second terminal short - circuit line section (ST2).

2. The microstrip oscillator according to claim 1, wherein the quasi - band - pass frequency - selective network included therein is configured as follows: the first terminal short - circuit line section (ST1) is grounded through the first via (S1); one end of the first terminal short - circuit line section (ST1) is coupled to the first terminal short - circuit resonator (SR1) by a slot, and the first terminal short - circuit resonator (SR1) is grounded through the second via (S2); the second terminal short - circuit line section (ST2) is grounded through the third via (S3); one end of the second terminal short - circuit line section (ST2) is coupled to the second terminal short - circuit resonator (SR2) by a slot, and the second terminal short - circuit resonator (SR2) is grounded through the fourth via (S4); the first terminal short - circuit line section (ST1) is coupled to the second terminal short - circuit line section (ST2) through an interdigital structure (CF); the first terminal short - circuit resonator (SR1) is coupled to the second terminal short - circuit resonator (SR2) by a slot.

3. The microstrip oscillator according to claim 1, wherein the quasi - band - pass frequency - selective network included therein can theoretically achieve a second - order band - pass frequency response, with one transmission zero on each side of its passband; the corresponding normalized coupling matrix [M] is as follows: Among them, Let S denote the first-terminal short-circuited stub (ST1), L denote the second-terminal short-circuited stub (ST2), 1 denote the first-terminal short-circuited resonator (SR1), and 2 denote the second-terminal short-circuited stub (ST2); m ij (where i and j each take S, L, 1, and 2 respectively) represents the normalized coupling coefficient between the first-terminal short-circuited stub (ST1), the second-terminal short-circuited stub (ST2), the first-terminal short-circuited resonator (SR1), and the second-terminal short-circuited resonator (SR2); The coupling coefficient k can be determined through the normalization coupling coefficient m by the following formula ij ij ;​ External quality factor Q e can be determined from the normalized coupling coefficient by the following formula: Among them, Q eS represents the external quality factor between the first terminal short circuit stub (ST1) and the first terminal short circuit resonator (SR1), Q eL represents the external quality factor between the second terminal short circuit stub (ST2) and the second terminal short circuit resonator (SR2), and FBW is the relative bandwidth.

4. The microstrip oscillator according to claim 1, wherein the quasi-bandpass frequency selection network included therein has five transmission zeros at finite frequencies, which are respectively labeled as f TZ1 , f TZ2 , f TZ3 , f TZ4 and f TZ5 ; wherein, Two resonators, the first-terminal shorted resonator (SR1) and the second-terminal shorted stub (ST2), are directly coupled to the source / load to form f TZ2 and f TZ3 ; These two transmission zeros are located on each side of the passband respectively, effectively improving the frequency selectivity and at the same time helping to form a relatively high group delay; The remaining three transmission zeros are located in the stopband, greatly improving the out-of-band rejection ability; These three transmission zeros are related to the special structures of the first-terminal shorted stub (ST1) and the second-terminal shorted stub (ST2); The input admittance Y in is expressed as Y in = -jY1 cotθ1 where Y1 and θ1 represent the characteristic admittance and electrical length respectively; a transmission zero occurs when Y in = ∞.

5. The design method of the quasi-bandpass frequency selection network included in the microstrip oscillator according to claim 1 is as follows: According to the oscillation frequency requirement of the microstrip oscillator, the initial structural parameter values of the quasi-bandpass frequency selection network are determined using the traditional filter synthesis method; the initial electrical length values of the first terminal short-circuit resonator (SR1) and the second terminal short-circuit stub (ST2) are set to a quarter wavelength corresponding to the oscillation frequency; the initial values of other parameters g1 and g2 are determined by comparing the ideal and actual coupling coefficients and the ideal and actual external quality factors; based on these initial values, the structural parameter values are further adjusted through simulation optimization to minimize the insertion loss and maximize the group delay of the quasi-bandpass frequency selection network at the oscillation frequency, and the final structural parameter values can be determined.

6. The structural parameters of the quasi-bandpass frequency selection network included in the microstrip oscillator according to claim 1 are as follows: l1 represents the line length of the slot coupling between the first terminal short-circuit resonator (SR1) and the second terminal short-circuit resonator (SR2), l2 represents the line length of the first terminal short-circuit resonator (SR1), and also represents the line length of the second terminal short-circuit resonator (SR2), l3 represents the line length of the cross-finger structure (CF), l4 represents the line length of the second terminal short-circuit stub (ST2) from the end to the position where it starts to couple with the second terminal short-circuit resonator (SR2), l5 represents the horizontal part line length of the coupling between the second terminal short-circuit stub (ST2) and the second terminal short-circuit resonator (SR2), l6 represents the line length of the first terminal short-circuit stub (ST1) from the access position of the cross-finger structure (CF) to the first metallization via (S1), w1 represents the line width of the first terminal short-circuit resonator (SR1), w2 represents the line width of the second terminal short-circuit stub (ST2), w3 represents the line width of the cross-finger structure (CF), g1 represents the slot width between the first terminal short-circuit resonator (SR1) and the second terminal short-circuit resonator (SR2), g2 represents the slot width between one end of the first terminal short-circuit stub (ST1) and the first terminal short-circuit resonator (SR1), d represents the diameter of the second metallization via (S2); when l1 = 5.80 mm, l2 = 23.58 mm, l3 = 2.00 mm, l4 = 11.20 mm, l5 = 4.80 mm, l6 = 31.56 mm, w1 = 0.80 mm, w2 = 0.60 mm, w3 = 0.22 mm, g1 = 1.65 mm, g2 = 0.28 mm, and d = 0.30 mm, the minimum insertion loss within the band of the quasi-bandpass frequency selection network is 4.51 dB, the peak value of the group delay is obtained at 2.004 GHz as 11.25 ns, the five transmission zeros are respectively located at 0.54 GHz, 1.87 GHz, 2.09 GHz, 2.63 GHz, and 3.61 GHz, and the suppression at 2f0 is -28.79 dB.

7. The structural parameters of the quasi-bandpass frequency selection network included in the microstrip oscillator according to claim 1 are as follows: l1 represents the line length of the slot coupling between the first terminal short resonator (SR1) and the second terminal short resonator (SR2), l2 represents the line length of the first terminal short resonator (SR1), and also represents the line length of the second terminal short resonator (SR2), l3 represents the line length of the cross-finger structure (CF), l4 represents the line length between the end of the second terminal short line segment (ST2) and the starting coupling position with the second terminal short resonator (SR2), l5 represents the horizontal part line length of the coupling between the second terminal short line segment (ST2) and the second terminal short resonator (SR2), l6 represents the line length between the access position of the first terminal short line segment (ST1) to the cross-finger structure (CF) and the first metallized via hole (S1), w1 represents the line width of the first terminal short resonator (SR1), w2 represents the line width of the second terminal short line segment (ST2), w3 represents the line width of the cross-finger structure (CF), g1 represents the slot width between the first terminal short resonator (SR1) and the second terminal short resonator (SR2), g2 represents the slot width between one end of the first terminal short line segment (ST1) and the first terminal short resonator (SR1), d represents the diameter of the second metallized via hole (S2); when l1 = 5.80 mm, l2 = 23.58 mm, l3 = 2.00 mm, l4 = 11.20 mm, l5 = 4.80 mm, l6 = 31.56 mm, w1 = 0.80 mm, w2 = 0.60 mm, w3 = 0.22 mm, g1 = 1.65 mm, g2 = 0.28 mm and d = 0.30 mm; at a DC bias voltage of 2.7 V and a power supply current of 12 mA, the oscillation frequency of the microstrip oscillator is 1.976 GHz, the output power is 9.61 dBm, the phase noise is -129.85 dBc / Hz@100 kHz, the out-of-band second harmonic suppression reaches 48.21 dBc, and the oscillator quality factor at 100 kHz away from the carrier is -200.66 dBc / Hz.

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