A method for controlling the peak value of inductor current ripple of a totem-pole bridgeless PFC converter

Abstract

The application discloses a totem pole bridgeless PFC converter inductance current ripple peak value control method, which is based on the inductance current ripple peak value of a totem pole bridgeless PFC converter, and the switching period of a control signal is changed through a variable switching frequency technical means, so as to realize the control of the inductance current ripple peak value of the totem pole bridgeless PFC converter. Specifically, first, the real-time inductance current ripple of a fixed switching frequency is predicted, then the switching period of the inductance current ripple when the inductance current ripple is controlled at the next moment is calculated and determined according to the ripple peak value limitation requirement, then the controller generates a control signal according to the new switching period, the control signal is sent to a switching tube through a driving module, and the control of the inductance current ripple peak value of the totem pole bridgeless PFC converter at the next moment is completed.

Description

Technical Field

[0001] This invention belongs to the field of power electronics technology, and more specifically, relates to a method for controlling the peak inductor current ripple of a totem-pole bridgeless PFC (Power Factor Correction) converter. Background Technology

[0002] The emergence of power electronics technology has had a profound impact on many technological fields. The electrical energy required by most electrical equipment is obtained through AC-DC conversion. However, traditional power electronic rectifier devices generate a large amount of harmonic pollution during operation, which leads to a decrease in power quality and an increase in energy consumption.

[0003] The most common and widely used method in active harmonic suppression circuits is power factor correction technology. This technology mainly operates by controlling the circuit with pulse width modulation to adjust the phase of the input current so that it closely follows the phase of the input voltage, making the power factor close to 1, thus fundamentally solving the harmonic problem.

[0004] Among various bridgeless PFC circuits, the totem-pole bridgeless PFC converter circuit uses the fewest switching devices and has relatively low conduction losses. Furthermore, the development of wide-bandgap semiconductor materials such as silicon carbide and gallium nitride has further reduced the losses of the converter circuit. However, in higher-frequency operating environments, the totem-pole bridgeless PFC converter circuit generates more severe harmonic distortion and electromagnetic interference (EMI) problems, limiting its application. Model-predicted VSFPWM switching frequency variation is based on a current ripple prediction model. Using the peak value of the current ripple as the control object, variable switching frequency PWM control is employed, and the switching period changes in real time, thus dispersing the EMI spectrum and preventing it from concentrating near integer multiples of the fixed switching frequency, thereby suppressing EMI peaks. Summary of the Invention

[0005] The purpose of this invention is to overcome the shortcomings of the prior art and provide a method for controlling the peak value of inductor current ripple in a totem-pole bridgeless PFC converter. The method uses the peak value of the current ripple as the control object and adopts variable switching frequency PWM control to disperse the EMI spectrum and reduce the EMI peak value.

[0006] To achieve the above-mentioned objective, the present invention provides a method for controlling the peak inductor current ripple of a totem-pole bridgeless PFC converter, characterized by comprising the following steps:

[0007] (1) Based on the topology of the totem pole bridgeless PFC converter, mark the high-frequency bridge arm switches S1 and S2 and the power frequency bridge arm switches S3 and S4, and mark the boost inductor L.

[0008] (2) Collect the input voltage V of the totem pole bridgeless PFC converter at time t. in (t), Input current I in (t) and output voltage V o (t);

[0009] (3) During a fixed switching period T c Below, predict the peak inductor current ripple i of the totem-pole bridgeless PFC converter after the switching change at time t+1. pk (t+1);

[0010]

[0011] Among them, i t D represents the current flowing through the boost inductor L after the switching change at time t; t+1 The duty cycle after the switch changes at time t+1;

[0012] (4) Calculate the duty cycle D after the switch changes at time t+1. t+1 ;

[0013] (4.1) Calculate the inductor current i after the switch changes at time t+1. t+1 :

[0014]

[0015] (4.2) The peak value of the inductor current ripple i pk (t+1) introduces the inductor current i t+1 :

[0016]

[0017] (4.3) Based on the topology of the totem-pole bridgeless PFC converter, the input reference value i from the outer voltage loop to the inner current loop is given. avg ;

[0018] Take i t+1 =i avg Therefore, we get:

[0019]

[0020] (5) Calculate the peak value of the inductor current ripple i pk (t+1) Switching period T when controlled v (t+1);

[0021]

[0022] Among them, I requiredThis is a limit value for the peak value of the inductor current ripple;

[0023] (6) The controller determines the switching period T v (t+1) Generates a control signal, which is then sent to the switching transistor through the drive module, thereby completing the control of the inductor current ripple peak of the totem pole bridgeless PFC converter at the next moment.

[0024] (7) When the next sampling time arrives, return to step (2).

[0025] The objective of this invention is achieved as follows:

[0026] This invention discloses a method for controlling the peak inductor current ripple of a totem-pole bridgeless PFC converter. Based on the peak inductor current ripple of the totem-pole bridgeless PFC converter, the switching period of the control signal is changed by varying the switching frequency, thereby controlling the peak inductor current ripple of the totem-pole bridgeless PFC converter. Specifically, the real-time inductor current ripple at a fixed switching frequency is first predicted. Then, based on the ripple peak value limit requirement, the switching period when the inductor current ripple is controlled at the next moment is calculated and determined. The controller then generates a control signal based on the new switching period, which is sent to the switching transistor through the drive module to complete the control of the peak inductor current ripple of the totem-pole bridgeless PFC converter at the next moment.

[0027] Meanwhile, the totem-pole bridgeless PFC converter inductor current ripple peak control method of the present invention also has the following beneficial effects:

[0028] (1) The present invention controls the peak value of the inductor current ripple of the totem pole bridgeless PFC converter by using the variable switching frequency PWM technology. Among the many variable switching frequency technologies that can realize the totem pole bridgeless PFC converter, this technology makes full use of the degree of freedom of the peak value of the inductor current ripple to achieve precise control of the current ripple.

[0029] (2) The present invention controls the switching frequency of the totem pole bridgeless PFC converter in real time to achieve control of the peak value of the inductor current ripple. By clamping the peak value of the inductor current ripple to the maximum value of the ripple current, the switching cycle is updated to be longer and the average switching frequency is reduced, which is beneficial to reduce the switching loss of the converter's switching transistor at high switching frequency.

[0030] (3) The present invention controls the switching frequency of the totem pole bridgeless PFC converter in real time, so that the EMI spectrum concentrated near an integer multiple of the fixed switching frequency when operating at a fixed switching frequency is dispersed, thereby achieving the effect of suppressing EMI peaks. Attached Figure Description

[0031] Figure 1 It is an existing bridgeless PFC circuit for totem poles;

[0032] Figure 2 This is a flowchart of a method for controlling the peak inductor current ripple of a totem pole bridgeless PFC converter according to the present invention.

[0033] Figure 3 It refers to the location of parasitic capacitors in the existing totem pole bridgeless PFC circuit;

[0034] Figure 4 It is the inductor current ripple within a single cycle of the positive and negative half-cycles of the existing circuit.

[0035] Figure 5 These are the time-domain waveforms of the input inductor current and the fundamental current implemented according to the present invention;

[0036] Figure 6 This is a comparison between the inductor current ripple achieved according to the present invention and the predicted ripple peak value;

[0037] Figure 7 This is a block diagram of the implementation of variable switching frequency PWM according to the present invention;

[0038] Figure 8 This is a variable switching frequency current ripple waveform implemented according to the present invention;

[0039] Figure 9 This is a block diagram of the totem-pole bridgeless PFC circuit control for VSFPWM based on the model prediction implemented according to the present invention.

[0040] Figure 10 This is a comparison diagram of conducted EMI of two modulation methods at different frequencies implemented according to the present invention. Detailed Implementation

[0041] The specific embodiments of the present invention will now be described with reference to the accompanying drawings to enable those skilled in the art to better understand the invention. It should be particularly noted that in the following description, detailed descriptions of known functions and designs that might obscure the main content of the invention will be omitted here.

[0042] Example

[0043] In this embodiment, as Figure 1As shown, in the totem-pole bridgeless PFC circuit, Vin is the AC input voltage source; L is the boost inductor; GaN power switches S1 and S2 form the high-frequency bridge arm, with complementary PWM drive signals; power switches S3 and S4 form the power frequency bridge arm, operating at the same frequency as the input signal and used as freewheeling diodes, typically N-channel MOSFETs; C is the output filter capacitor, and R is the equivalent load. The circuit operates in the positive and negative half-cycles depending on the AC input voltage. In the positive half-cycle, S2 is the main switch of the high-frequency bridge arm, S1 acts as the synchronous rectifier, power frequency bridge arm switch S4 is on, and S3 is off. Due to the highly symmetrical topology of the GaN totem-pole bridgeless PFC circuit, the operating modes in the positive and negative half-cycles are almost perfectly symmetrical.

[0044] Both the switching transistor and the diode are considered ideal devices, and their conduction losses and turn-on times are not included in the analysis. Since the switching transistor operates at a frequency of 100kHz, which is much higher than the frequency of the input AC voltage signal, the input voltage can be approximated as a constant value V during the analysis of one switching cycle. in Similarly, the capacitance value of the output capacitor C is chosen to be sufficiently large, so it can be approximated as a constant voltage source V during the analysis. o During the positive half-cycle, switch S4 in the power frequency bridge arm remains on, while switch S3 remains off. When the circuit switches to the negative half-cycle, the switching states of switches S3 and S4 are exactly opposite to those in the positive half-cycle. The GaN totem-pole bridgeless PFC circuit operating in continuous conduction mode can be mainly divided into two operating phases: the energy storage phase and the freewheeling phase. Furthermore, two switches on the same bridge arm are not allowed to conduct simultaneously; an appropriate dead time needs to be added during mode transitions.

[0045] The following is a detailed description of the inductor current ripple peak control method for a totem-pole bridgeless PFC converter according to the present invention, such as... Figure 2 As shown, it includes the following steps:

[0046] S1. Based on the topology of the totem pole bridgeless PFC converter, mark the high-frequency bridge arm switches S1 and S2 and the power frequency bridge arm switches S3 and S4, and mark the boost inductor L.

[0047] S2. Collect the input voltage V of the totem-pole bridgeless PFC converter at time t. in (t), Input current I in (t) and output voltage V o (t);

[0048] S3, during a fixed switching period T c Below, predict the peak inductor current ripple i of the totem-pole bridgeless PFC converter after the switching change at time t+1. pk (t+1);

[0049] In a single switching cycle within the power frequency half-cycle, the totem-pole bridgeless PFC converter is divided into an energy storage section and a freewheeling section based on the switching states of the switching transistors S1 and S2.

[0050] In this embodiment, as Figure 3 As shown, the rapid operation of the circuit switch transistor causes the circuit's operating state to switch quickly. The inductor charges during the energy storage phase and discharges during the freewheeling phase. Therefore, while transmitting the operating current, the inductor also transmits a significant amount of high-frequency noise into the circuit. This high-frequency noise tends to concentrate near multiples of the switching frequency, generating substantial differential-mode noise. Since the inductor charges during the energy storage phase and discharges during the freewheeling phase, it also transmits a significant amount of high-frequency noise into the circuit. This high-frequency noise tends to concentrate near multiples of the switching frequency, generating substantial differential-mode noise. Therefore, appropriate measures should be taken to suppress it. We will now elaborate on the energy storage and freewheeling phases as follows:

[0051] The voltage across the inductor in the energy storage section is the input voltage, and the inductor current is i. L The formula for rising is as follows:

[0052]

[0053] The voltage across the freewheeling section of the inductor is the difference between the input and output voltages, and the inductor current i L The formula for the decrease is as follows:

[0054]

[0055] The stable operating state output voltage V o Numerically greater than the input voltage V in The totem-pole bridgeless PFC converter is structurally perfectly symmetrical, therefore the analysis process for the negative and positive half-cycles is identical, and the waveforms are symmetrical. Based on the inductor current calculation formulas for the energy storage and freewheeling sections, the following can be obtained:

[0056]

[0057] Among them, i t D represents the current flowing through the boost inductor L after the switching change at time t; t+1 The duty cycle after the switch changes at time t+1;

[0058] In this embodiment, the prediction of the switching frequency change of VSFPWM is based on the circuit input inductor current ripple. Therefore, as Figure 4As shown, regardless of whether the circuit is in the positive or negative half-cycle of the power frequency, the ripple of the inductor current in a single switching cycle is a broken line shape, where (a) represents the waveform of one switching cycle in the positive half-cycle; and (b) represents the waveform of one switching cycle in the negative half-cycle.

[0059] A simulation of current ripple prediction for a totem-pole bridgeless PFC circuit was built using the Matlab / Simulink simulation platform. Figure 4 Input the time-domain waveforms of the inductor current and fundamental current. Simulation parameters: input AC voltage signal 220Vac, switching frequency 100kHz, inductance 1.5mH. Figure 5 The thin black solid line represents the time-domain waveform of the input inductor current, while the thick black solid line represents the fundamental inductor current obtained after Fourier transform processing. Figure 5 It can be observed that the actual input inductor current waveform is a high-frequency ripple signal superimposed on the fundamental current.

[0060] Figure 6 The inductor current ripple is compared with the predicted ripple peak value. The predicted peak ripple current completely encompasses the instantaneously changing ripple current, which shows that the current ripple prediction principle is theoretically correct and reliable, laying a theoretical foundation for the realization of variable switching frequency.

[0061] S4. Calculate the duty cycle D after the switch change at time t+1. t+1 ;

[0062] In this embodiment, the output of the inner current loop in the dual closed-loop control circuit of the totem-pole bridgeless PFC circuit is a signal carrying duty cycle information. Since there is no integral element in the duty cycle of the output modulation wave, the digital control system can significantly reduce the running time of the integral element, thereby improving the system control frequency and the converter's dynamic response. The duty cycle calculation process is as follows:

[0063] S4.1 Calculate the inductor current i after the switch changes at time t+1. t+1 :

[0064]

[0065] S4.2, reduce the peak value of the inductor current ripple i pk (t+1) introduces the inductor current i t+1 :

[0066]

[0067] S4.3. Based on the topology of the totem-pole bridgeless PFC converter, specify the input reference value i from the outer voltage loop to the inner current loop. avg ;

[0068] In this embodiment, in the dual closed-loop control circuit of the totem pole bridgeless PFC circuit, the output of the voltage outer loop is the input reference value of the current inner loop, and the output of the current inner loop is a signal carrying duty cycle information, so it can be used directly.

[0069] Take i t+1 =i avg Therefore, we get:

[0070]

[0071] S5. Calculate the peak value of the inductor current ripple i pk (t+1) Switching period T when controlled v (t+1);

[0072] In this embodiment, it is typically designed that the peak ripple of the input inductor current in a conventional totem-pole bridgeless PFC converter will inevitably be less than a certain value during stable operation. This value is the maximum ripple peak value in the circuit, and this value is limited to within the ripple limit. Based on the waveform of the current ripple during circuit operation, it can be concluded that the time when the current ripple reaches its maximum value only accounts for a small portion of the operating cycle. This degree of control freedom can be considered wasted. Assuming that all times are limited to the maximum value, the switching cycle can be changed while meeting the ripple peak control requirements.

[0073] Analyzing the formula for peak current ripple, if we take the initial current of each switching change as the reference, i.e., let i t =0, and we can see the peak current ripple i pk (t+1) is directly proportional to both the switching period and the duty cycle. Since the circuit design uses a switching frequency much higher than the power frequency, the input voltage and duty cycle can be considered fixed values ​​within a single switching cycle. Therefore, the peak current ripple is directly proportional to the switching period.

[0074] Therefore, in a fixed switching period T c Below, the predicted peak current ripple is i pk (t+1), then the peak value of the inductor current ripple will be clamped to the limit value I. required Switching cycle at time:

[0075]

[0076] Among them, I required This is a limit value for the peak value of the inductor current ripple;

[0077] S6, The controller operates according to the switching cycle T. v (t+1) Generates a control signal, which is then sent to the switching transistor through the drive module, thereby completing the control of the inductor current ripple peak of the totem pole bridgeless PFC converter at the next moment.

[0078] In this embodiment, after current ripple prediction, the block diagram for achieving variable switching frequency PWM control is as follows: Figure 7 As shown, after the duty cycle of the controller output is predicted by the current ripple peak value, the new switching period T is calculated. v (t+1), simultaneously generating a sampling signal and a carrier signal based on the new switching cycle. Finally, the carrier signal is compared with the duty cycle signal to obtain the corresponding PWM signal, which is output to the switching device. In the totem-pole bridgeless PFC circuit, the ripple of the input inductor current is unevenly distributed within the power frequency cycle. Therefore, the prediction model based on the peak value of the current ripple can update the switching cycle of the circuit in real time, and simultaneously update the carrier signal and the sampling signal.

[0079] S7. When the next sampling time arrives, return to step S2.

[0080] Figure 8 Current ripple waveform with variable switching frequency.

[0081] according to Figure 3 Simulation results of the current ripple show that the maximum current ripple at a fixed switching frequency is approximately 0.3A; therefore, the upper limit of the ripple is set to 0.3A. A variable switching frequency model was built using the Matlab / Simulink simulation platform, and the current ripple was extracted as follows: Figure 8 As shown, the peak current ripple was clamped to 0.3A at more times, making full use of the peak current ripple as a degree of freedom to update the switching frequency.

[0082] The model predicts the control block diagram of the totem-pole bridgeless PFC circuit for VSFPWM as follows: Figure 9 As shown in the circuit diagram, the PWM loop improved by the model-predicted VSFPWM is connected to the output of the current control loop, without affecting the control of the voltage and current loops. Within a single control cycle, the model-predicted VSFPWM module reads the current loop output signal command, which contains duty cycle information. Then, it obtains the predicted ripple current peak value through a current ripple peak prediction model, calculates the corresponding switching cycle based on the ripple current peak value limit, and updates the sampling signal and carrier signal simultaneously. Finally, it transmits the PWM control signal obtained by comparing the carrier signal with the duty cycle to the switching devices in the circuit to complete the control process of one switching cycle.

[0083] Figure 10Comparison of conducted EMI for two modulation methods at different frequencies: (a) conducted EMI with a fixed initial switching frequency of 50Hz; (b) conducted EMI with a variable initial switching frequency of 50Hz; (c) conducted EMI with a fixed initial switching frequency of 80Hz; (d) conducted EMI with a variable initial switching frequency of 80Hz; (e) conducted EMI with a fixed initial switching frequency of 100Hz; and (f) conducted EMI with a variable initial switching frequency of 100Hz.

[0084] To verify the impact of switching frequency on EMI, a comparison of conducted EMI spectra for two modulation methods—fixed switching frequency and variable switching frequency—at different initial switching frequencies is presented. The initial switching frequencies, from top to bottom, are 100kHz, 80kHz, and 50kHz. The figures clearly show that the EMI spectrum of VSFPWM is more stable in the high-frequency range and has much smaller fluctuations than that of CSFPWM. Moreover, the attenuation of the EMI spectrum generally increases gradually with the higher the initial frequency. VSFPWM utilizes the degree of freedom of the switching frequency to distribute the concentrated spectral energy over a wider range, reducing EMI spikes near integer multiples of the switching frequency and resulting in a more uniform energy distribution.

[0085] In summary, appropriately increasing the switching frequency helps VSFPWM distribute EMI energy more widely and evenly, thereby reducing EMI while also reducing switching losses.

[0086] Although the illustrative specific embodiments of the present invention have been described above to enable those skilled in the art to understand the invention, it should be understood that the invention is not limited to the scope of the specific embodiments. For those skilled in the art, various changes are obvious as long as they are within the spirit and scope of the invention as defined and determined by the appended claims, and all inventions utilizing the concept of the present invention are protected.

Claims

1. A method for controlling the peak inductor current ripple of a totem-pole bridgeless PFC converter, characterized in that, Includes the following steps: (1) Based on the topology of the totem pole bridgeless PFC converter, mark the high-frequency bridge arm switches S1 and S2 and the power frequency bridge arm switches S3 and S4, and mark the boost inductor L. (2) Collect the input voltage V of the totem pole bridgeless PFC converter at time t. in (t), Input current I in (t) and output voltage V o (t); (3) During a fixed switching period T c Below, predict the peak inductor current ripple i of the totem-pole bridgeless PFC converter after the switching change at time t+1. pk (t+1); Among them, i t D represents the current flowing through the boost inductor L after the switching change at time t; t+1 The duty cycle after the switch changes at time t+1; (4) Calculate the duty cycle D after the switch changes at time t+1. t+1 ; (4.1) Calculate the inductor current i after the switch changes at time t+1. t+1 : (4.2) The peak value of the inductor current ripple i pk (t+1) introduces the inductor current i t+1 : (4.3) Based on the topology of the totem-pole bridgeless PFC converter, the input reference value i from the outer voltage loop to the inner current loop is given. avg ; Take i t+1 =i avg Therefore, we get: (5) Calculate the peak value of the inductor current ripple i pk (t+1) Switching period T when controlled v (t+1); Among them, I required This is a limit value for the peak value of the inductor current ripple; (6) The controller determines the switching period T v (t+1) Generates a control signal, which is then sent to the switching transistor through the drive module, thereby completing the control of the inductor current ripple peak of the totem pole bridgeless PFC converter at the next moment. (7) When the next sampling time arrives, return to step (2).

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