Phased array antenna and antenna for a phased array antenna

Abstract

An antenna for a phased array antenna and a phased array antenna. The antenna for a phased array antenna includes a substrate, a radiating element arranged on the substrate, and a radiation pattern adjustment structure operable to reduce a central radiation provided by the radiating element during operation.

Description

Technical Field

[0001] This invention relates to a phased array antenna and an antenna for use with a phased array antenna. Background Technology

[0002] Phased array antennas can be used in various applications, such as radar and wireless telecommunications systems. Typically, a phased array antenna comprises multiple antenna elements and can be electronically scanned (e.g., by applying variable phase or time delays to excite different antenna elements). However, the scanning range of conventional phased array antennas can be limited because, at large scanning angles, angular variations in the element radiation field and the effects of inter-element coupling can lead to significant gain losses. In other words, when the beam of a phased array antenna is steered over a large range, the phased array antenna may experience a severe gain drop (e.g., greater than 3 dB over the scanning range). Summary of the Invention

[0003] In a first aspect, an antenna for or belonging to a phased array antenna is provided. The antenna includes a substrate, a radiating element disposed on the substrate, and a radiation pattern shaping structure operable to reduce the central radiation provided by the radiating element during operation. The central radiation may be central radiation in the H-plane and / or E-plane of the radiating element. The reduction in central radiation corresponds to a central dip in the radiation pattern. The radiation pattern shaping structure is provided partially or entirely by structural components of the antenna. In some embodiments, the antenna can also be used as a non-phased array antenna.

[0004] The radiation pattern adjustment structure is also operable to increase the beamwidth of the radiation provided by the radiating element during operation. The beamwidth of the radiation can be the beamwidth of the radiation in the H-plane and / or E-plane of the radiating element.

[0005] The substrate may include (or only include) one or more substrate layers and a ground plane. The ground plane is disposed on one side of the substrate. The substrate may be a printed circuit board (PCB) substrate, such as a single-layer or multi-layer PCB substrate.

[0006] The antenna also includes one or more feed structures. Examples of feed structures include probe feed structures, microstrip line feed structures, coplanar waveguide feed structures, slot feed structures, etc. In one example, the feed structure includes / is a slot feed structure, which includes a feed slot formed in a ground plane and a microstrip line disposed on the other side of the substrate (the side opposite to the ground plane).

[0007] Radiation elements can be arranged directly or indirectly on the substrate.

[0008] Antennas can be dipole antennas, patch antennas, slot antennas, dielectric resonator antennas, etc.

[0009] Optionally, the antenna is a dielectric resonator antenna, and the radiating element includes (or only includes) a dielectric resonator element. The dielectric resonator element can be made of one or more dielectric materials, and therefore can include a dielectric constant (made of one dielectric material) or an effective dielectric constant (made of multiple dielectric materials). The dielectric resonator element can be shaped like a cylinder, prism, etc. A cylinder can be a right cylinder. A cylinder can be a regular cylinder, an elliptical cylinder, a parabolic cylinder, a hyperbolic cylinder, etc. A prism can be a right prism. A prism can be a triangular prism, a rectangular prism, a cube, a polygonal prism, etc. The dielectric resonator element can be manufactured using additive manufacturing technology.

[0010] Optionally, the radiation pattern adjustment structure includes a radiation pattern adjustment element disposed on a substrate and defining an opening for receiving a portion of a dielectric resonator element. The radiation pattern adjustment element may be disposed directly or indirectly on the substrate. The dielectric resonator element and the radiation pattern adjustment element may be supported on the same surface (e.g., a generally flat surface), which may be provided by the substrate or other structures disposed on the substrate. The opening may be a through-hole. The radiation pattern adjustment element may be coaxially disposed with or centrally disposed within the opening. The cross-sectional shape of the radiation pattern adjustment element and the cross-sectional shape of the opening may be the same (the cross-sectional dimensions may be the same or different). Optionally, the radiation pattern adjustment element and the dielectric resonator element are not in direct contact. Optionally, a gap is defined between the radiation pattern adjustment element and the dielectric resonator element. This gap may be annular. Optionally, the radiation pattern adjustment element is an annular element surrounding the lower portion (closer to the substrate) of the dielectric resonator element. An annular element may refer to an element with a ring shape. The annular element may be circular. Optionally, the height of the radiation pattern adjustment element is less than half the height of the dielectric resonator element. Optionally, the radiation pattern adjustment element is a metal element made of one or more metallic materials. Optionally, the radiation pattern adjustment element is manufactured using additive manufacturing technology.

[0011] Optionally, the radiation pattern adjustment structure also includes a dielectric element disposed on the dielectric resonator element. The dielectric element is operable to promote resonance (one or more modes) in the dielectric resonator element during operation. The dielectric element may be made of one or more dielectric materials, and therefore may include a dielectric constant (made of one dielectric material) or an effective dielectric constant (made of multiple dielectric materials). The dielectric constant or effective dielectric constant of the dielectric element may be less than the dielectric constant or effective dielectric constant of the dielectric resonator element, thereby forming a quasi-magnetic interface between the dielectric element and the dielectric resonator element. The dielectric constant or effective dielectric constant of the dielectric element may be less than half the dielectric constant or effective dielectric constant of the dielectric resonator element. The height of the dielectric element may be less than the height of the dielectric resonator element. The dielectric element and the dielectric resonator element may be coaxial and have the same cross-sectional shape and / or dimensions. The shape of the dielectric element may be a cylinder, prism, etc. The cylinder may be a right circular cylinder, an elliptical cylinder, a parabolic cylinder, a hyperbolic cylinder, etc. The prism can be a right prism. The prism can be a triangular prism, a rectangular prism, a cube, a polygonal prism, etc. The dielectric element can be manufactured using additive manufacturing technology. The dielectric element and the dielectric resonator element can be integrally formed. In one example, the dielectric element and the dielectric resonator element form a cylinder that includes (or only includes) a lower cylindrical portion defined by the dielectric resonator element and an upper cylindrical portion defined by the dielectric element.

[0012] Optionally, the dielectric element is a first dielectric element, and the radiation pattern adjustment structure further includes a second dielectric element disposed on the first dielectric element. The second dielectric element may be made of one or more dielectric materials, and therefore may include a dielectric constant (made of one dielectric material) or an effective dielectric constant (made of multiple dielectric materials). The dielectric constant or effective dielectric constant of the second dielectric element may be greater than the dielectric constant or effective dielectric constant of the first dielectric element. The dielectric constant or effective dielectric constant of the second dielectric element may be at least twice the dielectric constant or effective dielectric constant of the first dielectric element. In one example, the dielectric constant or effective dielectric constant of the second dielectric element is the same as the dielectric constant or effective dielectric constant of the dielectric resonator element. The height of the second dielectric element may be less than the height of the first dielectric element. Two or all of the first dielectric element, the second dielectric element, and the dielectric resonator element may be coaxial. Two or all of the first dielectric element, the second dielectric element, and the dielectric resonator element may have the same cross-sectional shape and / or dimensions. The shape of the second dielectric element may be a cylinder, a prism, etc. The cylinder can be a right circular cylinder, an elliptical cylinder, a parabolic cylinder, a hyperbolic cylinder, etc. The prism can be a right prism. The prism can be a triangular prism, a rectangular prism, a cube, a polygonal prism, etc. The second dielectric element can be manufactured using additive manufacturing technology. The first and second dielectric elements can be integrally formed. The first dielectric element, the second dielectric element, and the dielectric resonator element can be integrally formed. In one example, the first dielectric element, the second dielectric element, and the dielectric resonator element form a cylinder that includes (or only includes) a lower cylindrical portion defined by the dielectric resonator element, an intermediate cylindrical portion defined by the first dielectric element, and an upper cylindrical portion defined by the second dielectric element. The top tip or top surface of the second dielectric element can provide a quasi-magnetic interface, which helps to adjust the field distribution. In one example, the radiation pattern adjustment structure includes only the first dielectric element, the second dielectric element, and the dielectric resonator element.

[0013] In one embodiment where the antenna feed structure includes a slot-feed structure with a feed slot, the antenna may further include a dielectric layer disposed on a substrate and located between the substrate and the dielectric resonator element. A gap is defined between the feed slot and the dielectric resonator element, which can cause mismatch and / or frequency shift. The dielectric layer can reduce the mismatch and / or frequency shift caused by the gap. The dielectric layer is a low-dielectric-constant dielectric layer.

[0014] In one implementation, the antenna may operate only in, or at least in, the X-band. The antenna may also operate in additional or alternative frequency bands.

[0015] The antenna can be used as a transmitter antenna. It can also be used as a receiver antenna or a transceiver antenna.

[0016] In a second aspect, a phased array antenna is provided, having at least one antenna of the first aspect. Each of the at least one antenna of the first aspect can be used as an antenna element of the phased array antenna. The phased array antenna can provide a scanning range of at least ±50°, at least ±60°, at least ±70°, at least ±72°, at least ±75°, or at least ±80°, and has a gain fluctuation of less than 2dB, less than 1.5dB, less than 1.25dB, less than 1.2dB, less than 1.1dB, less than 1.0dB, or less than 0.9dB within the scanning range.

[0017] Thirdly, a phased array antenna is provided, having a substrate, at least two radiating elements disposed on the substrate, and at least two radiation pattern adjustment structures. Each of the at least two radiation pattern adjustment structures is associated with a corresponding one of the at least two radiating elements and is operable to reduce the central radiation provided by the corresponding radiating element during operation. The at least two radiation pattern adjustment structures are provided partially or entirely by structural components of the phased array antenna.

[0018] The at least two radiating elements may include more than two radiating elements arranged in an array. The array may be a one-dimensional or two-dimensional array. The array may be a linear or planar array. The array may have a triangular or rectangular arrangement. The at least two radiating elements may be identical or may be radiating elements of different types, shapes, forms, and / or sizes. The radiating elements may be dielectric resonator elements.

[0019] The phased array antenna can provide a scanning range of at least ±50°, at least ±60°, at least ±70°, at least ±72°, at least ±75° or at least ±80°, and has a gain fluctuation of less than 2dB, less than 1.5dB, less than 1.25dB, less than 1.2dB, less than 1.1dB, less than 1.0dB or less than 0.9dB within the scanning range.

[0020] In one embodiment, each of the at least two radiating elements includes a corresponding dielectric resonator element, and the at least two radiation pattern adjustment structures may be provided by at least a single radiation pattern adjustment element. The single radiation pattern adjustment element is disposed on a substrate and defines at least two openings, each opening receiving a portion of the corresponding dielectric resonator element. The radiation pattern adjustment element may be shaped as a cylinder, prism, etc.

[0021] The at least two radiation pattern adjustment structures may each further include at least one dielectric element (e.g., a plate) disposed on a respective dielectric resonator element. The at least one dielectric element may be a first dielectric element (optionally, and a second dielectric element) of the first aspect.

[0022] In one embodiment, the at least one radiation pattern adjustment structure may be a radiation pattern adjustment structure of the first aspect.

[0023] Phased array antennas can be used as transmitter antennas, and can also be used as receiver antennas or transceiver antennas.

[0024] In a fourth aspect, an electronic device / system having the antenna of the first aspect is provided. This electronic device / system can be a communication device / system.

[0025] In a fifth aspect, an electronic device / system having a phased array antenna as described in the second aspect is provided. This electronic device / system can be a communication device / system.

[0026] In a sixth aspect, an electronic device / system with a phased array antenna as described in the third aspect is provided. This electronic device / system may be a communication device / system.

[0027] In a seventh aspect, a method for constructing a phased array antenna having multiple antenna elements is provided. This method can be implemented at least in part by a computer (e.g., a processor). The method includes: determining the (optimal) isolated element pattern of a single antenna element of the phased array antenna based on the array factor and the array beam envelope (gain curve) of the phased array antenna and considering the mutual coupling effects associated with the antenna elements; determining the design of a single antenna element based on the determined isolated element pattern; and manufacturing the phased array antenna based on the determined design of the single antenna element.

[0028] Optionally, the method further includes determining the design of the phased array antenna based on the single antenna element, and manufacturing the phased array antenna based on the design of the phased array antenna.

[0029] Optionally, the method further includes: defining the array beam envelope (gain curve) G of the phased array antenna. a Based on the array information of the phased array antenna, determine the design of the initial antenna element and the design of the corresponding phased array antenna with multiple such initial antenna elements; determine the array factor beam envelope G, which includes mutual coupling, associated with the phased array antenna. AF ; and determine the active reflection coefficient Γ of each feed port of the designed phased array antenna at different scanning angles. n (θ). Determining the isolated cell pattern also includes: determining the isolated cell pattern G0 based on the defined array beam envelope and the array factor beam envelope containing mutual coupling.

[0030] Optionally, the method further includes: receiving the array beam envelope (gain curve) G of the phased array antenna. aRelated inputs; receiving array information based on the phased array antenna, related to the design of the initial antenna element and the design of the corresponding phased array antenna with multiple such initial antenna elements; determining the array factor beam envelope G associated with the phased array antenna, including mutual coupling. AF ; and determine the active reflection coefficient Γ of each feed port of the designed phased array antenna at different scanning angles. n (θ). Determining the isolated cell pattern also includes determining the isolated cell pattern G0 based on the defined array beam envelope and the array factor beam envelope containing mutual coupling. One or more of these steps can be performed using a computer.

[0031] The array beam envelope can be substantially constant within the target scanning range. The array beam envelope may decrease rapidly outside the target scanning range. The array beam envelope can be represented as... Where θ is the scanning angle, f0(θ) is the normalized field pattern of the antenna element, η0 is the wave impedance in free space, and N is the total number of antenna elements in the phased array antenna.

[0032] The array factor beam envelope containing mutual coupling can be expressed as: Where θ is the scanning angle, Γ n (θ) represents the active reflection coefficient of the nth antenna element when the main beam scans to θ, and N is the total number of antenna elements in the phased array antenna.

[0033] The radiation pattern of an isolated cell can be represented as Where θ is the scanning angle, f0(θ) is the normalized field pattern of the antenna element, and η0 is the wave impedance in free space.

[0034] In one example, the isolated cell pattern G0 is determined based on the defined array beam envelope and the array factor beam envelope containing mutual coupling, according to the following equation: G0(θ) = G a (θ) / G AF (θ).

[0035] Other features and aspects of the invention will become apparent from consideration of the detailed description and accompanying drawings. Where appropriate and applicable, any feature described herein with respect to one aspect or embodiment may be combined with any other feature described herein with respect to any other aspect or embodiment.

[0036] Terms of degree such as “generally,” “about,” “roughly,” or similar (used depending on the context) are used to consider manufacturing tolerances, degradation, trends, tendencies, practical applications, etc. For example, when a numerical value is modified by a degree term such as “about,” this expression can include ±15%, ±10%, ±5%, ±2%, or ±1% of that value.

[0037] Unless otherwise stated, the terms “connection,” “coupling,” “linking,” “installation,” etc., are intended to cover both direct and indirect connections, couplings, links, installations, etc. Attached Figure Description

[0038] Embodiments of the invention will now be described by way of example with reference to the accompanying drawings, in which:

[0039] Figure 1 This is a flowchart illustrating a method for constructing a phased array antenna according to one embodiment of the present invention;

[0040] Figure 2 This is a schematic diagram of a multi-unit linear cylindrical dielectric resonator array;

[0041] Figure 3 This shows the array beam envelope G of the phased array antenna. a (θ), the isolated element pattern G0(θ) of the phased array antenna, and the array factor beam envelope G of the phased array antenna containing mutual coupling. AF A graph of the normalized curve of (θ) (the relationship between normalized gain in dB and scan angle in degrees).

[0042] Figure 4 This shows the scanning range θ for different targets. Aim A graph showing the normalized curves (the relationship between normalized gain in dB and scan angle in degrees) of different isolated unit radiation patterns G0(θ).

[0043] Figure 5 This illustrates different array factor beam envelopes G containing mutual coupling for different numbers of antenna elements in a phased array antenna. AF A graph of the normalized curves (θ) and different isolated unit patterns G0(θ) (the relationship between normalized gain in dB and scan angle in degrees).

[0044] Figure 6A This is a schematic diagram (perspective view) of a dielectric resonator antenna according to one embodiment of the present invention;

[0045] Figure 6B yes Figure 6A Side view of a dielectric resonator antenna;

[0046] Figure 6C yes Figure 6A A top view of a dielectric resonator antenna;

[0047] Figure 7 This is a schematic diagram illustrating an antenna design process (stage I) according to one embodiment of the present invention, showing the design "Antenna I" before modification and the design "Antenna II" after modification;

[0048] Figure 8A yes Figure 7 The simulated complex amplitude electric field distribution of the design "Antenna I" in the xz plane at 10.35 GHz;

[0049] Figure 8B yes Figure 7 The simulated complex amplitude electric field distribution of the "Antenna II" design at 10.35 GHz in the xz plane;

[0050] Figure 9 This shows the different values ​​of the metal ring height H1 (where the dielectric resonator element height H2 is 9 mm and the microstrip line length L1 is 19.6 mm). Figure 7 The design of "Antenna I" and Figure 7 The simulated radiation pattern of the "Antenna II" design in the H plane (the relationship between normalized gain in dB and scanning angle in degrees).

[0051] Figure 10 This shows the different values ​​of the metal ring height H1 (where the dielectric resonator element height H2 is 9 mm and the microstrip line length L1 is 19.6 mm). Figure 7 The design of "Antenna I" and Figure 7 The simulated reflection coefficient of the "Antenna II" design |S 11 | A curve graph;

[0052] Figure 11 This is a schematic diagram illustrating an antenna design process (stage II) according to one embodiment of the present invention, showing... Figure 7 The design of "Antenna II" (before modification) and the dielectric resonator antenna design of one embodiment of the present invention (after modification);

[0053] Figure 12 yes Figure 11 A simplified equivalent model for dielectric resonator antenna design;

[0054] Figure 13 It is shown Figure 11 The simulated radiation pattern (the relationship between normalized gain in dB and scanning angle in degrees) of the dielectric resonator antenna in the H plane when the upper dielectric plate height H4 (where the metal ring height H1 is 5 mm) is different.

[0055] Figure 14A It is based on Figure 11 A photograph (perspective view) of the dielectric resonator antenna designed and manufactured;

[0056] Figure 14B yes Figure 14A A photograph (bottom view) of a dielectric resonator antenna;

[0057] Figure 15 It is shown Figure 14A The measured and simulated curves of reflection coefficient (dB) and peak gain (dBi) of the dielectric resonator antenna;

[0058] Figure 16 It is shown Figure 14A A graph showing the simulated reflection coefficient (dB) of a dielectric resonator antenna with different sized air gaps between the dielectric resonator element and the feed slot.

[0059] Figure 17 yes Figure 14A Measurement and simulation of the radiation pattern of the dielectric resonator antenna at 10.65 GHz;

[0060] Figure 18A This is a schematic diagram (perspective view) of a dielectric resonator phased array antenna according to one embodiment of the present invention;

[0061] Figure 18B yes Figure 18A A top view of a dielectric resonator phased array antenna;

[0062] Figure 19A Based on Figure 18A A photograph (perspective view) of a dielectric resonator phased array antenna designed and manufactured;

[0063] Figure 19B yes Figure 19A A photograph (bottom view) of a dielectric resonator phased array antenna;

[0064] Figure 20 It is shown Figure 19A The measured and simulated reflection coefficient (dB) curves of elements 1, 3 and 5 of the dielectric resonator phased array antenna;

[0065] Figure 21 It is shown Figure 19A A graph showing the measured and simulated isolation between adjacent elements of a dielectric resonator phased array antenna;

[0066] Figure 22A It is shown Figure 19A A graph showing the simulated active voltage standing wave ratio (VSWR) of element 1 of a dielectric resonator phased array antenna.

[0067] Figure 22B It is shown Figure 19A A graph showing the measured active voltage standing wave ratio (VSWR) of element 1 of a dielectric resonator phased array antenna.

[0068] Figure 22C It is shown Figure 19AThe simulated active voltage standing wave ratio (VSWR) curve of element 5 of the dielectric resonator phased array antenna;

[0069] Figure 22D It is shown Figure 19A A graph showing the measured active voltage standing wave ratio (VSWR) of element 5 of the dielectric resonator phased array antenna.

[0070] Figure 22E It is shown Figure 19A A graph showing the simulated active voltage standing wave ratio (VSWR) of element 9 of a dielectric resonator phased array antenna.

[0071] Figure 22F It is shown Figure 19A A graph showing the measured active voltage standing wave ratio (VSWR) of element 9 of a dielectric resonator phased array antenna.

[0072] Figure 23A It is shown Figure 19A A graph showing the simulated scanning performance pattern (realized gain in dB versus scanning angle in degrees) of a dielectric resonator phased array antenna; and

[0073] Figure 23B It is shown Figure 19A The measured scanning performance pattern (realized gain in dB versus scanning angle in degrees) of a dielectric resonator phased array antenna is shown in the graph. Detailed Implementation

[0074] According to RC Hansen, Phased Array Antennas, 2nd Edition. Hoboken, NJ: Wiley, 2009, a true array pattern can be decomposed into the isotropic array factor (IAF), the isolated element pattern (IEP), and the impedance mismatch factor (IMF). Traditionally, the active element pattern (AEP) method is used to evaluate array patterns containing mutual coupling. The active element pattern is the product of the isolated element pattern and the impedance mismatch factor. For large arrays, the active element pattern of the center antenna element can roughly represent the active element patterns of almost all other antenna elements in the array. Therefore, the active element pattern of the center antenna element can be used to predict the array pattern. However, obtaining the active element pattern can be time-consuming because a sufficiently large array must be used for this simulation. Furthermore, this active element patterning method may not be suitable for small to medium-sized array antennas, because the mutual coupling effects between elements can vary considerably in arrays of this size.

[0075] The inventors of this invention, through research, experimentation, and testing, have designed a technique based on a gain compensation method to address the gain descent problem in phased array antennas. In one example of the proposed method, the isolated element pattern (IEP) and the array factor are considered together. The mutual coupling between antennas in the array is included when calculating the array factor. The isolated element pattern is optimized so that the element gain is approximately inversely proportional to the peak gain of the array factor. Therefore, the peak gain of the entire array beam can remain nearly constant over a wide scanning range. The inventors of this invention realized that, in order to mitigate gain ripple and / or reduce the complexity of array optimization, mutual coupling effects need to be considered during the antenna element design stage when designing phased array antennas.

[0076] The inventors of this invention, through research, experimentation, and testing, have designed an isolated element pattern method as an alternative to the active element pattern method. In one example of the isolated element pattern method, as an alternative to optimizing the active element pattern, the isolated element pattern is optimized to expand the scan range without significantly reducing gain. By incorporating the impedance mismatch factor into the isotropic array factor, the optimal isolated element pattern can be determined from the desired array pattern and the isotropic array factor including the impedance mismatch factor. The inventors of this invention recognized that for an ideal wide-angle scanning phased array, the peak array gain is almost constant across the scan range, and therefore the isolated element pattern and the beam envelope of the isotropic array factor including the impedance mismatch factor are complementary. Based on this, the inventors of this invention have designed a gain complementation technique based on the isolated element pattern for calculating the optimal element pattern of a wide-angle scanning phased array with low gain ripple.

[0077] First, we will use an ideal linear array as a reference to illustrate the gain complementation technique based on the isolated element pattern method for phased arrays.

[0078] In a linear phased array with N antenna elements, the conventional isotropic array factor can be given by the following equation.

[0079]

[0080] Where ψ = kd sinθ + β, k is the wavenumber, d is the element spacing, θ is the scan angle, and β is the asymptotic phase. In any main lobe direction, ψ equals 0 and the gain of the isotropic array factor reaches its maximum value N. Therefore, the beam envelope of the isotropic array factor at different scan angles is a constant N. However, in real-world phased arrays, mutual coupling is unavoidable and typically leads to a significant gain drop at large scan angles. In one implementation known as the isolated element pattern method, all mutual coupling effects are considered in the array factor. Under this premise, a gain complementation technique is proposed for the beam envelope of the isolated element pattern and the isotropic array factor including the impedance mismatch factor, for use in wide-angle scan phased arrays with low gain ripple.

[0081] By setting the isolated element pattern to G0(θ), the array beam envelope is defined as G a (θ), and the beam envelope containing the impedance mismatch factor and the isotropic array factor is defined as G. AF (θ), we can obtain: G a (θ)=G0(θ)G AF (θ).

[0082] When G is given a (θ) and G AFWhen (θ), the required G0(θ) can be simply calculated as

[0083] G0(θ)=G a (θ) / G AF (θ) (2)

[0084] Where -90°≤θ≤90°.

[0085] For a uniformly excited and equally spaced N-cell linear array with a scanning angle of θ, G a (θ), G0(θ) and G AF (θ) can be written as

[0086]

[0087]

[0088]

[0089] Where f0(θ) is the normalized field pattern of the element, η0 is the wave impedance in free space, and Γ n (θ) represents the active reflection coefficient of the nth element when the main beam scans to θ.

[0090] Generally speaking, the active reflection coefficient Γ n (θ) is difficult to calculate in advance, which makes it difficult to obtain an accurate G. AF (θ). This is because Γ n (θ) is determined by many factors, such as antenna structure, element spacing, and number of elements. Furthermore, Γ n (θ) varies among the elements in the array. However, Γ can be extracted using an electromagnetic field solver (such as ANSYS HFSS software). n (θ).

[0091] For wide-angle scanning phased arrays, G a (θ) is almost constant over the scanning range, therefore G0(θ) and G AF (θ) are complementary. Based on this relationship, an iterative process was developed to design wide-angle scanning phased arrays.

[0092] Figure 1 A flowchart of a method for constructing (including designing) a phased array antenna is shown in one embodiment. The method in this embodiment mainly includes six steps.

[0093] Step 1: Define the required normalization G a (θ). For a phased array antenna, the ideal beam envelope should remain constant within its target scanning range and decrease rapidly outside that range. In one example, G a(θ) can be represented as follows:

[0094]

[0095] Where 0° < θ Aim <90°. In this example, the gain of the main beam is within ±θ. Aim The value is 1 and it exceeds ±θ Aim It begins to decline at a certain time, and its rate of decline is determined by the coefficient n.

[0096] Step 2: Determine the array information (including the element spacing and number of elements for the array antennas) and design the initial antenna elements for the first iteration. Construct the corresponding array and perform simulations, for example, using ANSYS HFSS software.

[0097] Step 3: By extracting the active reflection coefficients of all ports in the array at different scanning angles, the provisional G is obtained according to equation (5). AF (θ).

[0098] Step 4: Calculate the required isolated element pattern using equation (2) and design the antenna based on (e.g., satisfying) the initial target radiation pattern (isolated element pattern). It can be found that the isolated element pattern is generally complementary to the array factor envelope within the relevant scan range.

[0099] Step 5: Use the newly designed cells from Step 4 to build a new array model and perform simulations (e.g., using ANSYS HFSS software) to verify the array's scanning performance.

[0100] Step 6: If the array requirements are met, manufacture the phased array antenna and perform measurements.

[0101] If the array requirements are not met in step 6, the method can return to step 3.

[0102] To explain G more clearly a (θ), G AF The relationship between G(θ) and G0(θ) is illustrated by constructing an example N-element linear antenna array based on a slot-fed cylindrical dielectric resonator antenna, as shown below. Figure 2 As shown. It should be understood that the proposed technique can be applied to any type of antenna element with the desired radiation pattern.

[0103] according to Figure 1 The first step is to determine G. a (θ). In this example, equation (6) is used for G. a (θ), where θ Aim =60°, n=20. Then, set the element spacing to half a wavelength and the number of elements N to 18. G AF(θ) can be obtained using ANSYS HFSS software. Finally, the required element pattern G0(θ) can be calculated according to equation (2).

[0104] Figure 3 G is shown a (θ), G AF The normalized curves of G(θ) and G0(θ). For example... Figure 3 As shown, G AF (θ) and G0(θ) in ±θ Aim Internal complementarity.

[0105] G a The target scanning range in (θ) ±θ Aim It is the main factor affecting the required isolated cell pattern.

[0106] Figure 4 Different required isolated cell patterns are shown for different target scanning ranges (±50 degrees, ±60 degrees, and ±70 degrees). For example... Figure 4 As shown, a wider beamwidth and a deeper central depression in the element pattern are required for a larger scanning range. Therefore, for a wide-angle scanning array with stable peak gain, a wide beamwidth element pattern with a central depression (reduced central radiation) can be designed first.

[0107] The number of elements, N, also affects the shape of the required element pattern. Figure 5 Different G values ​​are shown for different numbers of units N. AF (θ) and G0(θ). Reference Figure 5 As N increases, G AF (θ) descends faster at larger angles. Figure 5 It can be seen that the depth of the central depression also needs to be adjusted as N changes.

[0108] From the above, it can be determined that a wide-angle scanning phased array with low gain fluctuations requires a wide beam pattern with a central depression.

[0109] Although the method described in the above example refers only to a phased array antenna of a one-dimensional (1D) linear phased array, it should be noted that the method is not limited thereto and can also be applied to phased array antennas of two-dimensional (2D) planar phased arrays.

[0110] To verify the above design concept, or more generally, to provide an improved or alternative antenna, the present invention provides an antenna for a phased array antenna.

[0111] The antenna generally includes a substrate, radiating elements disposed directly or indirectly on the substrate, and a radiation pattern adjustment structure operable to reduce the central radiation provided by the radiating elements during operation. The central radiation can be central radiation in the H-plane and / or E-plane of the radiating element. Reduction of central radiation corresponds to a central dip in the radiation pattern. The radiation pattern adjustment structure is provided at least partially by structural components. The radiation pattern adjustment structure can also be operable to increase the beamwidth of the radiation provided by the radiating elements during operation. The beamwidth can be the beamwidth of the radiation in the H-plane and / or E-plane of the radiating element.

[0112] Antennas can be dipole antennas, patch antennas, slot antennas, dielectric resonator antennas, etc. Antennas also include feeding structures, which can be probe-fed structures, microstrip line-fed structures, coplanar waveguide-fed structures, slot-fed structures, etc.

[0113] The antenna can be a dielectric resonator antenna, and the radiating element can include a dielectric resonator element. A dielectric resonator element can be made of one or more dielectric materials, and therefore can include a dielectric constant or an effective dielectric constant. The dielectric resonator element can be shaped into a cylinder, prism, etc., and can be manufactured using additive manufacturing techniques.

[0114] The substrate may include one or more substrate layers and a ground plane disposed on one side of the substrate. The substrate may be provided by a PCB substrate, such as a single-layer or multi-layer PCB substrate.

[0115] A radiation pattern adjustment structure may include a radiation pattern adjustment element disposed directly or indirectly on a substrate and defining an opening (e.g., a through-hole) that surrounds a portion of a dielectric resonator element. The dielectric resonator element and the radiation pattern adjustment element may be supported on the same surface (e.g., a generally flat surface), which may be provided by the substrate or other structures disposed on the substrate. The radiation pattern adjustment element may be arranged coaxially with or centrally within the opening. The cross-sectional shape of the radiation pattern adjustment element and the cross-sectional shape of the opening may be the same (though the cross-sectional dimensions may differ). The radiation pattern adjustment element and the dielectric resonator element may not be in direct contact. A gap, which may be annular, may be defined between the radiation pattern adjustment element and the dielectric resonator element. The radiation pattern adjustment element may be an annular element (which may be circular) surrounding the lower portion of the dielectric resonator element (the portion closer to the substrate). The height of the radiation pattern adjustment element may be less than the height of the dielectric resonator element (e.g., less than half of it). The radiation pattern adjustment element may be a metallic element. The radiation pattern adjustment element may be manufactured using additive manufacturing techniques.

[0116] The radiation pattern adjustment structure may also include a dielectric element (e.g., a dielectric plate) arranged on the dielectric resonator element to, for example, facilitate resonance (one or more modes) in the dielectric resonator element during operation. The dielectric constant or effective dielectric constant of the dielectric element may be less than the dielectric constant or effective dielectric constant of the dielectric resonator element (e.g., less than half of it), thereby forming a quasi-magnetic interface between the dielectric element and the dielectric resonator element. The height of the dielectric element may be less than the height of the dielectric resonator element (e.g., less than half of it). The dielectric element and the dielectric resonator element may be coaxial and have the same cross-sectional shape and / or dimensions. The dielectric element may be shaped as a cylinder, prism, etc. The dielectric element may be manufactured using additive manufacturing techniques. The dielectric element and the dielectric resonator element may be integrally formed. The dielectric element and the dielectric resonator element may form a cylinder.

[0117] The radiation pattern adjustment structure may also include another dielectric element (e.g., a dielectric plate) disposed on the dielectric element. The dielectric constant or effective dielectric constant of this other dielectric element may be greater than that of the dielectric element (e.g., greater than twice its effective dielectric constant). The dielectric constant or effective dielectric constant of this other dielectric element may be the same as that of the dielectric resonator element. The height of this other dielectric element may be less than that of the dielectric element. The dielectric element, the other dielectric element, and the dielectric resonator element may be coaxial. The dielectric element, the other dielectric element, and the dielectric resonator element may have the same cross-sectional shape and / or dimensions. The other dielectric element may be shaped as a cylinder, prism, etc. The other dielectric element may be manufactured using additive manufacturing techniques. The dielectric element and the other dielectric element may be integrally formed. In one example, the dielectric element, the other dielectric element, and the dielectric resonator element form a cylinder. The top tip or top surface of the other dielectric element may provide a quasi-magnetic interface, which helps to adjust the field distribution.

[0118] The antenna may include a slot-fed structure having a feed slot and a low-dielectric-constant dielectric layer disposed on a substrate and between the substrate and a dielectric resonator element. The feed slot and the dielectric resonator element define a gap. The dielectric layer can reduce mismatch and / or frequency shift caused by the gap.

[0119] The antenna can operate only or at least in the X-band. It can be used as a transmitter antenna and may also be used as a receiver antenna or a transceiver antenna.

[0120] Figures 6A to 6CAn example of an antenna 600 for a phased array antenna according to one embodiment is shown. Antenna 600 includes a substrate 602, a dielectric resonator element 604 as a radiating element disposed on the substrate 602, and a radiation pattern adjustment structure. The radiation pattern adjustment structure includes a metal ring 606 disposed on the substrate 602 and two dielectric plates 608, 610 disposed on the dielectric resonator element 604. The radiation pattern adjustment structure is operable to reduce the central radiation provided by the dielectric resonator element 604 during operation.

[0121] refer to Figures 6A to 6C In this embodiment, the dielectric resonator element 604 is cylindrical, having a radius R1, a height H2, and a dielectric constant ε. r1 The dielectric resonator element 604 is mounted on a substrate 602, which is provided by a printed circuit board (PCB) having a square cross-section. The PCB has a side length L. g Thickness H0, substrate dielectric constant ε rs A ground plane is formed on the top surface of the PCB. Below the dielectric resonator element 604 is a rectangular slot 612 with a length L. s and width W s The ring is etched into the ground plane 602G to excite the dielectric resonator element 604. The annular metal ring 606 has an inner radius R2, thickness t, and height H1. The annular ring 606 is placed on the PCB to surround the lower part of the dielectric resonator element 604. Two dielectric plates 608 and 610 (one on top of the other) have heights H3 and H4 and a dielectric constant ε, respectively. r2 and ε r1 Two dielectric substrates 608 and 610 are mounted on a dielectric resonator element 604. The dielectric substrates 608 and 610 and the dielectric resonator element 604 are all cylindrical, and together they define a cylinder. A 50Ω microstrip line 614 with length L1 and width W1 is printed on the bottom of the PCB. The microstrip line 614 and slot 612 form part of the slot-fed structure of antenna 600. Antenna 600 is designed for operation in the X-band. ANSYS HFSS software is used to simulate and optimize the performance of antenna 600. A prototype antenna 1400 was fabricated and measured. The dielectric constants used for antennas 600 and 1400 are as follows: ε r1 =10±0.35, ε r2 =3±0.1 and ε rs = 3.38 ± 0.05. The design parameters for antennas 600 and 1400 are as follows: L g =29mm, L s =6.1mm, L1=20.2mm, W s=0.4mm, W1=1.82mm, t=1.5mm, R1=3.45mm, R2=5mm, H0=0.813mm, H1=4mm, H2=8.6mm, H3=2mm, H4=1mm. It should be noted that in some embodiments, the profile of the dielectric resonator element 604 can be reduced by using a dielectric resonator element 604 with a higher dielectric constant and / or by increasing the radius-to-height ratio of the dielectric resonator element 604.

[0122] Figure 7 Phase I of the antenna 600 design process is shown. This phase can be referred to as the wide-beam design phase. Figure 7 In the diagram, antenna "Antenna I" is a basic slot-fed dielectric resonator antenna, which has the capability to operate in HEM. 11δ The cylindrical dielectric resonator element 604 operates in mode. A large height-to-radius ratio is chosen to obtain a small footprint. In this example, by adding a metal ring 606 around the cylindrical dielectric resonator element 604, "Antenna I" can be transformed into "Antenna II," which helps to introduce a central depression (reduced central radiation) in the radiation pattern and expand the beamwidth of antenna 600.

[0123] Figure 8A and Figure 8B The simulated complex magnitudes of the total electric field distribution in the xz plane for "Antenna I" and "Antenna II" are shown respectively. Figure 8A and Figure 8B It can be seen that the near-field distributions in "Antenna I" and "Antenna II" are very different. For example... Figure 8A As shown, energy is uniformly radiated from the dielectric resonator element 604 of "Antenna I", resulting in a typical side-radiation pattern. Figure 8B As shown, the energy at the bottom of the dielectric resonator element 604 is confined by the metal ring 606 and diffracted at the edge of the metal ring 606. This causes more energy to be transferred to the side directions off the line of sight, resulting in a wide beam radiation pattern. The height of the metal ring 606 of “Antenna II” can affect the propagation path of electromagnetic (EM) waves and change the far-field radiation pattern.

[0124] Figure 9 A comparison of simulated H-plane radiation patterns for "Antenna I" and "Antenna II" with different H1 values ​​(H2 = 9 mm, L1 = 19.6 mm, other parameter values ​​are the same as listed above) is shown. The simulation results in the E-plane are similar to those in the H-plane (for simplicity, the simulation results in the E-plane are not shown here). Reference Figure 9The 3dB beamwidth is significantly increased by loading a metal ring 606 and is mainly controlled by the height H1 of the ring 606. Furthermore, the central depression of the radiation pattern can also be formed with a suitable H1, which is ideal.

[0125] Figure 10 The simulated reflection coefficients of "Antenna I" and "Antenna II" with different H1 values ​​(H2 is 9 mm, L1 is 19.6 mm, and other parameter values ​​are the same as listed above) are shown. Reference Figure 10 By loading a metal ring 606, the 10dB impedance bandwidth narrows, and it decreases as H1 increases. This is because the Q factor of the resonance is increased by the metal ring 606, resulting in a narrower bandwidth.

[0126] Figure 11 Phase II of the antenna 600 design process is shown. This phase can be referred to as the beamforming design phase. In this phase, the shape of the central recess in the element pattern is further adjusted, and additional parasitic structures are introduced. (See diagram for details.) Figure 11 As shown, in this stage, two dielectric plates 608 and 610 are placed on top of the dielectric resonator element 604, forming a dielectric cylinder. The upper dielectric plate 610 has the same dielectric constant ε as the dielectric resonator element 604. r1 The lower dielectric substrate has a much lower dielectric constant ε. r2 .

[0127] Figure 12 It shows Figure 11 A simplified equivalent model of the antenna design (modified, with a dielectric substrate) is provided. Due to the difference in dielectric constant, two quasi-magnetic surfaces, A and B, are formed (where upward waves are bounced without phase change) for waves propagating from the feed slot to the upper space. The lower quasi-magnetic surface B primarily contributes to the formation of the HEM in the dielectric resonator element 604. 11δ The resonance, and the upper quasi-magnetic surface A helps to adjust the field distribution in the dielectric plate 610, and thus the far field can be customized to some extent.

[0128] Figure 13 It shows different H4 (H1 is 5 mm, other parameter values ​​are the same as listed above) Figure 11 The simulated radiation pattern of the modified antenna design (with dielectric substrate) in the H-plane. (Reference) Figure 13 The shape of the central depression in the radiation pattern is effectively controlled by the height of the dielectric substrate 610, while the beamwidth remains almost unchanged. This feature facilitates the design of phased arrays with stable scanning gain.

[0129] Figure 14A and Figure 14BA prototype antenna 1400 is shown, which is based on the design of antenna 600 and manufactured using the parameter values ​​listed above. In this example, the cylindrical dielectric block of antenna 1400 (including the dielectric resonator element and two dielectric plates) is integrally formed using additive manufacturing technology. The prototype antenna 1400 was tested to verify its performance. Specifically, the reflection coefficient was measured using an Agilent E5230A vector network analyzer, and the radiation pattern and achieved gain were measured using a Satimo StarLab system.

[0130] Figure 15 The measured and simulated reflection coefficients of antenna 1400, as well as the peak realized gain, are shown. (Reference) Figure 15 The measured and simulated 10dB impedance bandwidths were 14.9% (9.86 GHz to 11.45 GHz) and 13.5% (9.66 GHz to 11.06 GHz), respectively. The measured peak realized gain varied between 3.5 dBi and 4.5 dBi, while the simulated peak realized gain varied between 3.2 dBi and 4.6 dBi, both within the overlapping bandwidth. There was a 2.8% frequency offset between the measured and simulated results. This was primarily due to manufacturing and experimental tolerances / errors. The inventors of this invention conceived that this could be caused by air gaps between the dielectric resonator elements and the feed slot during manual assembly.

[0131] Figure 16 The simulated reflection coefficients of antenna 1400 with different air gap sizes at 10.65 GHz are shown. (Reference) Figure 16 Tiny air gaps can cause significant frequency shifts, which can explain... Figure 15 The differences observed in the study.

[0132] Figure 17 The measured and simulated radiation pattern of antenna 1400 at 10.65 GHz is shown. Figure 17 As shown, a reasonable agreement can be observed between the measured and simulated results, and wide-beam radiation patterns with a central dip are obtained in both the E and H planes. The measured beamwidths of antenna 1400 in the E and H planes are 172° and 149°, respectively. In both principal planes, the measured co-polarized field is at least 18 dB stronger than the cross-polarized field. The simulated cross-polarized field in the E plane is much weaker than the measured results and cannot even be observed. Figure 17 As shown in the image.

[0133] Table I below summarizes some of the performance and features of the Antenna 1400.

[0134] Table I

[0135]

[0136]

[0137] *The footprint of the main radiator (excluding the ground plane), λ0 is the free-space wavelength at the center frequency.

[0138] The above-described designs of antennas 600 and 1400 can be used to construct wide-angle phased array antennas. In some embodiments, a phased array antenna is provided, which includes one or more antennas 600 and 1400 as antenna elements. The phased array antenna can provide a scan range of at least ±50°, at least ±60°, at least ±70°, at least ±72°, at least ±75°, or at least ±80°, and has a gain ripple of less than 2dB, less than 1.5dB, less than 1.25dB, less than 1.2dB, less than 1.1dB, less than 1.0dB, or less than 0.9dB within this scan range. It should be noted that low gain ripple over a wider scan range will inevitably result in at least a slight reduction in the peak gain of the array antenna.

[0139] As an example, antenna 600 is designed to construct a 9-element H-plane linear phased array antenna 1800. Figure 18A and Figure 18B A phased array antenna 1800 with nine antenna elements 1800-1 to 1800-9 is shown. Since antenna elements 1800-1 to 1800-9 in antenna 1800 are substantially identical and substantially the same as those in antenna 600, the same reference numerals plus "1200" are used to denote identical / similar features (e.g., substrate 602 corresponds to substrate 1802, dielectric resonator element 604 corresponds to dielectric resonator element 1804, etc.). Furthermore, for simplicity, reference numerals are not repeated for different identical antenna elements. The following discussion focuses on aspects that differ from or are not described relative to antenna 600.

[0140] In this embodiment, the element spacing of antenna 1800 is p (p = 0.45λ0, where λ0 is the free-space wavelength at 10.35 GHz). Each antenna element 1800-1 to 1800-9 includes two dielectric plates placed on top of the dielectric resonator elements, as described above with respect to antenna 600. However, unlike antenna 600, in antenna 1800, the loaded metal ring 606 is replaced by a generally rectangular metal block 1806. The metal block 1806 has multiple openings, each receiving a corresponding dielectric resonator element 1804 (its bottom / lower portion). Using a single metal block 1806 has a small impact on radiation performance compared to using multiple metal rings because the electrical boundary at the bottom remains almost unchanged. Using a single metal block 1806 can significantly reduce assembly complexity compared to using multiple metal rings. If metal rings (which can be used in some other embodiments) are used, the dielectric resonator elements of each antenna element in the array must be aligned one after another, which can be challenging and / or time-consuming. Compared to the design of antenna 600, Figure 18A Another difference in the design relates to the ground plane. To mitigate the aforementioned air gap effect, a low-dielectric-constant dielectric layer 1816 with a thickness of H5 and a dielectric constant of 3 is attached to the ground plane 1802G of the substrate 1802 before integrating the dielectric cylinder. Since the difference in dielectric constant between the air and the additional dielectric layer 1816 is sufficiently small, the effective dielectric constant load on the feed slot 1812 will not change significantly even if an air gap is introduced during the assembly or manufacturing of the antenna 1800. Therefore, the mismatch and frequency offset problems caused by the air gap are mitigated.

[0141] The phased array antenna 1800 was simulated and optimized in the X-band using ANSYS HFSS software. A prototype antenna 1900 was fabricated and measured. The dielectric constants for antennas 1800 and 1900 are as follows: ε r1 =10±0.35, ε r2 =3±0.1 and ε rs = 3.38 ± 0.05. The design parameters for antennas 1800 and 1900 are as follows: L g =125mm, L s =9.6mm, L1=21mm, L2=117mm, W1=1.82mm, W2=13mm, W s =0.4mm, W G =29mm, R1=3.45mm, R2=5mm, H0=0.813mm, H1=3.1mm, H2=9mm, H3=2mm, H4=6.4mm, H5=0.6mm.

[0142] Figure 19A and Figure 19B A prototype phased array antenna 1900 is shown, which is based on the design of antenna 1800 and manufactured using the values ​​of the parameters listed above (numbers 1-9 are shown for the dielectric resonator antenna elements). In this example, the cylindrical dielectric block of antenna 1800 (including the dielectric resonator elements and two dielectric plates) is integrally formed using additive manufacturing technology. The metal block is also manufactured using additive manufacturing technology. The prototype phased array antenna 1900 was tested to verify its performance. Specifically, the reflection coefficient was measured using an Agilent E5230A vector network analyzer, and the radiation pattern and achieved gain were measured using a Satimo StarLab system.

[0143] Figure 20 The measured and simulated reflection coefficients of elements 1, 3, and 5 of the phased array antenna 1900 are shown. (Reference) Figure 20 Reasonable agreement can be observed between the measured and simulated results, and the frequency shift is much smaller than that of a single antenna element due to the loading of a low-dielectric-constant dielectric layer. The measured reflection coefficient has an overlap of 10 dB impedance bandwidth of 5.2% (10.04 GHz to 10.58 GHz).

[0144] Figure 21 The measured and simulated isolation between adjacent cells is shown. (Reference) Figure 21 , measurement and simulation of |S 12 |and|S 45 | Less than -19dB within the impedance bandwidth.

[0145] To better understand impedance matching during operation (scanning), the active reflection coefficient at different scanning angles was also investigated. The active reflection coefficient at the m-th port can be calculated using the following equation.

[0146]

[0147] Where S mn These are passive S-parameters in complex form, where k is the wavenumber, d is the element spacing, and θ is the scan angle.

[0148] Using the measured passive S-parameter matrix, the active reflection coefficient of each cell measured at different scanning angles is obtained by equation (7).

[0149] Figures 22A to 22F The active voltage standing wave ratio (VSWR) of units 1, 5, and 9 at different scanning angles is shown. Specifically, Figure 22A and Figure 2 2B shows the measurement and simulation results for Unit 1, respectively. Figure 22C and Figure 22D The measurement and simulation results for Unit 5 are shown respectively. Figure 22E and Figure 22F The measurement and simulation results for Unit 9 are shown respectively.

[0150] like Figures 22A to 22F As shown, due to strong mutual coupling, the active VSWR deteriorates at large angles. This is reasonable, as this design is not optimized for wide-angle impedance matching. Overall, the active VSWR of almost all dielectric resonator antenna elements in the array (except element 5) is less than 3 over the relevant bandwidth. The discrepancy between the measured and simulated results is primarily due to manufacturing and experimental tolerances / errors.

[0151] In one example, the array pattern of the phased array antenna 1900 is obtained using the unit-excitation active element pattern (UEAEP) method. In the UEAEP method, array scanning performance can be synthesized from the active element pattern (without approximation). The active element pattern, including mutual coupling effects, is measured for each element in the array of the phased array antenna 1900. Throughout the measurement process, the phased array antenna 1900 is fixed on the turntable to maintain the relative position information between elements. When measuring the active element pattern of one element, a cable carrying the excitation signal is connected to that element, while other elements are connected to or terminated with a 50Ω matched load. By measuring the antenna elements one by one, nine different active element patterns are obtained. Assuming uniform amplitude and different asymptotic phases, the measured active element patterns are post-processed to calculate the array pattern at different scanning angles.

[0152] Figure 23A and Figure 23B The measured and simulated scanning performance of the phased array antenna 1900 at 10.35 GHz is shown. The function of cos(θ) is also plotted on... Figure 23A and Figure 23B For ease of comparison, as shown in the figure, the gain of the phased array antenna 190° along the scanning angle does not decrease as a function of cos(θ). This is because the array gain is intentionally reduced on the wide side and improved at large angles. This flexible control of the array gain is primarily due to the adjustment of the beam radiation pattern of the dielectric resonator antenna elements. (Reference) Figure 23A The simulated main beam can scan from -72° to +72° with peak gain varying only between 13.3 dBi and 14.1 dBi, and the peak sidelobe level (SLL) is less than -6.1 dB over a wide scan range. (Reference) Figure 23BThe measured main beam can scan from -72° to +72° with a gain varying between 12.5 dBi and 13.4 dBi, and the maximum sidelobe level within the relevant scan range is less than -6.9 dB. The discrepancy between the measured and simulated results is primarily due to manufacturing and experimental tolerances / errors. Sidelobe levels can be further suppressed by optimizing the input amplitude distribution.

[0153] Table II below summarizes some of the performance and features of the Antenna 1900.

[0154] Table II

[0155]

[0156] The above-described embodiments of the present invention provide a novel gain-complementary technique for phased arrays (phased array antennas) with stable scanning gain. In these embodiments, optimized elements in the phased array require a wide beam radiation pattern with a central dip for wide-angle scanning. This technique is demonstrated using different example antenna designs.

[0157] More generally, this invention provides a technique for designing wide-angle scanning phased arrays or phased array antennas with low gain ripple. This technique can obtain optimal isolated element patterns for phased arrays of any size, taking into account mutual coupling effects. This can be achieved by calculating mutual coupling in the array factor. The optimal isolated element pattern can be derived from the desired array beam envelope and the beam envelope of the new array factor. Generally, the antenna elements of this invention have wide beam patterns and beam adjustment capabilities suitable for specific applications. Some embodiments of this invention provide an adjustable beam cylindrical dielectric resonator antenna loaded with an outer metal ring and two top dielectric plates, and having a wide beamwidth and a central recess (adjustable by design) in both the E-plane and H-plane. Some embodiments of this invention provide a multi-element H-plane linear phased array whose H-plane main beam can be scanned over a wide range with very low gain ripple. Some embodiments of this invention allow for flexible and rapid steering of stable high-gain beams over a large coverage area for target search or tracking. Some embodiments of this invention are particularly suitable for radar and various wireless communication applications.

[0158] Some embodiments of the present invention may include one or more of the following advantages. In some embodiments of the present invention, gain complementarity techniques can obtain optimal isolated element patterns for phased arrays of any size, taking into account mutual coupling effects. It can effectively reduce gain ripple and simplify the array optimization process. In some embodiments of the present invention, antennas (e.g., adjustable beam antennas) can flexibly control the center dip in the control pattern while keeping the beamwidth almost constant. This characteristic can compensate for the gain drop of the main beam at large scanning angles. In some embodiments of the present invention, phased array antennas can steer high-gain beams with very low gain ripple over a wide range. Some embodiments of the present invention may include one or more other advantages not specifically described.

[0159] Those skilled in the art will understand that various changes and / or modifications can be made to the invention as shown in certain embodiments to provide other embodiments of the invention. Therefore, the embodiments described herein should be considered illustrative rather than restrictive in all respects. Example optional features of some aspects of the invention are set forth in the Summary of the Invention section. Some embodiments of the invention may include one or more of these optional features (some of which are not specifically shown in the drawings). Some embodiments of the invention may omit (i.e., lack) one or more of these optional features (some of which are not specifically shown in the drawings). For example, the invention can be applied to array antennas with different array arrangement structures (e.g., linear, planar, triangular, etc.). The technology of the invention can be applied to different antenna types, including but not limited to dielectric resonator antennas, patch antennas, slot antennas, dipole antennas, etc. In some embodiments, the antenna construction may differ from that shown. For example, dielectric resonator elements, dielectric plates, metal rings / blocks may have different shapes, sizes, forms, etc. In some embodiments, the dielectric constant or effective dielectric constant of dielectric resonator elements, dielectric plates, etc., may differ from the values ​​shown. In some embodiments, the antenna of the invention can be arranged to operate in other frequency ranges / bands, not limited to the X-band. The degree of reduction in central radiation provided by the radiation pattern adjustment structure (including the range and value of the radiation reduction) can vary for different implementations, i.e., it is not limited to those described above.

Claims

1. An antenna for a phased array antenna, comprising: a substrate; a dielectric resonator element arranged on the substrate; a radiation pattern adjustment structure operable to reduce, during operation, a central radiation provided by the dielectric resonator element; and wherein the radiation pattern adjustment structure comprises a ring-shaped element disposed on the substrate, the ring-shaped element defining an opening that receives a first portion of the dielectric resonator element such that the first portion is surrounded by the ring-shaped element; the dielectric resonator element having a second portion that extends from the opening in a direction away from the substrate. the central radiation is a central radiation in an H-plane and / or an E-plane of the dielectric resonator element.

2. The antenna of claim 1, wherein, the radiation pattern adjustment structure is further operable to increase, during operation, a beamwidth of radiation provided by the dielectric resonator element.

3. The antenna of claim 2, wherein, the beamwidth is a beamwidth of radiation in the H-plane and / or the E-plane of the dielectric resonator element.

4. The antenna of claim 3, wherein, the ring-shaped element and the dielectric resonator element are not in direct contact.

5. The antenna of claim 1, wherein, the ring-shaped element is a metallic element.

6. The antenna of claim 1, wherein, the ring-shaped element has a shape of a circular ring.

7. The antenna of claim 1, wherein, a height of the ring-shaped element is less than half a height of the dielectric resonator element.

8. The antenna of claim 1, wherein, the radiation pattern adjustment structure further comprises a dielectric element arranged on the dielectric resonator element, and the dielectric element is operable to facilitate resonance in the dielectric resonator element during operation.

9. The antenna of claim 1, wherein, a dielectric constant or an effective dielectric constant of the dielectric element is less than a dielectric constant or an effective dielectric constant of the dielectric resonator element, thereby forming a quasi-magnetic interface between the dielectric element and the dielectric resonator element.

10. The antenna of claim 9, wherein, the dielectric constant or the effective dielectric constant of the dielectric element is less than half the dielectric constant or the effective dielectric constant of the dielectric resonator element.

11. The antenna of claim 10, wherein, a height of the dielectric element is less than a height of the dielectric resonator element.

12. The antenna of claim 9, wherein, the dielectric element and the dielectric resonator element are coaxial and have a same cross-sectional shape and / or size.

13. The antenna of claim 9, wherein, the dielectric element and the dielectric resonator element are integrally formed.

14. The antenna of claim 9, wherein, 15. The antenna of claim 13, the dielectric element and the dielectric resonator element form a cylinder; and wherein wherein the dielectric resonator element defines a lower cylindrical portion and the dielectric element defines an upper cylindrical portion.

16. The antenna of claim 9, the dielectric element is a first dielectric element; and wherein the radiation pattern adjustment structure further comprises a second dielectric element arranged on the first dielectric element. wherein, 17. The antenna of claim 16, a dielectric constant or an effective dielectric constant of the second dielectric element is greater than a dielectric constant or an effective dielectric constant of the first dielectric element. a dielectric constant or an effective dielectric constant of the second dielectric element is the same as a dielectric constant or an effective dielectric constant of the dielectric resonator element. wherein a height of the second dielectric element is less than a height of the first dielectric element.

18. The antenna of claim 17, wherein, the first dielectric element, the second dielectric element, and the dielectric resonator element are integrally formed.

19. The antenna of claim 16, wherein, 21. The antenna of claim 16, 20. The antenna of claim 16, wherein, ​ ​ wherein The first dielectric element, the second dielectric element, and the dielectric resonator element form a cylinder; and wherein the dielectric resonator element defines a lower cylindrical portion, the first dielectric element defines an intermediate cylindrical portion, and the second dielectric element defines an upper cylindrical portion.

22. The antenna according to claim 1, wherein, The antenna further comprises: a slot feed structure having a feed slot, the feed slot and the dielectric resonator element defining a gap.

23. A phased array antenna comprising at least one antenna according to claim 1.

24. The phased array antenna of claim 23, wherein, The phased array antenna is operable to provide a scan range of at least ±50° and to have a gain fluctuation of less than 2dB over the scan range.

25. The phased array antenna of claim 23, wherein, The phased array antenna is operable to provide a scan range of at least ±70° and to have a gain fluctuation of less than 2dB over the scan range.

26. The phased array antenna of claim 23, wherein, The phased array antenna is operable to provide a scan range of at least ±70° and to have a gain fluctuation of less than 1dB over the scan range.

27. A phased array antenna comprising: a substrate; at least two dielectric resonator elements arranged on the substrate; at least two radiation pattern adjustment structures, each of the radiation pattern adjustment structures being associated with a respective one of the at least two dielectric resonator elements and being operable to reduce a central radiation provided by the respective dielectric resonator element during operation; and wherein each of the radiation pattern adjustment structures comprises a ring element disposed on the substrate, the ring element defining an opening that receives a first portion of the dielectric resonator element such that the first portion is surrounded by the ring element; the dielectric resonator element having a second portion that extends from the opening in a direction away from the substrate. ​

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