A Low Rectifier Current Ripple LLC Converter Based on a Non-Equivariant Ratio Series Input Transformer

Abstract

The present invention discloses a low-rectifier-current-ripple LLC converter based on a non-equivalent-ratio series-input transformer, comprising: an inverter circuit, a resonant circuit, and a rectifier circuit; the inverter circuit converts direct current into a pulsating square wave, the resonant circuit adopts two non-equivalent-ratio transformers with series-connected primary sides, denoted as the first transformer and the second transformer, and the turns ratios and the inductance values of the exciting inductances of the transformers are matched to construct different transformer exciting currents, and the difference between the exciting currents is used to increase the square-wave approximation degree of the transformer secondary-side current. The magnitude of the difference between the transformer exciting currents can be adjusted through the design of the converter parameters to adjust the smoothing effect on the rectified output current ripple. The rectifier circuit converts alternating current into direct current. While retaining the same soft-switching characteristics as the LLC converter, the present invention solves the problem of large rectified output current ripple existing in the LLC converter, reduces the variation range of the switching frequency, and can achieve high efficiency and higher power density.

Description

Technical Field

[0001] The present invention relates to the technical field of LLC resonant DC / DC converters in power electronic converters, especially LLC converters with a series input transformer. Background Art

[0002] LLC converters have excellent soft-switching characteristics, including zero-voltage switching (ZVS) of the primary switching tube within the entire operating range and zero-current switching (ZCS) of the secondary rectifier diode within the under-resonant region. Thanks to these characteristics, LLC converters are widely used in many DC / DC power fields such as battery energy storage, server power supplies, and new energy electric vehicles. However, correspondingly, LLC converters also have some problems. One important problem is the large pulsating current at the rectifier output. Since the LLC converter is essentially a series-parallel resonant converter, its working principle determines that only a capacitive filter can be used on the output side. Then the pulsating current output by the rectifier bridge is completely filtered by the capacitor in order to obtain a flat output current after filtering. At this time, the filtering capacitor has a problem of large current pulsation. Since electrolytic capacitors themselves are not tolerant of pulsating current, capacitors tolerant of pulsating current are often used, such as the scheme of using film capacitors, multi-layer ceramic (MLCC) capacitors in combination with electrolytic capacitors. On the one hand, the film capacitors and MLCC capacitors added to meet the requirement of low output pulsating current will correspondingly increase the circuit cost and also reduce the power density of the converter. On the other hand, even when capacitors tolerant of pulsating current are used in combination, the electrolytic capacitor is still an important factor affecting the reliability of the whole converter. In the case of long-term exposure to pulsating current, the life of the electrolytic capacitor will decrease, and the greater the pulsating current it withstands, the more obvious the reduction in life. Summary of the Invention

[0003] The technical problem to be solved by the present invention is to provide a low-rectifier-current-pulsation LLC converter based on a non-equivalent-ratio series input transformer, which can solve the problem of large pulsating current at the rectifier output of the LLC converter and at the same time take into account the requirement of high efficiency.

[0004] To solve the above technical problem, the present invention provides a low-rectifier-current-pulsation LLC converter based on a non-equivalent-ratio series input transformer, including: an inverter circuit, a resonant circuit, and a rectifier circuit; the inverter circuit converts direct current into a pulsating square wave, the resonant circuit uses two non-equivalent-ratio transformers with series-connected primary sides, denoted as the first transformer and the second transformer. By matching the turns ratio and the inductance value of the exciting inductance of each transformer, different transformer exciting currents are constructed. The difference between the exciting currents is used to increase the square-wave approximation of the transformer secondary current. By designing the converter parameters, the magnitude of the difference between the transformer exciting currents is adjusted to realize the adjustment of the suppression effect on the rectifier output current pulsation, and the rectifier circuit converts alternating current into direct current.

[0005] Preferably, the inverter circuit is selected from a full-bridge circuit, a half-bridge circuit, a multi-level circuit, or other circuits that can generate a square-wave or quasi-rectangular-wave AC output.

[0006] Preferably, the rectifier circuit is selected from an uncontrolled full-bridge rectifier, a full-wave rectifier circuit, or a fully-controlled full-bridge rectifier or full-wave rectifier circuit composed of fully-controlled power devices.

[0007] Preferably, the resonant circuit includes a resonant capacitor, a resonant inductor, a first transformer, and a second transformer. The resonant capacitor is connected in series with the resonant inductor, then in series with the first transformer, and then in series with the second transformer.

[0008] Preferably, the resonant inductor is an independent inductor or the primary leakage inductance of a transformer, or is composed of an independent inductor and the leakage inductance of a transformer.

[0009] Preferably, the rectified output of the secondary side of the first transformer and the rectified output of the secondary side of the second transformer are connected in parallel to the output filter capacitor.

[0010] Preferably, both transformers are single transformers or equivalent transformers obtained by series and / or parallel connection of multiple transformers.

[0011] Preferably, the two transformers are transformers with independent magnetic cores, or can also be magnetic integrated transformers composed of a single magnetic circuit combined magnetic core.

[0012] Preferably, the peak-to-peak value of the exciting current of the transformer with a larger turns ratio is greater than the peak-to-peak value of the exciting current of the transformer with a smaller turns ratio. The greater the difference in the peak-to-peak values of the exciting currents of the transformers, the more significant the effect of the converter in suppressing the current ripple of the secondary rectified output.

[0013] Preferably, when combining multiple transformers into one transformer, it can be either one magnetic core for each transformer, or a magnetic integrated transformer composed of a single magnetic core, or a combination of an independent magnetic core transformer and a magnetic integrated transformer.

[0014] Preferably, when the secondary rectifier circuit is a fully-controlled circuit, the fully-controlled device operates in a synchronous rectification mode.

[0015] The beneficial effects of the present invention are as follows: The resonant circuit of the present invention includes a resonant inductor, a resonant capacitor, and two transformers with non-equal turns ratios connected in series on the primary side. Each of the two transformers can be a single transformer or can be equivalently formed by multiple transformers as a whole. The resonant inductor, the resonant capacitor, and the exciting inductors of the two transformers in the resonant network can form an LLC resonance; the converter in the present invention has an inductive input impedance and inherits the advantages of zero-voltage switching (ZVS) of the primary-side switching tube and zero-current switching (ZCS) of the secondary-side rectifier diode of the LLC resonant converter; the converter in the present invention constructs different exciting currents by using non-equal ratio transformers, and uses the different exciting currents to increase the square-wave approximation of the secondary-side current of the transformer, which has the effect of suppressing the pulsation of the rectified output current, and the suppression effect of the rectified output current pulsation can be adjusted by adjusting the difference between the exciting currents of the transformers; in addition to the outstanding effects of reducing the filter capacitor, improving the power density and reliability, and reducing the cost brought by suppressing the rectified output pulsating current, when the same frequency conversion control is adopted under the same input and output conditions as the LLC resonant converter, the converter of the present invention can also reduce the variation range of the switching frequency, especially can reduce the lowest switching frequency, thereby improving the utilization rate of the magnetic core, reducing the volume and weight of the magnetic components, and further contributing to improving the power density and reducing the cost. BRIEF DESCRIPTION OF THE DRAWINGS

[0016] Figure 1 is a schematic diagram of the circuit structure of the converter of the present invention.

[0017] Figure 2 is the working waveform diagram of the present invention in Mode 1.

[0018] Figure 3(a) is the circuit modal diagram of the present invention in Mode 1 from t 0 to t 1 .

[0019] Figure 3(b) is the circuit modal diagram of the present invention in Mode 1 from t 1 to t 2 .

[0020] Figure 3(c) is the circuit modal diagram of the present invention in Mode 1 from t 2 to t 3 .

[0021] Figure 3(d) is the circuit modal diagram of the present invention in Mode 1 from t 3 to t 4 .

[0022] Figure 3(e) is the circuit modal diagram of the present invention in Mode 1 from t 4 to t 5 .

[0023] Figure 4 This is the working waveform diagram of the present invention in Mode 2.

[0024] Figure 5(a) is the circuit modal diagram of the present invention in Mode 2 from t 0 to t 1 .

[0025] Figure 5(b) is the circuit modal diagram of the present invention in Mode 2 from t 1 to t 2 .

[0026] Figure 5(c) is the circuit modal diagram of the present invention in Mode 2 from t 2 to t 3 .

[0027] Figure 5(d) is the circuit modal diagram of the present invention in Mode 2 from t 3 to t 4 .

[0028] Figure 6 This is the schematic diagram of the working waveform of the LLC using an equi-ratio series input transformer in the present invention.

[0029] Figure 7 This is the schematic diagram of the working waveform when the difference in exciting current designed in the present invention is relatively large.

[0030] Figure 8 This is the schematic diagram of the working waveform when the difference in exciting current designed in the present invention is relatively small. Detailed implementation manners

[0031] As Figure 1 shown, a low-rectifier-current-ripple LLC converter based on a non-equi-ratio series input transformer includes: an inverter circuit, a resonant circuit, and a rectifier circuit; the inverter circuit converts direct current into pulsating square waves, the resonant circuit uses two non-equi-ratio transformers with their primary sides in series, denoted as the first transformer and the second transformer, and matches the turns ratios and the inductance values of the exciting inductances of each transformer to construct different exciting currents of the transformers, increases the square-wave approximation degree of the secondary-side current of the transformer by using the difference between the exciting currents, and adjusts the size of the difference between the exciting currents of the transformers through the design of the converter parameters to achieve the adjustment of the suppression effect on the ripple of the rectified output current, and the rectifier circuit converts alternating current into direct current.

[0032] When the converter adopts the variable-frequency control method, it can adjust the magnitude of the output voltage or achieve a constant output voltage. When the switching operating frequency of the converter is less than the resonant frequency of the circuit, the converter operates in the boost and unity-gain mode (hereinafter referred to as "Mode 1"), and when the switching frequency is higher than the resonant frequency of the circuit, the converter operates in the buck mode (hereinafter referred to as "Mode 2").

[0033] Set the turns ratio of transformer T1 to be greater than that of transformer T2. The LLC converter using a non-equal turns ratio matrix transformer can be roughly divided into two operating modes, namely the boost and unity gain mode (Mode 1) and the buck mode (Mode 2). For ease of analysis, define the resonant frequency f of the converter r and the equivalent voltage gain M:

[0034]

[0035]

[0036] First, introduce Mode 1. When the converter operates in this mode, the switching frequency of the converter is in the region less than and near the resonant frequency fr, and the equivalent voltage gain of the converter is greater than 1 and near 1, which is the boost mode and the unity gain mode. Figure 2 Fig. 3 is the typical operating waveform of the converter in Mode 1.

[0037] Half of the operating cycle of Mode 1 can be divided into five operating modes.

[0038] [t0~t1]: As shown in Fig. 3(a), at t0, the switching tubes Q2 and Q3 are turned off, and the converter enters the dead time. The resonant current flows through the junction capacitance of the primary switching tubes. During this period, the resonant current charges the junction capacitance of Q1 and Q4 and discharges the junction capacitance of Q2 and Q3. The resonant current is clamped by iLm1 during t0~t1. The secondary side of T1 is open, and the primary side Lr1, Cr1, and Lm1 resonate. The absolute value of the amplitude of iLr1 is greater than iLm2, and energy flows from the primary side to the secondary side through T2, and the secondary side D6 and D7 are conducting.

[0039] [t1~t2]: As shown in Fig. 3(b), at t1, the junction capacitance of the switching tubes Q2 and Q3 is charged to V1, the junction capacitance of the switching tubes Q1 and Q4 is discharged, and the body diodes of Q1 and Q4 are conducting. After that, iLr1 starts to increase in the reverse direction, and a difference appears with iLm1. The secondary side D1 and D4 of T1 are conducting, the clamping voltage of Lm1 changes direction, iLm1 starts to increase in the reverse direction. Since the clamping voltage of Lm2 does not change direction, iLm2 continues to increase negatively. At the same time, the secondary side i1 increases and i2 decreases.

[0040] [t2~t3]: As shown in Fig. 3(c), at t2, the switching tubes Q1 and Q4 are turned on. Since before turning on, the DS voltage of Q1 and Q4 has been clamped to the conduction voltage drop of the body diode, Q1 and Q4 are approximately turned on with zero voltage. During t2~t3, the resonant currents iLr1 and iLm1 continue to increase in the reverse direction. The clamping voltage of Lm2 does not change direction, iLm2 continues to increase negatively. iLr1 operates between iLm2 and iLm1. The difference between iLr1 and iLm1 continues to increase, and the difference between iLr1 and iLm2 continues to decrease. i1 continues to increase and i2 continues to decrease.

[0041] [t3 - t4]: As shown in Fig. 3(d), at time t3, the resonant current iLr1 drops to the magnitude of iLm2, and the secondary current i2 of T2 drops to 0. Thereafter, the resonant current continues to rise in the reverse direction, the difference between iLr1 and iLm2 continues to increase, i2 continues to rise, D6 and D7 are cut off, D5 and D8 are turned on, the clamping voltage of Lm2 changes direction, and iLm2 begins to rise in the reverse direction. During t3 - t4, the resonant current is above iLm1 and iLm2, and D1, D4, D5, and D8 are turned on.

[0042] [t4 - t5]: As shown in Fig. 3(e), at time t4, the resonant current iLr1 drops to the magnitude of iLm1, and i1 drops to zero. Thereafter, the resonant current iLr1 is clamped by the current iLm1 of Lm1, the secondary side of T1 is open - circuited, D1 and D4 achieve zero - current turn - off, and the primary - side Lr1, Cr1, and Lm1 resonate in series. iLr1 is always greater than iLm2. The switching tubes D5 and D8 on the secondary side of T2 remain turned on. The secondary - side resonant current i2 of T2 is the difference between the primary - side resonant current and the exciting current iLm2, that is, N2(iLm1 - iLm2). Since both iLm1 and iLm2 change linearly during this period, i2 also changes linearly.

[0043] For comparative analysis, Figure 4 the typical operating waveforms of an LLC converter with an equal turns - ratio when the switching frequency is less than the resonant frequency are given. Comparing Figure 4 with Figure 2 , it can be found that the main differences between the two are that the non - equal - turns - ratio LLC converter has one more Figure 2 mode [t2 - t3] than the equal - turns - ratio LLC converter, and there are differences in the operating modes of the two converters during the series - resonance interval (i.e., Figure 2 [t4 - t5] in Figure 2 ). As known from Figure 2 , during [t2 - t3], the secondary - side current i1 of T1 changes approximately linearly and is more similar to a square wave. At the same time, during [t4 - t5], i2 changes linearly, which also increases the similarity between the waveform of i2 and a square wave, facilitating the realization of a low - ripple rectified output current. Observing

[0044] Figure 2 it is not difficult to find that the value of i1 at time t3 determines the square - wave approximation degree of i1, and i1(t3)=N1(iLm2 - iLm1). The value of i2 during the interval [t4 - t5] determines the square - wave approximation degree of i2, and during this period i2 = N2(iLm1 - iLm2). The greater the difference between iLm1 and iLm2 at time t3 and during the interval [t4 - t5], the more similar the secondary - side current of the converter is to a square wave, and the smaller the ripple of the rectified output current. It should be particularly noted that it is a prerequisite that the absolute value of iLm1 is greater than iLm2 at time t3 and during the interval [t4 - t5].

[0044] When the switching frequency is further away from the resonant frequency, the converter will enter the second operating mode, i.e., Mode 2. At this time, the equivalent voltage gain of the converter is less than 1, which is a buck mode. Fig. 5 shows the typical waveforms of the converter in Mode 2.

[0045] It can be seen from Figure 2 that during [t2 - t3], the secondary current i1 of T1 approximately linearly changes, and is more approximate to a square wave. At the same time, during [t4 - t5], i2 shows a linear change, which also increases the approximation degree of the waveform of i2 to a square wave, facilitating the realization of a low rectified output pulsating current. Observing Figure 2 it is not difficult to find that the value of i1 at t3 determines the square wave approximation degree of i1, and i1(t3) = N1(iLm2 - iLm1). The value of i2 during the interval [t4 - t5] determines the square wave approximation degree of i2, and during this period, i2 = N2(iLm1 - iLm2). The greater the difference between iLm1 and iLm2 at t3 and during the interval [t4 - t5], the more approximate the secondary current of the converter is to a square wave, and the smaller the pulsation of the rectified output current. It is particularly worth noting that it is a prerequisite that the absolute value of iLm1 is greater than iLm2 at t3 and during the interval [t4 - t5].

[0046] Half a cycle of Mode 2 can be divided into 4 operating modes.

[0047] [t0 - t1]: As shown in Fig. 5(a), at t0, the switching tubes Q2 and Q3 are turned off, and the converter enters the dead time. The resonant current flows through the junction capacitances of the primary switching tubes. During this period, the resonant current charges the junction capacitances of Q1 and Q4 and discharges the junction capacitances of Q2 and Q3. The amplitude of the resonant current rapidly drops. Since the resonant current is still greater than the magnetizing currents of transformers T1 and T2 at this time, the secondary rectifier diodes D2, D3, D6, and D7 remain conducting.

[0048] [t1 - t2]: As shown in Fig. 5(b), at t1, the junction capacitances of the switching tubes Q2 and Q3 are charged to V1, the junction capacitances of the switching tubes Q1 and Q4 are discharged, the body diodes of Q1 and Q4 conduct. At the same time, the resonant current drops to the same magnitude as iLm1, and the secondary current i1 of T1 drops to 0. After that, the resonant current continues to drop, and the resonant current always operates above iLm1, i1 continues to rise, D2 and D3 turn off, D1 and D4 conduct, the clamping voltage of Lm1 commutes, and iLm1 starts to rise reversely. During t1 - t2, iLm2 is still less than the resonant current, D6 and D7 remain conducting, and the amplitude of iLm2 continues to rise negatively.

[0049] [t2 - t3]: As shown in Fig. 5(c), at time t2, switches Q1 and Q4 turn on. Since the DS voltages of Q1 and Q4 have been clamped to the forward voltage drop of the body diodes before turning on, Q1 and Q4 turn on at approximately zero voltage. The resonant currents iLr1 and iLm1 continue to decrease, and iLm2 continues to rise negatively because the clamping voltage of Lm2 has not commutated.

[0050] [t3 - t4]: As shown in Fig. 5(d), at time t3, the resonant current iLr1 decreases to the magnitude of iLm2, and the secondary current i2 of T2 decreases to 0. After that, the resonant current continues to decrease. The resonant current is above iLm2, and i2 continues to rise. D6 and D7 are turned off, and D5 and D8 are turned on. The clamping voltage of Lm2 commutates, and iLm2 begins to rise in the reverse direction. During t3 - t4, the resonant current is above iLm1 and iLm2, and D1, D4, D5, and D8 are turned on.

[0051] During the [t2 - t3] mode, the secondary current i1 of T1 changes approximately linearly, and i1 is closer to a square wave, enabling the converter to rectify and output a current with lower ripple. By observing Fig. 5, it can be found that the square wave approximation degree of i1 is determined by the value of i1 at time t3. At time t3, i1(t3) = N1(iLm2 - iLm1). The greater the difference between iLm1 and iLm2, the smaller the ripple of the rectified output current of the converter at this time. Similarly, it is a prerequisite that the absolute value of iLm1 is greater than iLm2.

[0052] The key to the new scheme for suppressing the pulsating current proposed by the present invention lies in using the non - equal turns ratio of the transformer to construct different exciting currents, where the requirements are as follows: on the premise of N1 > N2, in Mode 1, it is satisfied that the absolute value of iLm1 is greater than iLm2 during the time t3 and in the interval [t4 - t5], and the greater the difference between iLm1 and iLm2 during the time t3 and in the interval [t4 - t5], the smaller the ripple of the rectified output current; when working in Mode 2, it is satisfied that at time t3, the absolute value of iLm1 is greater than iLm2, and the greater the difference between iLm1 and iLm2, the smaller the ripple of the rectified output current of the converter at this time. At the same time, it can be found that when the converter works in Mode 1, the converter can simultaneously achieve ZVS of the primary - side switches and ZCS of the secondary - side rectifier diodes of the large - turns - ratio transformer. Since the power distribution ratio of the transformer can be approximately regarded as the ratio of the turns ratio, that is to say, the secondary - side rectifier diodes of the main - power transformer can achieve ZCS. Therefore, in order to achieve high efficiency, when designing the parameters of the non - equal turns ratio LLC converter, the converter can be made to work in Mode 1 as much as possible.

[0053] For the structure as Figure 1The converter shown is simulated. Through simulation tests, it can be seen that the solution proposed in the present invention can effectively reduce the pulsation of the rectified output current. At the same time, by setting the designed excitation current difference to different values, the suppression effect of the converter in the present invention on the pulsation of the output rectified current is different. The following takes the LLC with an equal-ratio series input transformer (representing the traditional LLC), the LLC proposed in the present invention with a non-equal turns ratio series input transformer and the peak-to-peak difference of the excitation current being A (a suitable value), and the LLC with a non-equal turns ratio series input transformer and the peak-to-peak difference of the excitation current being B (a suitable value) as examples, where A > B. Figure 6 is the working waveform of the LLC with an equal-ratio series input transformer. Figure 7 is the working waveform of the LLC with a non-equal turns ratio series input transformer and the peak-to-peak difference of the excitation current being A. As can be seen from the figure, the effective value of the pulsating current flowing through the output filter capacitor of the latter is significantly smaller than that of the former. Therefore, it shows that the LLC with a non-equal turns ratio series input transformer can improve the problem of large pulsation of the LLC rectified output current. Figure 8 is the working waveform of the LLC with a non-equal turns ratio series input transformer and the peak-to-peak difference of the excitation current being B. As can be seen from the figure, compared with the case where the peak-to-peak difference of the designed excitation current is A, the effective value of the pulsating current borne by the output filter capacitor with the peak-to-peak difference of the excitation current set to B is larger. Therefore, the larger the designed excitation current difference is set, the better the suppression effect of the converter in the present invention on the pulsation of the output rectified current. It is worth noting that the pulsating current borne by the filter capacitor of the converter proposed in the present invention at this time is still smaller than that of the LLC with an equal-ratio series input transformer.

Claims

1. A low rectifier current pulsation LLC converter based on a non-equal ratio series input transformer, characterized in that: include: An inverter circuit, a resonant circuit and a rectifier circuit; the inverter circuit converts direct current into a pulsating square wave, the resonant circuit includes a resonant capacitor, a resonant inductor, a first transformer and a second transformer, the resonant capacitor is connected in series with the resonant inductor, then connected in series with the first transformer, and then connected in series with the second transformer, the corresponding excitation current peak-to-peak value of the transformer with a larger transformation ratio is greater than the excitation current peak-to-peak value of the transformer with a smaller transformation ratio, the resonant circuit uses two transformers with non-equal transformation ratios connected in series on the primary side, recorded as the first transformer and the second transformer, the transformation ratios and excitation inductances of each transformer are matched to construct different transformer excitation currents, the square wave approximation of the transformer secondary current is increased by using the difference between the excitation currents, the difference between the transformer excitation currents is adjusted by converter parameter design to achieve the smoothing effect of the rectifier output current pulsation, the larger the excitation current difference is set, the better the suppression effect, the rectifier circuit converts alternating current into direct current, the rectifier output of the secondary side of the first transformer and the rectifier output of the secondary side of the second transformer are connected in parallel to the output filter capacitor, and the converter adjusts the output voltage or achieves the constant output voltage when adopting the variable frequency control mode.

2. The low rectifier current pulsation LLC converter based on non-equal ratio series input transformer according to claim 1, characterized in that: The inverter circuit selects a full-bridge circuit, a half-bridge circuit, a multi-level circuit or other circuits that can generate square wave or quasi-rectangular wave AC output.

3. The low rectifier current pulsation LLC converter based on non-equal ratio series input transformer according to claim 1, characterized in that: The rectifier circuit selects an uncontrolled full-bridge rectifier, a full-wave rectifier circuit or a fully-controlled full-bridge rectifier, a full-wave rectifier circuit composed of fully-controlled power devices.

4. The low rectifier current pulsation LLC converter based on non-equal ratio series input transformer according to claim 1, characterized in that: The resonant inductor is an independent inductor or a primary leakage inductor of a transformer, or is composed of an independent inductor and a primary leakage inductor of a transformer.

5. The low rectifier current pulsation LLC converter based on non-equal ratio series input transformer according to claim 1, characterized in that: Both transformers are single transformers or equivalent transformers obtained by connecting a plurality of transformers in series or / and in parallel; both transformers are magnetic integrated transformers formed by a single magnetic core with a magnetic circuit combination.

6. The low rectifier current pulsation LLC converter based on non-equal ratio series input transformer according to claim 1, characterized in that: When a transformer is composed of multiple transformers, each transformer may have a magnetic core, or a magnetically integrated transformer composed of a single magnetic core may be used, or a combination of an independent magnetic core transformer and a magnetically integrated transformer may be used.

7. The low rectifier current pulsation LLC converter based on non-equal ratio series input transformer according to claim 5, characterized in that: When the secondary side rectification circuit is a fully controlled circuit, the fully controlled device adopts a synchronous rectification working mode.

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