NLADRC Voltage Control Method for an Efficient Power Converter Used in Electric Vehicles

Abstract

The present invention discloses an NLADRC voltage control method for an efficient power converter for electric vehicles, which relates to the technical field of NLADRC voltage control for efficient power converters for electric vehicles, and includes establishing an equivalent model of a CLLLC resonant converter based on the fundamental wave analysis method; solving the voltage gain by using a linear circuit solution method through the FHA equivalent model of the CLLLC resonant converter; and establishing an improved anti-integral saturation NLADRC voltage control strategy. The method of the present invention significantly improves the anti-disturbance ability and stability by applying an improved non-linear auto-disturbance rejection control strategy with anti-integral saturation. By using a non-linear filtering function, this strategy effectively suppresses the fluctuations in the response, accelerates the stabilization speed, and improves the overall control stability and robustness.

Description

Technical Field

[0001] The present invention relates to the technical field of electric vehicles, and particularly to a NLADRC voltage control method for an efficient power converter for electric vehicles. Background Art

[0002] With the booming development of the electric vehicle industry, as a core component of electric vehicles, the research direction of on-board chargers mainly focuses on high-frequency efficiency, high power density, and high-performance control technologies. Gallium nitride high electron mobility transistor (GaN High Electron Mobility Transistor, GaN HEMT) has characteristics such as fast switching speed and low loss, which is beneficial for on-board chargers to increase the switching frequency and the overall efficiency of the machine, but also brings challenges in circuit topology and control technology. Therefore, in this paper, an in-depth study is carried out on the control method of a two-stage bidirectional on-board charger based on GaN HEMT.

[0003] The on-board charger must first meet the requirements of power quality, that is, it has functions such as improving the power factor and reducing the current THD to reduce harmonic pollution to the power grid; secondly, it needs to have a large output voltage range to meet the charging requirements of the battery. Since GaN HEMT can effectively suppress the reverse recovery effect, the totem-pole bridgeless PFC has been widely used and has become a research hotspot in high-frequency PFC topologies. The latter stage of the on-board charger mainly provides a safe and reliable wide-range voltage output. Currently, CLLLC resonant converters are mostly used. The prior art has studied the synchronous rectification of CLLLC resonant converters to further reduce the conduction loss brought by the body diode.

[0004] Aiming at problems such as poor disturbance rejection ability of the CLLLC resonant converter, a nonlinear active disturbance rejection voltage control strategy based on improved anti-integral saturation is proposed. By improving the anti-integral saturation design and the nonlinear filtering function, the stability and robustness of the system are effectively improved. Summary of the Invention

[0005] The purpose of this part is to outline some aspects of the embodiments of the present invention and briefly introduce some preferred embodiments. Some simplifications or omissions may be made in this part, as well as in the abstract and title of the specification of this application, to avoid obscuring the purpose of this part, the abstract, and the title of the invention. However, such simplifications or omissions cannot be used to limit the scope of the present invention.

[0006] In view of the problems existing in the above or prior art, the present invention is proposed.

[0007] Therefore, the technical problem solved by the present invention is: the poor disturbance rejection ability of the CLLLC resonant converter.

[0008] In order to solve the above technical problems, the present invention provides the following technical solutions: a NLADRC voltage control method for a high-efficiency power converter for electric vehicles, which includes establishing a CLLLC resonant converter equivalent model based on a fundamental wave analysis method;

[0009] The voltage gain is solved by using the FHA equivalent model of the CLLLC resonant converter and the linear circuit solution method.

[0010] An improved anti-integral saturation NLADRC voltage control strategy is established.

[0011] As a preferred solution of the NLADRC voltage control method of the high-efficiency power converter for electric vehicles of the present invention, wherein: the CLLLC resonant converter equivalent model established based on the fundamental wave analysis method includes:

[0012] The full-bridge equivalent replacement of the CLLLC resonant converter is performed, and the secondary side of the transformer is converted to the primary side to obtain the FHA model of the CLLLC resonant converter; where V AB_FHA It is the sine wave voltage equivalent to the voltage between the midpoints of the inverter bridge arm through FHA; L r1 and C r1 are the resonant inductance and capacitance of the primary and primary sides respectively; let n p is the transformer ratio, the resonant inductance of the secondary side converted to the primary side Resonant capacitor Req is the load obtained by using FHA to equalize the resistive load on the output side;

[0013] Due to the switching of the switching devices in the inverter unit, the voltage between the midpoints of the bridge arms is actually V in and -V in The square wave voltage that jumps between is shown as follows:

[0014]

[0015] Among them, T s is the switching period, i.e. T s =1 / f s , where f s is the switching cycle;

[0016] Performing Fourier transform on the above equation gives the following equation:

[0017]

[0018] Where n is the integer parameter of the Fourier series expansion;

[0019] In the fundamental wave analysis method, it is believed that the energy of the input signal mainly comes from the fundamental wave component, and the remaining high-order harmonics are negligible; the fundamental wave component and its effective value are extracted as follows:

[0020]

[0021] Since the input of CLLLC needs to pass through the inverter unit, the actual current flowing in the resonant cavity is an alternating current. According to the idea of the fundamental wave analysis method, it is equivalent to a sine wave as follows:

[0022]

[0023] I r1 is the effective value of the resonant current i r1 (t) in the primary side of the transformer, is V AB_FHA (t) and the phase difference between i r1 (t). Similarly, the fundamental component and its effective value of the voltage between the midpoints C and D of the two bridge arms of the rectifier unit are obtained:

[0024]

[0025] where θ is the phase difference of the output side relative to the input side;

[0026] The current in the secondary resonant cavity is expressed as:

[0027]

[0028] where I r2 is the effective value of the resonant current ir2(t) in the secondary side of the transformer;

[0029] The resonant cavity passes through the rectifier unit to obtain the output current, so the output average current is:

[0030]

[0031] where I o is the output current, P o is the output power, V o is the output voltage, and R o is the output resistive load;

[0032] It can be seen from the above formula that V CD_FHA (t) and i r2 (t) are in the same phase. Therefore, Ro is equivalent to the equivalent output resistance Req of the FHA model:

[0033]

[0034] As a preferred scheme of the NLADRC voltage control method for the high-efficiency power converter for electric vehicles described in the present invention, wherein: for the FHA equivalent model based on the CLLLC resonant converter, by using the linear circuit solving method, the voltage gain is solved, including

[0035] The voltage gain of the CLLLC resonant converter is expressed as:

[0036]

[0037] Wherein,

[0038]

[0039] Where ω s is the switching angular frequency;

[0040] The CLLLC resonant converter is defined as follows:

[0041]

[0042] Let x = y = 1, then the structure of the converter can achieve bidirectional symmetry, realize the consistency of bidirectional voltage gain and control, and simplify parameter design and control;

[0043] To further obtain the gain expression of the converter, it is necessary to take the normalized frequency f n = f s / f r = ω s / ω r , f s and ω s are the actual switching frequency and the corresponding angular frequency; The characteristic impedance The quality factor Q = Z r / R eq , Substituting the above definitions into the formula, the gain expression of the converter can be obtained as:

[0044]

[0045] As a preferred scheme of the NLADRC voltage control method for the high-efficiency power converter for electric vehicles described in the present invention, wherein: The establishment of the improved anti-integral saturation NLADRC voltage control strategy includes,

[0046] Designing the voltage controller based on LADRC;

[0047] Designing the voltage controller based on the improved anti-integral saturation NLADRC.

[0048] As a preferred scheme of the NLADRC voltage control method for the high-efficiency power converter for electric vehicles described in the present invention, wherein: The design of the voltage controller based on LADRC includes,

[0049] Reducing the CLLLC resonant converter to a second-order model, and its transfer function structure is shown as the following formula:

[0050]

[0051] According to the above formula, let y be the output quantity, i.e., the secondary side output, and u be the input quantity, i.e., the primary side DC voltage input. Then we have:

[0052]

[0053] Among them, a0, a1, c0, c1, and b0 are undetermined coefficients in the LADRC. a0, a1, c0, and c1 are related to reducing the mathematical model of the converter to the second order.

[0054] Select the state variable x as Then the state variable includes the total disturbance, and the corresponding state equation is written as:

[0055]

[0056] Therefore, a third-order LESO is established as:

[0057]

[0058] Among them, β1, β2, and β3 are observer parameters; z1 and z2 are the observed values of the system state variables, and z3 is the observation of the total disturbance of the controlled object. If the parameters are set appropriately, the correct state variable x can be estimated.

[0059] In order to achieve the expected control, the LSEF is set as:

[0060]

[0061] Among them, k p 、k d are controller parameters; r is the given target value.

[0062] As a preferred scheme of the NLADRC voltage control method for the high-efficiency power converter for electric vehicles described in the present invention, among them: the design of the voltage controller based on the improved anti-integral saturation NLADRC includes,

[0063] The NLADRC is realized by adding the following nonlinear filtering function on the basis of the LADRC:

[0064]

[0065] Among them, e is the error; α is the nonlinear factor, which characterizes the nonlinear degree of the filtering function, and 0 ≤ α ≤ 1; δ is the filtering factor, which characterizes the width of the linear interval.

[0066] The following function is used as the nonlinear filtering function for integral anti-saturation:

[0067]

[0068] Among them, e is the error, ke is the proportionality coefficient with ke > 0, Klim is the upper and lower limits of integration, and ε is the maximum error.

[0069] Advantages of the present invention: By applying the improved anti-integral saturation non-linear active disturbance rejection control strategy, the present invention significantly improves the disturbance rejection ability and stability. Using the non-linear filtering function, this strategy effectively suppresses the fluctuations in the response, speeds up the stabilization speed, and enhances the stability and robustness of the overall control. At the same time, the improved anti-integral saturation algorithm prevents the saturation problem of the integral term, making the voltage control more accurate and enhancing the fast response ability. These improvement measures can maintain a stable voltage output and excellent control accuracy in the face of different load disturbances. Description of the Drawings

[0070] To more clearly illustrate the technical solutions of the embodiments of the present invention, the drawings required for the description of the embodiments will be briefly introduced below. Obviously, the drawings in the following description are only some embodiments of the present invention. For those of ordinary skill in the art, other drawings can be obtained based on these drawings without creative efforts. Among them:

[0071] Figure 1 It is a topological structure diagram of a resonant converter.

[0072] Figure 2 It is a FHA model diagram of a CLLLC resonant converter.

[0073] Figure 3 It is a converter gain curve diagram with different k values under the same Q value.

[0074] Figure 4 It is a converter gain curve diagram with different Q values under the same k value.

[0075] Figure 5 It is a LADRC control block diagram of a CLLLC resonant converter.

[0076] Figure 6 It is an Anti-reset Windup structure diagram.

[0077] Figure 7 It is an improved Anti-reset Windup structure diagram.

[0078] Figure 8 It is an image diagram of a non-linear filtering function.

[0079] Figure 9 It is a comparison waveform diagram of the disturbance rejection performance of the converter before and after using ADRC.

[0080] Figure 10 Output voltage waveform diagram of a converter using improved NLADRC voltage control.

[0081] Figure 11 Waveform diagram of the anti-load disturbance comparison experiment. Specific implementation manners

[0082] To make the above objects, features and advantages of the present invention more obvious and understandable, the specific implementation manners of the present invention will be described in detail below with reference to the accompanying drawings of the specification.

[0083] In the following description, many specific details are set forth to facilitate a full understanding of the present invention. However, the present invention may be implemented in other ways different from those described herein, and those skilled in the art may make similar extensions without departing from the connotation of the present invention. Therefore, the present invention is not limited by the specific embodiments disclosed below.

[0084] Secondly, the so-called "one embodiment" or "embodiment" herein refers to a specific feature, structure or characteristic that may be included in at least one implementation manner of the present invention. The appearances of "in one embodiment" in different places in this specification do not all refer to the same embodiment, nor are they separate or alternative embodiments that exclude each other with other embodiments.

[0085] Embodiment 1

[0086] Referring to Figures 1 to 8 , which is an embodiment of the present invention, a NLADRC voltage control method for an efficient power converter for electric vehicles is provided, including:

[0087] S1: Establish an equivalent model of the CLLLC resonant converter based on the fundamental wave analysis method.

[0088] It can be seen from Figure 1 that the topological structure of the CLLLC mainly consists of three parts: an inverter unit, a resonant unit, and a rectifier unit.

[0089] Furthermore, the establishment of the equivalent model of the CLLLC resonant converter based on the fundamental wave analysis method includes

[0090] performing a full-bridge equivalent substitution on the CLLLC resonant converter, and converting the secondary side of the transformer to the primary side to obtain the FHA model of the CLLLC resonant converter;

[0091] where V AB_FHA is a sinusoidal voltage equivalent to the voltage between the midpoints of the inverter unit bridge arms through FHA; L r1 and C r1 are the resonant inductor and resonant capacitor on the primary side of the primary side respectively; let n p be the transformer turns ratio, and the resonant inductor on the secondary side converted to the primary side Resonant capacitor Req is the load equivalent by using FHA for the resistive load on the output side, and the specific calculation process will be given below.

[0092] Due to the switching of the switching devices in the inverter unit, the voltage between the midpoints of the bridge arms is actually a square wave voltage that jumps between V in and -V in as shown in the following equation:

[0093]

[0094] where, T s is the switching period, that is, T s = 1 / f s where f s is the switching frequency;

[0095] Performing Fourier transform on the above equation gives the following equation:

[0096]

[0097] where, n is the integer parameter of the Fourier series expansion;

[0098] In the fundamental wave analysis method, it is considered that the energy of the input signal mainly comes from the fundamental wave component, and the remaining high-order harmonic components are ignored; the fundamental wave component and its effective value can be extracted from the above equation and expressed as:

[0099]

[0100] Since the input of CLLLC needs to pass through the inverter unit, the actual current flowing in the resonant cavity is an alternating current. According to the idea of the fundamental wave analysis method, it is equivalent to a sine wave as shown in the following equation:

[0101]

[0102] I r1 is the effective value of the resonant current i r1 (t) on the primary side of the transformer, is the phase difference between V AB_FHA (t) and i r1 (t). Similarly, the fundamental wave component and its effective value of the voltage between the midpoints CD of the two bridge arms of the rectifier unit are obtained:

[0103]

[0104] where, θ is the phase difference of the output side relative to the input side;

[0105] The current in the secondary resonant cavity is expressed as:

[0106]

[0107] Among them, I r2 is the effective value of the resonant current ir2(t) on the secondary side of the transformer;

[0108] The resonant cavity obtains the output current through the rectification unit, so the output average current is:

[0109]

[0110] Among them, I o is the output current, P o is the output power, V o is the output voltage, R o is the output resistive load;

[0111] It can be seen from the above formula that V CD_FHA (t) and i r2 (t) are in the same phase, so Ro is equivalent to the equivalent output resistance Req of the FHA model:

[0112]

[0113] S2: Through the FHA equivalent model of the CLLLC resonant converter, using the linear circuit solution method to solve the voltage gain.

[0114] Furthermore, the voltage gain is solved by the linear circuit solution method through the FHA equivalent model of the CLLLC resonant converter, including

[0115] The voltage gain of the CLLLC resonant converter is expressed as:

[0116]

[0117] Among them,

[0118]

[0119] Among them, ω s is the switching angular frequency;

[0120] The following definitions are made for the CLLLC resonant converter:

[0121]

[0122] The resonant network of the CLLLC resonant converter consists of five components, and there are theoretically multiple resonant points, which increases the difficulty of parameter design and topology control. Analysis Figure 2 shows that when x = y = 1, the structure of the converter can achieve bidirectional symmetry, realizing the consistency of bidirectional voltage gain and control, and simplifying parameter design and control;

[0123] To further obtain the gain expression of the converter, the normalized frequency f n = f s / f r = ω s / ω r , f s and ω s are the actual switching frequency and the corresponding angular frequency; the characteristic impedance The quality factor Q = Z r / R eq . Substituting the above definitions into the formula, the gain expression of the converter can be obtained as:

[0124]

[0125] Furthermore, according to the above formula, the converter frequency response gain curves with different k values under the same Q value and the converter frequency response gain curves with different Q values under the same k value are respectively as Figure 3 , shown in Figure 4

[0126] It should be noted that from Figure 3 it can be seen that when the quality factor Q remains unchanged, that is, under the condition of constant load, on the one hand, as the k value increases, the maximum gain of the converter decreases, the switching frequency corresponding to the maximum gain decreases, and the under-resonant amplification region gradually becomes non-linear. If the k value is too large, the converter will even lose the amplification ability; on the other hand, as the k value increases, the minimum gain will also increase, and the gain adjustment ability in the over-resonant region gradually becomes worse. Therefore, from this perspective, the k value should be taken as small as possible to ensure that the converter has a wide gain range in the over-resonant state

[0127] From Figure 4 it can be known that when the k value is fixed, the quality factor Q represents the weight of the load. The larger the quality factor Q, the heavier the load. At the same frequency, the voltage gain is also smaller. When the load is too heavy, in the under-resonant region, the linear regulation ability of the frequency to the gain will be lost, and even the situation where the gain is less than 1 will occur. Therefore, when designing parameters, the maximum load-carrying capacity of the converter also needs to be considered, and the maximum quality factor should be set reasonably

[0128] S3: Establish an improved anti-integral saturation NLADRC voltage control strategy

[0129] The charging process of the OBC is relatively complex and requires double closed-loop control of the voltage and current loops to achieve. Similar to the speed limit loop of the motor, the voltage loop acts as a voltage limit loop to limit the charging voltage, and the current loop controls the charging current according to the charging efficiency curve of the power battery. For the CLLLC converter, PI control is sufficient to achieve the control, but the actual working conditions of the OBC are complex and there are various disturbances during operation. The anti-disturbance ability of PI control is poor and it is difficult to maintain stable operation

[0130] Therefore, in view of the anti-disturbance problems such as load disturbance of the CLLLC converter, an improved anti-integral saturation NLADRC voltage control strategy is proposed in this paper to improve the anti-disturbance performance of the converter. Based on the Linear Active Disturbance Rejection Control (LADRC), this strategy adds a non-linear filtering function and improves the Anti-reset Windup integral anti-saturation algorithm for the integral variables inside the control system to accelerate the convergence of internal variables and the stability of the system.

[0131] Furthermore, the establishment of the improved anti-integral saturation NLADRC voltage control strategy includes:

[0132] Designing the voltage controller based on LADRC;

[0133] Designing the voltage controller based on the improved anti-integral saturation NLADRC.

[0134] Even further, the design of the voltage controller based on LADRC includes:

[0135] Reducing the order of the CLLLC resonant converter to a second-order model, and its transfer function structure is shown as follows:

[0136]

[0137] Linear Active Disturbance Rejection Control mainly includes two parts: the Extended State Observer (ESO) and the Linear State Error Feedback (LSEF). The ESO is the core of LADRC, which is used to estimate the current state variables according to the total disturbance of the system. The LSEF is a control law similar to negative feedback. Compared with the PI control, the LSEF no longer simply subtracts the given value from the feedback, but adds the differential components of each order of state variables to make up for the deficiencies of the PI in terms of rapidity, etc. The control block diagram of LADRC is as Figure 5 shown

[0138] According to the above formula, let y be the output quantity, that is, the secondary side output, and u be the input quantity, that is, the primary side DC voltage input. Then:

[0139]

[0140] Among them, a0, a1, c0, c1, and b0 are undetermined coefficients in LADRC. a0, a1, c0, and c1 are related to reducing the order of the mathematical model of the converter to the second order;

[0141] Select the state variable x as The state variables then contain the total disturbance, and the corresponding state equation is written as:

[0142]

[0143] Therefore, a third-order LESO is established as:

[0144]

[0145] where β1, β2, and β3 are the observer parameters; z1 and z2 are the observed values of the system state variables, and z3 is the observed total disturbance of the controlled object; if the parameters are set appropriately, the correct state variable x can be estimated.

[0146] To achieve the desired control, the LSEF is set as:

[0147]

[0148] where k p and k d are the controller parameters; r is the given target value.

[0149] Furthermore, the design of the voltage controller based on the improved anti-integral saturation NLADRC includes

[0150] NLADRC is implemented by adding the following non-linear filtering function on the basis of LADRC:

[0151]

[0152] where e is the error; α is the non-linear factor, which characterizes the non-linearity of the filtering function, 0 ≤ α ≤ 1; δ is the filtering factor, which characterizes the width of the linear interval.

[0153] It should be noted that the non-linear filtering function fal(e, α, δ) has the characteristics of large gain for small errors and small gain for large errors, ensuring that the estimated value can converge faster and track the given value, and the estimated value will be relatively more stable.

[0154] As can be seen from the third-order LESO, the observed values of the system variables and the disturbance observed values z1, z2, and z3 all need to be integrated from the differential estimated values obtained by the ESO, and they all belong to the integral terms, which have an important impact on the convergence, stability, and rapidity of the system. Since there is a saturation problem in the integral terms, an improved integral anti-saturation algorithm is adopted in this paper. This method is based on Anti-resetWindup, and its structure is as follows Figure 6As shown, this method is simple and effective but lacks robustness, has a large degree of arbitrariness in compensation, and has low precision. On this basis, this paper improves it. Similarly, a nonlinear filtering function is set for the integral term to achieve the effect of amplifying small errors and saturating large errors, accelerating the integral stability and anti-saturation. The improved Anti-reset Windup structure is as Figure 7 shown.

[0155] The role of the nonlinear filtering function is to amplify small errors and saturate large errors. The saturation effect can accelerate the integral desaturation and prevent the integral from accumulating infinitely and diverging. The amplification effect is to enable the integral to converge quickly. The conventional linear function with upper and lower limits does not have an amplification effect within the limit range, and the function transition stage is not smooth. Therefore, the following function is used as the nonlinear filtering function for integral anti-saturation in this paper:

[0156]

[0157] where e is the error, k e is the proportional coefficient with k e >0, 0 < χ < 1, K lim is the upper and lower limits of the integral, and ε is the maximum error. The waveform of its function is as Figure 8 shown.

[0158] In summary, the present invention accelerates the convergence of the internal parameters of the ADRC by adding a nonlinear filtering function and an improved anti-integral saturation algorithm, inherits the excellent disturbance rejection ability of the LADRC, improves the robustness and rapidity of the control system, and improves the output disturbance rejection performance to cope with complex working conditions.

[0159] Embodiment 2

[0160] Referring to Figures 9 to 11 , as the second embodiment of the present invention, a NLADRC voltage control method for an electric vehicle high-efficiency power converter is provided and simulated and verified.

[0161] CLLLC Resonant Converter Simulation and Experiment

[0162] Simulation Analysis

[0163] As Figure 9 shown is the simulation comparison of load disturbance based on PI control and LADRC control.

[0164] Among them, Fig. (a) is the output voltage waveform diagram of the converter based on PI control. A load disturbance is applied at 0.1 s. It can be seen that the voltage rises to 370 V and returns to stability after about 10 ms. Fig. (b) is the output voltage waveform of the converter based on LADRC. When the load is suddenly reduced at 0.25 s, the voltage does not change significantly. It can be seen that the disturbance rejection performance of the converter is greatly improved after adopting ADRC. However, the rapidity of the converter decreases significantly. It takes about 0.22 s to achieve stable output with an overshoot of about 4.1%. While the converter under PI control can complete stable control in 0.02 s, and there is a voltage ripple of 4 - 5 V after the converter under LADRC control stabilizes.

[0165] As Figure 10 shown is the output voltage waveform diagram of the improved anti-integral saturation NLADRC.

[0166] Among them, compared with LADRC, after improvement, due to the addition of a filtering function and an integral anti-windup strategy, the rapidity is slightly improved. Stable voltage output can be achieved in about 0.18 s, the overshoot is extremely small and can be ignored, and the output voltage ripple is also relatively small, being 2 - 3 V.

[0167] Experimental verification

[0168] As Figure 11 shown are the disturbance rejection comparison waveforms of different control strategies under low voltage. Among them, Fig. (a) is the disturbance rejection waveform under PI control. After suddenly applying a load, the output voltage drops slightly by 2 - 3 V and there is a certain amount of ripple. Fig. (b) is the voltage control strategy based on the improved anti-integral saturation NLADRC. After the disturbance appears, the voltage fluctuation cannot be captured, and the output voltage does not change before and after the disturbance. Since the test voltage is low and the load change is not obvious, but the effectiveness of the proposed control strategy can be verified by comparing the capture situation of the oscilloscope.

[0169] If the function is implemented in the form of a software functional unit and sold or used as an independent product, it can be stored in a computer-readable storage medium. Based on such an understanding, the technical solution of the present invention, in essence, or the part that contributes to the prior art or a part of this technical solution can be embodied in the form of a software product. This computer software product is stored in a storage medium and includes several instructions for causing a computer device (which can be a personal computer, a server, or a network device, etc.) to execute all or part of the steps of the methods in various embodiments of the present invention. And the aforementioned storage medium includes: USB flash drives, mobile hard disks, read-only memories (ROM, Read-Only Memory), random access memories (RAM, Random Access Memory), magnetic disks, or optical discs and other various media that can store program codes.

[0170] The logic and / or steps represented in the flowchart or otherwise described herein can, for example, be considered as a definable sequence of executable instructions for implementing logical functions, and can be embodied in any computer-readable medium for use by or in connection with an instruction execution system, apparatus, or device, such as a computer-based system, a system including a processor, or other systems that can fetch and execute instructions from the instruction execution system, apparatus, or device. As used in this specification, a "computer-readable medium" can be any device that can contain, store, communicate, propagate, or transport a program for use by or in connection with an instruction execution system, apparatus, or device.

[0171] More specific examples (non-exhaustive list) of computer-readable media include the following: an electrical connection part (electronic device) having one or more wirings, a portable computer diskette (magnetic device), a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or flash memory), an optical fiber device, and a portable compact disc read-only memory (CDROM). Additionally, the computer-readable medium can even be paper or other suitable media on which a program can be printed, because the program can be obtained electronically, for example, by optically scanning the paper or other media, followed by editing, interpretation, or other suitable processing as necessary, and then stored in a computer memory.

[0172] It should be understood that the various parts of the present invention can be implemented in hardware, software, firmware, or a combination thereof. In the above embodiments, multiple steps or methods can be implemented using software or firmware stored in a memory and executed by a suitable instruction execution system. For example, if implemented in hardware, as in another embodiment, any one or a combination of the following techniques well known in the art can be used: discrete logic circuits having logic gate circuits for implementing logical functions on data signals, application specific integrated circuits having appropriate combinational logic gate circuits, programmable gate arrays (PGAs), field programmable gate arrays (FPGAs), etc. It should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and not to limit them. Although the present invention has been described in detail with reference to the preferred embodiments, those of ordinary skill in the art should understand that the technical solutions of the present invention can be modified or equivalently replaced without departing from the spirit and scope of the technical solutions of the present invention, and they should all be covered by the scope of the claims of the present invention.

[0173] It should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention rather than to limit them. Although the present invention has been described in detail with reference to the preferred embodiments, those of ordinary skill in the art should understand that the technical solutions of the present invention can be modified or equivalently replaced without departing from the spirit and scope of the technical solutions of the present invention, and they should all be covered within the scope of the claims of the present invention.

Claims

1. A NLADRC voltage control method for a high-efficiency power converter for an electric vehicle, characterized in that: include, Establish the equivalent model of CLLLC resonant converter based on fundamental wave analysis method; The voltage gain is solved by using the FHA equivalent model of the CLLLC resonant converter and the linear circuit solution method. Establish an improved anti-integral saturation NLADRC voltage control strategy; The improved anti-integral saturation NLADRC voltage control strategy includes: Design the voltage controller based on LADRC; The voltage controller is designed based on the improved anti-integral saturation NLADRC; The design of the voltage controller based on the improved anti-integral saturation NLADRC includes: NLADRC is implemented on the basis of LADRC by adding a nonlinear filter function as shown below: Among them, e is the error; α is the nonlinear factor, which represents the nonlinearity of the filter function, 0≤α≤1; δ is the filter factor, which represents the width of the linear interval; The following function is used as the integral anti-saturation nonlinear filter function: Among them, e is the error, k e k is the proportionality coefficient e >0,0<χ<1,K lim are the upper and lower limits of integration, and ε is the maximum error.

2. The NLADRC voltage control method for a high-efficiency power converter for an electric vehicle according to claim 1, wherein: The CLLLC resonant converter equivalent model established based on the fundamental wave analysis method includes: The full-bridge equivalent replacement of the CLLLC resonant converter is performed, and the secondary side of the transformer is converted to the primary side to obtain the FHA model of the CLLLC resonant converter; where V AB_FHA It is the sine wave voltage equivalent to the voltage between the midpoints of the inverter bridge arm through FHA; L r1 and C r1 are the resonant inductance and capacitance of the primary and primary sides respectively; let n p is the transformer ratio, the resonant inductance of the secondary side converted to the primary side Resonant capacitor Req is the load obtained by using FHA to equalize the resistive load on the output side; Due to the switching of the switching devices in the inverter unit, the voltage between the midpoints of the bridge arms is actually V in and -V in The square wave voltage that jumps between is shown as follows: Among them, T s is the switching period, i.e. T s =1 / f s , where f s is the switching cycle; Performing Fourier transform on the above equation gives the following equation: Where n is the integer parameter of the Fourier series expansion; In the fundamental wave analysis method, it is believed that the energy of the input signal mainly comes from the fundamental wave component, and the remaining high-order harmonics are negligible; the fundamental wave component and its effective value are extracted as follows: Since the input of CLLLC needs to pass through the inverter unit, the current flowing through the resonant cavity is actually an alternating current. According to the idea of ​​fundamental wave analysis, it is equivalent to a sine wave as follows: I r1 is the transformer primary resonant current i r1 (t) effective value, Yes V AB_FHA (t) and i r1 (t) The phase difference between them, similarly, the fundamental component and effective value of the voltage between the midpoints CD of the two bridge arms of the rectifier unit are obtained: Where θ is the phase difference between the output side and the input side; The current in the secondary resonant cavity is expressed as: Among them, I r2 is the effective value of the transformer secondary resonant current ir2(t); The resonant cavity obtains the output current through the rectifier unit, so the output average current is: Among them, I o is the output current, P o is the output power, V o is the output voltage, R o is the output resistive load; From the above formula, we can see that V CD_FHA (t) and i r2 (t) are in phase, so Ro is equivalent to the equivalent output resistance Req of the FHA model:

3. The NLADRC voltage control method for a high-efficiency power converter for an electric vehicle as claimed in claim 2, characterized in that: The FHA equivalent model based on the CLLLC resonant converter uses a linear circuit solution method to solve the voltage gain, including: The voltage gain of the CLLLC resonant converter is expressed as: in, where ω s is the switching angular frequency; The CLLLC resonant converter is defined as follows: If x=y=1, the converter structure can achieve bidirectional symmetry, achieve bidirectional voltage gain and control consistency, and simplify parameter design and control; To further obtain the gain expression of the converter, it is necessary to take the normalized frequency f n =f s / f r =ω s / ω r , f s and ω s is the actual switching frequency and the corresponding angular frequency; characteristic impedance Quality factor Q = Z r / R eq , substituting the above definition into the formula, the gain expression of the converter can be obtained as:

4. The NLADRC voltage control method for a high-efficiency power converter for an electric vehicle as claimed in claim 3, characterized in that: The design of the voltage controller based on LADRC includes: The CLLLC resonant converter is reduced to a second-order model, and its transfer function structure is shown in the following equation: According to the above formula, let y be the output quantity, that is, the secondary side output, and u be the input quantity, that is, the primary side DC voltage input, we have: Among them, a0, a1, c0, c1, and b0 are the undetermined coefficients in LADRC, and a0, a1, c0, and c1 are related to the mathematical model of the converter being reduced to the second order; Choose the state variable x as Then the state variable contains the total disturbance, and the corresponding state equation is written as: Therefore, the third-order LESO is established as: Among them, β1, β2, β3 are observer parameters; z1, z2 are the observed values ​​of the system state variables, and z3 is the total disturbance observation of the controlled object; if the parameters are set appropriately, the correct state variable x can be estimated; To achieve the desired control, the LSEF is set to: Among them, k p , k d is the controller parameter; r is the given target value.

Images