Control method for power conversion system

Abstract

The invention provides a control method for a power conversion system, the power conversion system is used for inverting the voltage of a DC bus and then outputting the inverted voltage to a power grid, and the control method comprises the following steps: obtaining a current error value based on an inversion output current from the power conversion system and a current reference value; obtaining a resonance control output based on the current error value; obtaining a first duty ratio intermediate value based on the resonant control output, the gain of the inverter output current and the gain of the voltage difference between the positive side and the negative side of the direct current bus from the power conversion system; and obtaining a pulse width modulation wave duty ratio based on the network voltage and the first duty ratio intermediate value, wherein the pulse width modulation wave duty ratio is used for controlling the output of the power conversion system.

Description

Technical Field

[0001] The present invention relates to the field of control, and in particular to a resonant linear quadratic control system and a control method for a power conversion system. Background Art

[0002] The power conversion system (PCS), also known as the converter, can control the charging and discharging process of the battery, perform AC / DC conversion, and directly supply power to the grid-connected AC grid as a current source inverter CSI or an off-grid voltage source inverter VSI. The PCS consists of a DC / AC bidirectional converter, a control unit, etc.

[0003] The PCS controller receives background control instructions through communication, and controls the converter to charge or discharge the battery according to the sign and size of the power instruction, thereby adjusting the active power and reactive power of the power grid. The PCS controller communicates with the battery management system (BMS) through the CAN interface to obtain the status information of the battery pack, which can realize protective charging and discharging of the battery and ensure safe operation of the battery. Therefore, a control system and control method for PCS with low stability error and high performance is required. Summary of the invention

[0004] Based on the above problems in the prior art, the present invention provides a control method for a power conversion system, wherein the power conversion system is used to invert the voltage of a DC bus and output it to a power grid. The control method includes:

[0005] obtaining a current error value based on an inverter output current from the power conversion system and a current reference value;

[0006] obtaining a resonant control output based on the current error value;

[0007] Obtaining a first duty cycle intermediate value based on the resonance control output, the gain of the inverter output current, and the gain of the voltage difference between the positive and negative sides of the DC bus of the power conversion system;

[0008] A pulse width modulation wave duty cycle is obtained based on the grid voltage and the first duty cycle intermediate value, and the pulse width modulation wave duty cycle is used to control the output of the power conversion system.

[0009] In one embodiment, the control method further includes: multiplying different phases of the inverter output current from the power conversion system by corresponding gain coefficients to obtain the gain of the inverter output current.

[0010] In one embodiment, the control method further includes: multiplying the voltage difference between the positive and negative sides of the DC bus from the power conversion system by a corresponding gain coefficient to obtain a gain of the voltage difference between the positive and negative sides of the DC bus.

[0011] In one embodiment, the control method further comprises: multiplying the grid voltage input vector by The grid voltage gain is obtained, where B p is the duty cycle input matrix, B g is the grid voltage input matrix,

[0012]

[0013] Wherein, L is the inverter inductance of the power conversion system, V bus is the DC bus voltage.

[0014] In one embodiment, the step of obtaining a resonant control output based on the current error value further includes:

[0015] The resonant control output is obtained by multiplying the current error value by the transfer function of the resonant controller, where the transfer function is:

[0016]

[0017] ω o is the grid voltage angular frequency, k5 k6 are the gain coefficients of the resonant control output matrix, and s is the operator differential of the control transfer function.

[0018] In one embodiment, the step of obtaining a resonant control output based on the current error value further includes:

[0019] The current error value is compared with the input matrix B of the resonant control c The product of the state vector of the resonant control and the state matrix A of the resonant control c The product of is added and then integrated to obtain the state vector of the resonance control;

[0020] Multiply the state vector of the resonant control by the output matrix C of the resonant control c After that, the resonance control output is obtained;

[0021] in, C c =[k5 k6].

[0022] In one embodiment, the power grid includes three phases, and the control method further includes:

[0023] Based on the output current effective value of the inverter active and reactive control loop from the grid-connected inverter control, and the ratio of the output AC αβ axis voltage of the power conversion system to the AC standard sinusoidal voltage in the αβγ coordinate system, the current reference value of the αβ axis is obtained.

[0024] In one embodiment, the control method further includes:

[0025] Obtaining a voltage difference error value based on a bus voltage difference reference value and a voltage difference between the positive and negative sides of a DC bus from the power conversion system;

[0026] Based on the voltage difference error value, a current reference value of the γ-axis is obtained through proportional-integral control.

[0027] In one embodiment, the control method further includes:

[0028] A current error value is obtained based on the current reference value of the αβγ axis, the output AC capacitor current compensation value of the power conversion system, and the inverter output current from the power conversion system.

[0029] In one embodiment, the control method further includes:

[0030] Obtaining a current loop duty cycle based on the first duty cycle intermediate value and the inverse of the inverter modulation wave gain coefficient;

[0031] Obtaining a feedforward duty cycle based on the grid voltage and the DC bus voltage;

[0032] The pulse width modulation wave duty cycle is obtained based on the current loop duty cycle and the feedforward duty cycle.

[0033] The present invention also provides a control system for a power conversion system, wherein the power conversion system is used to invert the voltage of a DC bus and output it to a power grid, and the control system comprises:

[0034] A first subtractor, configured to receive an inverter output current and a current reference value from the power conversion system, and output a current error value;

[0035] a resonant controller, configured to receive the current error value and output a resonant control output;

[0036] a second subtractor, configured to receive the resonance control output, the gain of the inverter output current, and the gain of the voltage difference between the positive and negative sides of the DC bus from the power conversion system, and to subtract the gain of each phase of the inverter output current and the gain of the voltage difference between the positive and negative sides of the DC bus from each phase in the resonance control output to obtain a first duty cycle intermediate value;

[0037] The third subtractor is used to output a pulse width modulation wave duty cycle based on the grid voltage and the first duty cycle intermediate value, and the pulse width modulation wave duty cycle is used to control the inverter output current of the power conversion system.

[0038] In one embodiment, the control system further includes a first multiplier for receiving the inverter output current from the power conversion system, multiplying different phases thereof by corresponding gain coefficients, and outputting the gain of the inverter output current.

[0039] In one embodiment, the control system further includes a second multiplier for receiving the DC bus positive and negative voltage difference from the power conversion system, multiplying it by a corresponding gain coefficient, and outputting the gain of the DC bus positive and negative voltage difference.

[0040] In one embodiment, the control system further comprises a third multiplier for receiving a grid voltage input vector and multiplying it by The gain of the output grid voltage after B p is the duty cycle input matrix, B g is the grid voltage input matrix,

[0041]

[0042] Wherein, L is the inverter inductance of the power conversion system, V bus is the DC bus voltage.

[0043] In one embodiment, the transfer function of the resonant controller is:

[0044]

[0045] Among them, ω o is the grid voltage angular frequency, k5 k6 are the gain coefficients of the resonant control output matrix, and s is the operator differential of the control transfer function.

[0046] In one embodiment, the resonant controller comprises:

[0047] The fourth multiplier is used to receive the current error value and multiply it by the input matrix B of the resonance control c After output;

[0048] a first adder, configured to receive the output from the fourth multiplier and the gain of the integration of the output of the first adder, and output a differential of a resonance control state vector;

[0049] an integrator, configured to receive the differential of the resonance control state vector and output a state vector of the resonance control;

[0050] A fifth multiplier is used to receive the state vector of the resonance control and multiply it by the state matrix A of the resonance control c Then output to the first adder;

[0051] A sixth multiplier is used to receive the state vector of the resonance control and multiply it by the output matrix C of the resonance control. c Then, outputting the resonance control output;

[0052] in, C c =[k5 k6].

[0053] In one embodiment, the power grid includes three phases, and the control system further includes:

[0054] The seventh multiplier is used to receive the output current effective value of the inverter active and reactive control loop from the grid-connected inverter control, and the ratio of the output AC αβ axis voltage to the AC standard sinusoidal voltage of the power conversion system in the αβγ coordinate system, and output the current reference value of the αβ axis.

[0055] In one embodiment, the control system further comprises:

[0056] a fourth subtractor, configured to receive a bus voltage difference reference value and a voltage difference between the positive and negative sides of a DC bus from the power conversion system, and output a voltage difference error value;

[0057] The proportional-integral controller is used to receive the voltage difference error value and output a current reference value of the γ axis.

[0058] In one embodiment, the first subtractor is further used to receive the current reference value of the αβγ axis, add the output AC capacitor current compensation value of the power conversion system to it, subtract the inverter output current from the power conversion system, and output the current error value.

[0059] In one embodiment, the control system further comprises:

[0060] an eighth multiplier, configured to receive the first duty cycle intermediate value, multiply the intermediate value by the inverse of the inverter modulation wave gain coefficient, and output the current loop duty cycle;

[0061] Feedforward, for obtaining a feedforward duty cycle based on the grid voltage and the DC bus voltage;

[0062] The third subtractor is used to obtain the pulse width modulation wave duty cycle based on the current loop duty cycle and the feedforward duty cycle.

[0063] The control system and control method for a power conversion system of the present invention provide a novel resonant linear quadratic regulator, which includes a resonant controller and a linear quadratic regulator. The influence of the DC bus positive and negative voltage difference on the inverter output current is attenuated by including the DC bus positive and negative voltage difference in the state vector, and low stability error and high performance are obtained. BRIEF DESCRIPTION OF THE DRAWINGS

[0064] Figure 1 FIG. 4 is a circuit diagram of a power conversion system according to an embodiment of the present invention.

[0065] Figure 2 A schematic diagram of a control system for a power conversion system according to an embodiment of the present invention is shown.

[0066] Figure 3 A schematic diagram of a resonant controller according to an embodiment of the present invention is shown.

[0067] Figure 4 Shown include Figure 3 Schematic diagram of a control system of a resonant controller for a power conversion system.

[0068] Figure 5 A schematic diagram of a control system for a power conversion system according to an embodiment of the present invention is shown.

[0069] Fig. 6A The open-loop control margin Bode plot is shown.

[0070] Figure 6B A closed-loop Bode plot is shown.

[0071] Figure 7 A schematic diagram of a control system for a power conversion system according to another embodiment of the present invention is shown.

[0072] Fig. 8A The open-loop control margin Bode plot is shown.

[0073] Figure 8B The voltage balance closed loop step response is shown.

[0074] Fig. 9A Shows the L / r L ±50% current open loop margin.

[0075] Fig. 9B The current root locus curve at L±50% is shown.

[0076] Fig.10 A schematic diagram of a power conversion system with high impedance is shown.

[0077] Fig.11A schematic diagram of a high impedance inductor-capacitor-inductor (LCL) controlled object is shown.

[0078] Fig. 12A The root locus curve of a high impedance current system is shown.

[0079] Fig. 12B The step response of the current control system is shown.

[0080] Fig.13A The simulation output of the power conversion system is shown.

[0081] Fig. 13B The grid current harmonic values ​​of the power conversion system are shown.

[0082] Fig.14A The simulation output of the power conversion system is shown.

[0083] Fig. 14B The grid current harmonic values ​​of the power conversion system are shown. DETAILED DESCRIPTION

[0084] In order to make the purpose, technical solution and advantages of the present invention more clearly understood, the present invention will be further described in detail below through specific embodiments in conjunction with the accompanying drawings. It should be noted that the embodiments given in the present invention are only for illustration and do not limit the scope of protection of the present invention.

[0085] Figure 1 FIG. 2 shows a circuit diagram of a power conversion system according to an embodiment of the present invention. Figure 1 As shown, the power conversion system is used to receive direct current from the battery 101 and convert the direct current into three-phase alternating current to supply power to the power grid 102. The power grid 102 includes three phases, namely phase A, phase B and phase C. The power conversion system includes parallel X bridge arms and Y bridge arms, and the X bridge arms and the Y bridge arms are connected between the DC bus and the power grid 102. The parallel X bridge arms and Y bridge arms are used for power expansion. In actual applications, only one bridge arm can be used or more parallel bridge arms can be used. Among them, capacitors C1 and C2 are half-side capacitors of the DC bus, the voltage Vbus+ is the positive terminal voltage of the DC bus, and the voltage Vbus- is the negative terminal voltage of the DC bus. Phase A in the X bridge arm includes a switch tube Q XA1 ,Q XA2 ,Q XA3 ,Q XA4 , and the inductor L XA , by controlling the switch tube Q XA1 ,Q XA2 ,Q XA3 ,Q XA4 The duty cycle controls the inverter output current i of phase A in the X bridge arm XAThe principle of the B phase and the C phase in the X bridge arm is the same, which will not be repeated here. The Y bridge arm is similar to the X bridge arm, which will not be repeated here. The power conversion system also includes inverter AC capacitors C3, C4, and C5, and the capacitors C3, C4, and C5 are connected to point O (i.e., the ground terminal).

[0086] The A-phase grid-connected inverter current ia of the power grid 102 is the A-phase inverter output current i in the X bridge arm. XA The A-phase inverter output current i YA The sum of the B-phase grid-connected inverter current ib is the B-phase inverter output current i in the X bridge arm. XB The B-phase inverter output current i in the Y bridge arm YB The sum of the C-phase grid-connected inverter current ic is the C-phase inverter output current i in the X bridge arm. XC The C-phase inverter output current i in the Y bridge arm YC The grid voltage is usually fixed, so it is only necessary to control the grid inverter current ia, ib, ic to be a pure sine wave without harmonic interference. That is, it is necessary to control the inverter output current ia, ib, ic of the power conversion system.

[0087] In the present invention, for convenience, the inverter inductors L of the X and Y bridge arms are XA , L XB , L XC , L YA , L YB , L YC are the same, their inverter inductance value is L, and the internal resistance of the inductor is r L ; Make the DC bus half capacitors C1 and C2 the same, their capacitance values ​​are C bus ; Make the inverter AC capacitors C3, C4 and C5 the same, and their capacitance values ​​are all C ac Those skilled in the art should understand that the theoretical calculation and control method of the present invention are applicable when the inverter inductance values ​​of the X and Y bridge arms are different, the DC bus half capacitors are different, and the inverter AC capacitors are different.

[0088] In this power conversion system, the inverter output current ia, ib, ic is controlled by controlling the duty cycle of the switch tube in each bridge arm, and the voltage difference v between the positive and negative sides of the DC bus is controlled by controlling the duty cycle of the switch tube in each bridge arm. dff Make the positive and negative half axes of the DC bus symmetrical. dff When the value is 0, it means that the DC bus is balanced. Therefore, the state vector x p It can be expressed as:

[0089] x p =[i α ,i β ,iγ ,v dff ] T (1)

[0090] The inverter output current i α ,i β ,i γ They are the three-phase inverter output currents in the αβγ coordinate system. For the convenience of subsequent control, it is usually necessary to transform the ABC three-phase coordinate system to the αβγ coordinate system through Clark transformation, where the γ component is the common mode component or zero sequence component. That is, the inverter output currents ia, ib, and ic are transformed by Clark transformation to obtain the inverter output current i α ,i β ,i γ , current ia, ib, ic and current i α ,i β ,i γ Corresponding.

[0091] The output vector y is represented as:

[0092] y=[i α ,i β ,i γ ] T (2)

[0093] The duty cycle control vector u is expressed as:

[0094] u=[d α ,d β ,d γ ] T (3)

[0095] where d α ,d β ,d γ are the duty cycles of the pulse width modulation wave in the αβγ coordinate system respectively.

[0096] Grid voltage input vector e g It is expressed as:

[0097] e g =[e α ,e β ,e γ ] T (4)

[0098] where e α ,e β ,e γ are the three-phase grid voltages in the αβγ coordinate system respectively.

[0099] From this we can get the state equation:

[0100]

[0101] Where s is the operator differential of the control transfer function; is the integral transfer function; L is the inverter inductance, r L is the internal resistance of the inverter inductor, where the inductors of each branch are the same. bus is the DC bus voltage, that is, the half-side bus voltage, C bus is the half capacitance of the DC bus, that is Figure 1 The values ​​of capacitors C1 and C2. α S β S γ is the three-phase switching function of the αβγ coordinate system, whose value is 0 (off) or 1 (on), that is, S x =0 / 1; (x=α / β / γ). no is the voltage between the neutral line (N) and the midpoint (O) of the AC capacitor, that is, Figure 1 The voltage between the points N and O shown in FIG. 1 is the voltage between the neutral line of the power grid 102 and the ground of the power conversion system. bat is the battery current, Δi a is the sinusoidal positive and negative half-cycle current difference of the A-phase output current ia (which is the effective value or balanced value); Δi b is the difference between the positive and negative half-cycle sinusoidal current of the B-phase output current ib; Δi c It is the difference between the positive and negative half-cycle sinusoidal current of the C phase output current ic. γ is the output current of the γ axis.

[0102] The positive and negative DC bus voltage difference loop is the controlled object, and the voltage and loop are the controlled objects:

[0103]

[0104] Among them, Δi a =Δi b =Δi c =Δi.

[0105] In the case of a three-phase balanced power grid:

[0106]

[0107] v ao +v bo +v co =0 (8)

[0108]

[0109] Among them, v ao , v bo , v coare the three-phase voltages between the inverter three phases A, B, C and the midpoint O of the AC capacitor, v an , v bn , v cn They are the three-phase voltages between the inverter three phases A, B, C and the neutral line N.

[0110] When the three levels are equal, we get:

[0111]

[0112] Among them, d α , d β , d γ are the duty cycle of the pulse width modulation wave in the αβγ coordinate system, S α , S β , S γ They are the three-phase switching functions of the αβγ coordinate system respectively.

[0113] Based on the above equations, the mathematical model of the power conversion system is obtained:

[0114]

[0115] The mathematical model of the power conversion system includes the state equation x p ' and output equation y, where the state equation is the state vector x of the controlled object p The first-order differential of . p is the state matrix of the controlled object, B p is the duty cycle input matrix, B g is the grid voltage input matrix, C p The inverter current output matrix is:

[0116]

[0117]

[0118] Figure 2 A schematic diagram of a control system for a power conversion system according to an embodiment of the present invention is shown. As shown in the figure, the control system includes a subtractor 201, a current resonant controller 202, a multiplier 206, a multiplier 207, a subtractor 203, a multiplier 208, a subtractor 204, a multiplier 209, and a mathematical model 205.

[0119] The subtractor 201 is used to receive the output vector y from the power conversion system and the three-phase inverter current reference vector r in the αβγ coordinate system, where r = [r α ,r β ,r γ ] T , subtract the inverter current reference vector r from the output vector y to obtain the control system error value e.

[0120] The resonant controller 202 is used to receive the control system error value e from the subtractor 201 and output the resonant control output vector y c The resonant controller 202 includes a transfer function D r (s), output vector y c is the control system error value e and transfer function D r The output vector y is c =[y cα ,y cβ ,y cγ ] T .

[0121] The multiplier 206 is used to receive the inverter output current i in the state vector of the power conversion system. α ,i β ,i γ , multiply it by the gain coefficients k1, k2, k3 respectively, and output the inverter output current gain k1i α , k2i β , k3i γ In one embodiment, the multiplier 206 may be omitted, and the gain coefficients k1, k2, and k3 are all 1.

[0122] The multiplier 207 is used to receive the voltage v in the state vector from the power conversion system. dff , multiply it by the gain factor k4, and output voltage v dff Gain k4v dff In one embodiment, the multiplier 207 may be omitted, and the gain coefficient k4 is 1.

[0123] Subtractor 203, used to receive the resonant control output vector y from the resonant controller 202 c , the current i in the state vector of the power conversion system α ,i β ,i γ The gain of the power conversion system, and the voltage v in the state vector dff The gain will output the vector y c Subtract the current i from each phase α ,i β ,i γ gain, and subtract the voltage v dff The gain of the duty cycle is obtained by m , where u m =[d mα ,d mβ ,d mγ ] T .d mα , dmβ , d mγ It is expressed as:

[0124]

[0125] Although the subtractor is used as an example in the above embodiment, those skilled in the art should understand that, depending on the signs of the gain coefficients k1, k2, k3, k4 and different control methods, the subtractor 203 can also be implemented as an adder, or at least one subtraction operation in equation 14 can be implemented as an addition operation.

[0126] Multiplier 208, for receiving grid voltage input vector e g , multiply it by In one embodiment, the multiplier 208 can be omitted.

[0127] Subtractor 204, used to receive the duty cycle intermediate vector u from subtractor 203 m and the grid voltage input vector e from the multiplier 208 g The gain of is subtracted and the duty cycle control vector u is output. Therefore, in the control system of the present invention, the duty cycle control vector u is obtained based on the output current of the power conversion system, the voltage difference between the positive and negative sides of the DC bus, and the grid voltage.

[0128] Multiplier 209, for receiving the grid voltage input vector e g , multiply it by B g In one embodiment, the multiplier 209 can be omitted.

[0129] The mathematical model 205 includes the state equation and output equation of the power conversion system, which is the hardware modeling of the power conversion system and can simulate the output of the power conversion system. The mathematical model 205 is used to receive the duty cycle control vector u from the subtractor 204 and the optional grid voltage input vector e from the multiplier 209. g The gain and output state vector x p and the output vector y.

[0130] Figure 3 FIG. 3 is a schematic diagram of a resonant controller according to an embodiment of the present invention. The transfer function of the resonant controller 302 is:

[0131]

[0132] Among them, ω o is the grid voltage angular frequency, k5 k6 are the gain coefficients of the resonant control output matrix.

[0133] Define the state vector:

[0134]

[0135] Among them, x c is the resonant control state vector, x c1 is the resonant state vector current, x c2 is the equivalent differential of the resonant state vector current, y c is the resonance control output vector.

[0136] Converting it to the s domain is equivalent to:

[0137]

[0138] Among them, X c1 (s) is the resonant state vector current x c1 The transfer function, X c2 (s) is the equivalent differential of the resonant state vector current x c2 , E(s) is the transfer function of the control system error value e.

[0139] Therefore, we get:

[0140]

[0141]

[0142]

[0143] C c =[k5 k6]

[0144] Among them, x c ' is the differential of the resonant control state vector, A c is the state matrix of the resonant control, B c is the input matrix of the resonant control, C c is the output matrix of the resonant control.

[0145] Based on the above equations (15)-(18), again refer to Figure 3 The resonant controller 302 includes a multiplier 313, an adder 314, an integrator 315, a multiplier 316 and a multiplier 317. The multiplier 313 is used to receive the control system error value e and multiply it by the input matrix B of the resonant control. c The adder 314 is used to receive the output from the multiplier 313 and the output from the multiplier 317, and add them together to obtain the differential x of the resonance control state vector. c The integrator 315 is used to receive the differential x of the resonance control state vector from the adder 314. c ', integrate it and output the state vector x of the resonance controlc The multiplier 317 is used to receive the state vector x of the resonance control c , multiply it by the state matrix A of the resonant control c The multiplier 316 is used to receive the state vector x of the resonance control from the integrator 315. c , multiply it by the output matrix C of the resonant control c After that, the resonant control output vector y is output c .

[0146] Figure 4 Shown include Figure 3 Schematic diagram of the control system of the resonant controller used in the power conversion system. Figure 2-Figure 4 As shown, it is a resonant linear quadratic regulator (LQR) control system.

[0147] Create the augmented matrix state vector:

[0148] x=[x p ,x c ] T

[0149] u=-Kx

[0150]

[0151] Among them, x p is the state vector of the controlled object, x c is the state vector of the resonant control, x is x p With x c The augmented state vector composed of , u is the duty cycle control vector, K is the feedback gain matrix of the augmented state vector, and r is the inverter current reference vector.

[0152] therefore,

[0153]

[0154] Among them, A is the state matrix of the augmented vector, B is the duty cycle input matrix of the augmented vector, and B r Enter the matrix for the given value of the augmented vector, B t is the grid voltage input matrix of the augmented vector, C a is the inductor current output matrix of the augmented vector.

[0155] Substituting the inversion parameters gives:

[0156]

[0157]

[0158] Optimal calculation of linear quadratic regulator:

[0159] Since the inverter current reference vector r and the grid voltage input vector e in equation (20) g As an uncontrollable quantity, it can be ignored and the linear state equation can be written as:

[0160]

[0161] The performance objective function J of quadratic optimal control is:

[0162]

[0163] Among them, Q is the state performance weight matrix of optimal control, and R is the control energy weight matrix of optimal control.

[0164] The LQR Riccati matrix equation is:

[0165] PA+A T P-PBR -1 B T P+Q=0 (24)

[0166] Where P is the solution matrix of the Riccati matrix equation, A is the state matrix of the augmented vector, and A T is the transposed matrix of the state matrix A, B is the duty cycle input matrix of the augmented vector, B T is the transposed matrix of the duty cycle input matrix B, R -1 is the inverse matrix of the control energy weight matrix R.

[0167] Then the optimal control can be obtained:

[0168] u * =-R -1 B T Px=-Kx

[0169]

[0170] Among them, u * is the input matrix under the optimal control state, J * is the performance target value under the optimal control state, x is the augmented state vector, x T is the transposed vector of the augmented state vector x, x(0) is the zero state of the augmented state vector, x T (0) is the transposed matrix of the zero state of the augmented state vector.

[0171] Optimal parameter adjustment simulation / experiment:

[0172] The control state performance weight matrix Q is:

[0173] Q=diag(q1,q2,q3,q4,q5,q6) (26)

[0174] Among them, diag is a diagonal matrix, q1 q2…q6 are the weight coefficients of the state performance weight matrix, among which q1 q2q3 are the performance weight coefficients of the three-phase inverter output current, q4 is the performance weight coefficient of the DC bus differential loop, q5 is the current error performance weight coefficient of the resonant controller, and q6 is the current error differential performance weight coefficient of the resonant controller.

[0175] The weights q1 to q3 are selected so that the three-phase inverter output current i α ,i β ,i γ Sine fast tracking with large bandwidth:

[0176] e≤1%

[0177] q1~q3=q∈(0~10 -4 ) (27)

[0178] Select v dff To attenuate different bus influences:

[0179] q4=0 (28)

[0180] The weight coefficients q5 / q6 of the resonant controller will affect the current sinusoidal 50 / 60Hz tracking, so q5 and q6 may be large. In combination with q1 to q3, sufficient bandwidth and control margin are obtained:

[0181]

[0182] The control energy weight matrix R of the optimal control is selected as:

[0183] R=diag(1,1,1) (30)

[0184] Table 1 shows the system parameter values:

[0185] Table 1

[0186]

[0187] Figure 5 A schematic diagram of a control system for a power conversion system according to an embodiment of the present invention is shown. Figure 5 The control system and Figure 2 The control system in is equivalent to that in the example above. In order to analyze the stability of the system, Figure 2 The control system in is transformed into the s-domain to obtain Figure 5 Transfer function based control systems in .

[0188] like Figure 5 As shown, the control system includes a subtractor 501, a current resonant controller 502, a multiplier 506, a multiplier 507, a subtractor 503, an adder 504, a multiplier 508, and a mathematical model 505.

[0189] The subtractor 501 is used to receive the inverter output current i from the power conversion system 505. αβγ , and the three-phase inverter current reference value r in the αβγ coordinate system αβγ , the three-phase inverter current reference value r αβγ With the inverter output current i αβγ After subtraction, the control system error value e is obtained.

[0190] The resonant controller 502 is used to receive the control system error value e from the subtractor 501 and output the resonant control output y cαβγ The resonant controller 502 includes a transfer function D r (s), resonant control output y cαβγ is the control system error value e and transfer function D r The product of (s).

[0191] The multiplier 506 is used to receive the inverter output current i from the power conversion system 505. αβγ (i α ,i β ,i γ ), multiply its different phases by the corresponding gain coefficients k1, k2, k3, and output the inverter output current i α ,i β ,i γ The gain k1i α , k2i β , k3i γ In one embodiment, the multiplier 506 may be omitted, and the gain coefficients k1, k2, and k3 are 1.

[0192] The multiplier 507 is used to receive the voltage v from the power conversion system. dff , multiply it by the gain factor k4, and output voltage v dff Gain k4v dff In one embodiment, the multiplier 507 may be omitted, and the gain coefficient k4 is 1.

[0193] Subtractor 503, used to receive the resonance control output y from the resonance controller 502 cαβγ , inverter output current i α ,i β ,i γ The gain, and voltage vdff The gain of the resonant control output y cαβγ Subtract the current i from each phase α ,i β ,i γ gain, and subtract the voltage v dff The gain of the first duty cycle intermediate value d mαβγ1 .

[0194] Adder 504, used to receive the grid voltage e αβγ , and the first duty cycle intermediate value d mαβγ1 , add them together and output the second duty cycle intermediate value d mαβγ2 .

[0195] Multiplier 508, for receiving the second duty cycle intermediate value d mαβγ2 , multiply it by the inverter modulation wave gain coefficient K pwm The reciprocal of Output pulse width modulation wave duty cycle d αβγ The inverter modulation wave gain coefficient K pwm The derivation of is known in the art and will not be described in detail here.

[0196] The power conversion system 505 is used to receive the pulse width modulation wave duty cycle d αβγ , and the grid voltage e αβγ , and output the DC bus positive and negative voltage difference v dff And the inverter output current i αβγ .

[0197] In one embodiment, the feedback gain of each phase of the αβγ coordinate system is:

[0198] K α =(k1,k2,k3,k4)=(0.201905,0,0,0.0001,-1125.05,-9936.76)

[0199] K β =(k1,k2,k3,k4)=(0,0.201905,0,0.0001,-1125.05,-9936.76)

[0200] K γ =(k1,k2,k3,k4)=(0,0,0.201905,0.0001,-1125.05,-9936.76) (31)

[0201] Among them, K α is the α phase state feedback gain matrix in the αβγ coordinate system, K β is the β-phase state feedback gain matrix in the αβγ coordinate system, Kγ is the γ phase state feedback gain matrix in the αβγ coordinate system.

[0202] The transfer function is:

[0203]

[0204]

[0205] Among them, G ivm (s) is the transfer function from inverter output voltage to inverter output current, i.e., the current loop controlled object, G vi (s) is the transfer function from the inverter output current to the voltage difference between the positive and negative sides of the DC bus, that is, the voltage difference loop controlled object.

[0206] System open loop / closed loop:

[0207]

[0208]

[0209] Among them, G i_ol (s) is the open-loop system transfer function of current control, G i_cl (s) is the closed-loop system transfer function of current control. Therefore, the amplitude-frequency curve of the final system is obtained, such as Figure 6A-6B shown. Fig. 6A The open-loop control margin Bode plot is shown, where the phase margin PM is 45.5deg and the gain margin GM is 13.2dB, both of which are much greater than 0. Figure 6B A closed-loop Bode plot is shown.

[0210] Figure 7 The schematic diagram of the control system for a power conversion system according to another embodiment of the present invention is shown. The entire control system uses two serial control loops, an inner current loop and an outer voltage loop. The inner current loop is an inverter current resonant linear quadratic control, and the outer voltage loop is a DC bus positive and negative voltage difference v dff The voltage difference loop proportional integral (PI) control. The external voltage loop helps to eliminate the imbalance of positive and negative DC buses, ensure inverter modulation balance, and stable grid-connected current.

[0211] like Figure 7 As shown, the PQ loop 709 is the inverter active and reactive power control loop of the grid-connected inverter control, which is used to output the effective value of the current I rms The multiplier 710 is used to receive the effective value of the current I rms , and the AC αβ axis voltage v of the power conversion system ac_αβ (t) and the standard AC sinusoidal voltage v sin_αβ (t) ratio v ac_αβ (t):vsin_αβ (t), and then multiply them to output the current reference value r of the αβ axis αβ .

[0212] The outer voltage loop includes a subtractor 711 and a proportional integral (PI) controller 712 for calculating the DC bus voltage difference v dff The bus voltage difference reference value generates the γ-axis current reference value r γ The subtractor 711 is used to receive the bus voltage difference reference value (eg, 0V) and the DC bus positive and negative voltage difference v from the power conversion system. dff , and outputs a voltage error value after subtracting it. The PI controller 712 is used to receive the voltage error value from the subtractor 711 and output a current reference value r of the γ axis. γ The current reference value r of the αβγ coordinate system is obtained αβγ . The transfer function of the PI controller is:

[0213]

[0214] Among them, D vdff (s) is the voltage difference loop controller, K p is the proportional gain coefficient, T i is the integration time constant.

[0215] The inner current loop includes a subtractor 701, a resonant controller 702, a subtractor 703, a multiplier 704, a feedforward 713 and a subtractor 708, which are used to generate a pulse width modulation wave duty cycle d αβγ .

[0216] The subtractor 701 is used to receive the current reference value r of the αβγ axis. αβγ , minus the inverter output current i of the power conversion system αβγ , plus the AC capacitor current compensation value i cap_comp , and output the control system error value e.

[0217] The resonant controller 702 is used to receive the control system error value e from the subtractor 701 and output the resonant control output y cαβγ The resonant controller 702 includes a transfer function D r (s), resonant control output y cαβγ is the control system error value e and transfer function D r The product of (s).

[0218] The multiplier 706 is used to receive the inverter output current i from the power conversion system. αβγ (i α ,i β ,i γ), multiply them by the gain coefficients k1, k2, k3 respectively, and output the inverter output current gain k1i α , k2i β , k3i γ In one embodiment, the multiplier 706 may be omitted, and the gain coefficients k1, k2, and k3 are 1.

[0219] The multiplier 707 is used to receive the voltage v from the power conversion system. dff , multiply it by the gain factor k4, and output voltage v dff Gain k4v dff In one embodiment, the multiplier 207 may be omitted, and the gain coefficient k4 is 1.

[0220] Subtractor 703, used to receive the resonance control output y from the resonance controller 702 cαβγ , inverter output current i α ,i β ,i γ The gain, and voltage v dff The gain of the resonant control output y cαβγ Each phase in the subtracts the inverter output current i αβγ gain, and subtract the voltage v dff The gain of the first duty cycle intermediate value d mαβγ1 .

[0221] Multiplier 704, used to receive the first duty cycle intermediate value d mαβγ1 , multiply it by the inverter modulation wave gain coefficient K pwm The reciprocal of The output current loop duty cycle d iloop .

[0222] Feedforward 713, for inverter output voltage V based on power conversion system inv , and the DC bus voltage v bus , and get the feedforward duty cycle d vff The feedforward system is a known technology in the art and will not be described in detail here. In one embodiment, the inverter output voltage V inv Equal to the grid voltage.

[0223] Subtractor 708, used to receive the current loop duty cycle d iloop , minus the feedforward duty cycle d from the feedforward 713 vff Output pulse width modulation wave duty cycle d αβγ .

[0224] The power conversion system 705 is used to receive the pulse width modulation wave duty cycle d αβγ , and output the DC bus positive and negative voltage difference vdff And the inverter output current i αβγ .

[0225] The transfer function is:

[0226]

[0227]

[0228] Among them, G id (s) is the transfer function from duty cycle to inverter output current, G vi_dff (s) is the transfer function from the inverter output current to the voltage difference between the positive and negative sides of the DC bus, that is, the voltage difference loop controlled object.

[0229] Fig. 8A The open-loop control margin Bode plot is shown, where the phase margin PM is 59.1deg. Figure 8B The voltage balance closed-loop step response curve is shown, and the system eventually converges.

[0230] Robust high impedance stability analysis:

[0231] Parameter changes of key hardware components, such as L / r L ±50%, i.e., inductance L and inductor resistance r L Deviation range, nominal value changes -50% to +50%; C bus ±15%, that is, the deviation range of the DC bus half capacitance, the nominal value changes from -15% to +15%; C ac ±15%, that is, the deviation range of the inverter AC capacitor, the nominal value changes from -15% to +15%. Fig. 9A Shows the L / r L ±50% current open loop margin. Fig. 9B The current root locus curve at L±50% is shown. Fig. 9A and Fig. 9B It shows that the system has good margin and stability.

[0232] Fig.10 A schematic diagram of a power conversion system with high impedance is shown. High impedance is defined as Zg = 15% * Zs, where Z g It is the general term for high impedance of power grid input, Z s It is the equivalent output impedance of the power conversion system at rated power.

[0233] Input impedance inductance ratio K g is the ratio of the inductive value to the resistive value of the input high impedance, that is:

[0234]

[0235] Among them, Lg The high impedance inductor of the grid input, r g A high impedance resistor is input to the grid.

[0236] Fig.11 The schematic diagram of the high impedance LCL controlled object is shown in Figure 1. Under high impedance of the grid, the transfer function of the modulation duty cycle to the inductor current under high impedance input is:

[0237]

[0238] Considering sL+r L , and the grid resistance-inductance ratio sL g +r g , the transfer function from the modulation duty cycle to the inductor current under high impedance input is:

[0239]

[0240] Then the open-loop system transfer function of the inverter output current control under high impedance input is:

[0241]

[0242] Among them, G id (s) or G id_lcl (s) is the transfer function from duty cycle to inverter output current in high impedance state, i.e., the current controlled object, G vi (s) is the transfer function from the inverter output current to the DC bus voltage difference in the high impedance state, that is, the voltage difference loop controlled object.

[0243] Fig. 12A The high impedance current system root locus curve is shown, where the nominal value of the grid input high impedance value Zg varies by +1% to 50%. Fig. 12B The step response of the current control system is shown, where the grid input has a high impedance value Z g The nominal value change is 15%. As can be seen from 12A-12B, the nominal value change of the grid input high impedance value Zg is stable and reliable when it is below 30%.

[0244] Fig.13A The simulation output of the power conversion system is shown, grid-connected 120kW, 3-phase, without N system, from top to bottom, inverter A / B / C three-phase voltage, inverter output current of inverter bridge arm X, inverter grid A / B / C three-phase current, DC bus positive and negative voltage. Fig. 13B The harmonic value of the grid current of the power conversion system is shown, wherein the harmonic value THD_Io=2.26%. Therefore, the control system for the power conversion system of the present invention has high performance and high reliability.

[0245] Fig.14AThe simulation output of the power conversion system is shown, grid-connected 120kW, 3-phase, without N system, from top to bottom, DC bus positive voltage, DC source (rectifier) ​​input A phase current, inverter bridge arm X A phase inverter output current. Fig. 14B The harmonic value of the grid current of the power conversion system is shown, wherein the harmonic value THD_Io=2.4%. Therefore, the control system for the power conversion system of the present invention has high performance and high reliability.

[0246] Although the power conversion system and the power grid in the above embodiments include three phases, the present invention is not limited thereto, and the control system and control method of the present invention can also be applied to a power conversion system and a power grid including only one phase or more phases.

[0247] For ease of understanding, the above embodiments are described in the form of hardware systems, but those skilled in the art should understand that the adders, subtractors, multipliers, resonant controllers, proportional-integral controllers, etc. in the above embodiments can all be implemented as software and / or hardware.

[0248] Based on the above embodiments, the present invention further provides a control method for a power conversion system, wherein the power conversion system is used to invert the voltage of a DC bus and output it to a power grid, and the control method includes:

[0249] obtaining a current error value based on an inverter output current from the power conversion system and a current reference value;

[0250] obtaining a resonant control output based on the current error value;

[0251] Obtaining a first duty cycle intermediate value based on the resonance control output, the gain of the inverter output current, and the gain of the voltage difference between the positive and negative sides of the DC bus of the power conversion system;

[0252] A pulse width modulation wave duty cycle is obtained based on the grid voltage and the first duty cycle intermediate value, and the pulse width modulation wave duty cycle is used to control the output of the power conversion system.

[0253] The control system of the present invention provides a novel resonant linear quadratic regulator D r_lqr (s), which includes a resonant compensator D r (s), and the enhanced linear quadratic regulator D lqr (s). The state vector contains the voltage difference v between the positive and negative sides of the DC bus. dff and through the voltage difference v dff To attenuate its effect on the inverter output current i abcWhen used in grid-connected inverters, based on algorithm analysis, the optimal controller for inverter sinusoidal 50 / 60 Hz can be obtained, and low stability error and high performance are obtained under simulation / experiment.

[0254] Based on the coordinate system αβγ, the bus voltage balance control G vdff (s) System and resonant linear quadratic regulator D r_lqr (s) together as a γ loop, constitute the control system of the power conversion system to ensure the stable balance of the bus voltage and to attenuate the lower frequency harmonics of the current generated by the inverter.

[0255] The resonance control system D of the present invention r_lqr (s), when used in a high-impedance grid scenario, basic stability analysis and simulation show that the control system of the present invention operates stably with sufficient margin.

[0256] The control system of the present invention is based on the grid-connected interleaved inverter module, which helps to obtain good grid current performance (low harmonics and power factor -1 to 1). Finally, simulation and experiments show that the control system of the present invention works well in terms of stability and dynamics, and obtains good performance.

[0257] The resonant linear quadratic regulator of the present invention can also be used in rectification systems, and dual / single-directional converters.

[0258] Although the present invention has been described through preferred embodiments, the present invention is not limited to the embodiments described herein but includes various changes and modifications that may be made without departing from the scope of the present invention.

Claims

1. A control method for a power conversion system, wherein the power conversion system is used to invert the voltage of a DC bus and output it to a power grid, the control method comprising: obtaining a current error value based on an inverter output current from the power conversion system and a current reference value; obtaining a resonant control output based on the current error value; Obtaining a first duty cycle intermediate value based on the resonance control output, the gain of the inverter output current, and the gain of the voltage difference between the positive and negative sides of the DC bus of the power conversion system; A pulse width modulation wave duty cycle is obtained based on the grid voltage and the first duty cycle intermediate value, and the pulse width modulation wave duty cycle is used to control the output of the power conversion system.

2. The control method for a power conversion system according to claim 1, wherein: The control method further includes: multiplying different phases of the inverter output current from the power conversion system by corresponding gain coefficients to obtain the gain of the inverter output current.

3. The control method for a power conversion system according to claim 1, wherein: The control method further includes: multiplying the voltage difference between the positive and negative sides of the DC bus from the power conversion system by a corresponding gain coefficient to obtain the gain of the voltage difference between the positive and negative sides of the DC bus.

4. The control method for a power conversion system according to claim 1, wherein: The control method further comprises: multiplying the grid voltage input vector by The grid voltage gain is obtained, where B p is the duty cycle input matrix, B g is the grid voltage input matrix, Wherein, L is the inverter inductance of the power conversion system, V bus is the DC bus voltage.

5. The control method for a power conversion system according to claim 1, wherein: The step of obtaining a resonant control output based on the current error value further includes: The resonant control output is obtained by multiplying the current error value by the transfer function of the resonant controller, where the transfer function is: ω o is the grid voltage angular frequency, k5 k6 are the gain coefficients of the resonant control output matrix, and s is the operator differential of the control transfer function.

6. The control method for a power conversion system according to claim 5, wherein: The step of obtaining a resonant control output based on the current error value further includes: The current error value is compared with the input matrix B of the resonant control c The product of the state vector of the resonant control and the state matrix A of the resonant control c The product of is added and then integrated to obtain the state vector of the resonance control; Multiply the state vector of the resonant control by the output matrix C of the resonant control c After that, the resonance control output is obtained; in, C c =[k5 k6].

7. The control method for a power conversion system according to claim 1, wherein: The power grid includes three phases, and the control method further includes: Based on the output current effective value of the inverter active and reactive control loop from the grid-connected inverter control, and the ratio of the output AC αβ axis voltage of the power conversion system to the AC standard sinusoidal voltage in the αβγ coordinate system, the current reference value of the αβ axis is obtained.

8. The control method for a power conversion system according to claim 7, wherein: The control method further comprises: Obtaining a voltage difference error value based on a bus voltage difference reference value and a voltage difference between the positive and negative sides of a DC bus from the power conversion system; Based on the voltage difference error value, a current reference value of the γ-axis is obtained through proportional-integral control.

9. The control method for a power conversion system according to claim 8, wherein: The control method further comprises: A current error value is obtained based on the current reference value of the αβγ axis, the output AC capacitor current compensation value of the power conversion system, and the inverter output current from the power conversion system.

10. The control method for a power conversion system according to any one of claims 1 to 9, wherein: The control method further comprises: Obtaining a current loop duty cycle based on the first duty cycle intermediate value and the inverse of the inverter modulation wave gain coefficient; Obtaining a feedforward duty cycle based on the grid voltage and the DC bus voltage; The pulse width modulation wave duty cycle is obtained based on the current loop duty cycle and the feedforward duty cycle.

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