Bipolar voltage self-balancing converter and control method

Abstract

The invention provides a bipolar voltage self-balancing converter and a control method. The bipolar voltage self-balancing converter comprises a first port, a second port, a first capacitor, a first output capacitor, a second output capacitor, a first load resistor, a second load resistor, a coupling inductor, a full-bridge circuit and a half-bridge circuit, only one coupling inductor is used for transmitting energy at the primary side and the secondary side, so that the number of magnetic elements is reduced; according to the provided control method, linear fitting processing is carried out on the intra-bridge phase shift duty ratio of the full-bridge circuit through voltage gain and the inter-bridge phase shift duty ratios of the full-bridge circuit and the half-bridge circuit, and then the bipolar voltage self-balancing converter is controlled by adjusting the first duty ratio after linear fitting. Under the unbalanced load condition, the converter achieves zero-voltage switching in the full-voltage and full-load range, and therefore the problems that a bipolar voltage balance output converter is difficult to achieve soft switching in the full-load range, the number of devices is large, and control is complex are solved.

Description

Technical Field

[0001] The present invention relates to the technical field of switching power supplies, and in particular to a bipolar voltage self-balancing converter and a control method thereof. Background Art

[0002] In recent years, DC distribution systems do not need to synchronize phase and frequency with the power grid, and do not require complex circuits and operating conditions. Therefore, they are recognized as the core driving force of future distributed power generation. They can provide high-quality power conversion and are ideal for renewable energy such as photovoltaics and fuel cells.

[0003] The DC power distribution system can be divided into unipolar DC system and bipolar DC system according to the number of voltage levels provided by the DC bus to the load. Although the unipolar DC system has been widely used in the past, it is increasingly difficult to meet the requirements of many emerging applications in terms of efficiency and reliability. In order to solve the limitations of the unipolar DC system, the bipolar DC system came into being. In the bipolar DC system, the DC bus is configured in a three-wire configuration, which can provide three DC voltage levels and flexibly interface with DC-DC converters with various voltage ranges. This architecture provides higher flexibility and reliability, can better meet the needs of various applications, and also makes it a more effective choice to meet the needs of modern efficient and reliable power supply; however, the imbalance of load resistance, line resistance and renewable energy distribution will cause the neutral point voltage of the bipolar DC system to be unbalanced. This imbalance will affect the balance of the bipolar voltage, resulting in increased power loss and triggering line overcurrent protection.

[0004] In order to solve the problem of heavy voltage imbalance in bipolar DC systems, the prior art has proposed some voltage balancer topologies, which are derived from the basic non-isolated two-port converter; however, due to the lack of additional control freedom in these topologies, it is difficult to achieve precise control of the bipolar bus voltage; some researchers have summarized the bipolar voltage balancing schemes of isolated and non-isolated converters in detail from different perspectives; among them, the DC-DC converter with isolated bipolar output is usually configured by two-stage power conversion, and the DAB converter is cascaded with the buck-boost balancer to convert the input voltage into the required bipolar DC output voltage. However, such a two-stage configuration requires many devices, which reduces its power density and conversion efficiency and increases the equipment cost. In order to solve the shortcomings caused by the existence of the voltage balancer, researchers have proposed to integrate the isolation converter and the voltage balancer into one by using switch multiplexing or sharing the bridge arm branch. In the traditional DAHB converter, voltage balance can be achieved by adding an inductor and a capacitor, but it is only a voltage balancer based on a buck converter and cannot achieve flexible buck-boost regulation; there are also documents proposing a converter whose voltage balance operation is achieved by relying on two additional inductors, which can fully obtain a balanced bipolar voltage level under the entire unbalanced load state, thereby working under zero voltage switching, but the additional inductor used for voltage balance usually needs to have a higher inductance, so the volume is larger and the cost is increased. There are also documents showing that the voltage balance operation can be integrated into the DAB converter without a balancing inductor, and a single-stage voltage balancer with high-frequency current isolation based on the TL-DAB converter without additional circuits is proposed, but it uses more than twice the number of switch tubes of the traditional DAB converter, the circuit structure is very complex, and the control is also cumbersome; there is also a DAB-based bipolar voltage balance converter that has excellent performance in bipolar DC distributed systems. However, the introduction of flux balance control makes the control more complicated. Summary of the invention

[0005] The present invention provides a bipolar voltage self-balancing converter and a control method, which aims to solve the problem that the bipolar voltage balanced output converter is difficult to realize full load range soft switching and has a large number of components and complex control.

[0006] In order to achieve the above object, the present invention provides a bipolar voltage self-balancing converter, comprising:

[0007] A first port, a second port, a first capacitor, a first output capacitor, a second output capacitor, a first load resistor, a second load resistor, a coupled inductor, a full-bridge circuit, and a half-bridge circuit;

[0008] The positive electrode of the first port is respectively connected to the first end of the first capacitor and the first input end of the full-bridge circuit;

[0009] The negative electrode of the first port is respectively connected to the second end of the first capacitor and the second input end of the full-bridge circuit;

[0010] The first end of the coupled inductor is connected to the midpoint of the first bridge arm in the full-bridge circuit;

[0011] The second end of the coupled inductor is connected to the midpoint of the second bridge arm in the full-bridge circuit;

[0012] The third end of the coupled inductor is connected to the midpoint of the bridge arm in the half-bridge circuit;

[0013] The fourth end of the coupled inductor is respectively connected to the second end of the first output capacitor, the first end of the second output capacitor, the second end of the first load resistor, and the first end of the second load resistor;

[0014] The positive electrode of the second port is respectively connected to the first end of the first load resistor, the first end of the first output capacitor, and the first output end of the half-bridge circuit;

[0015] The negative electrode of the second port is respectively connected to the second end of the second load resistor, the second end of the second output capacitor, and the second output end of the half-bridge circuit.

[0016] To further explain, the coupled inductor is equivalent to the leakage inductance, magnetizing inductance and ideal transformer;

[0017] A first end of the leakage inductor is connected to a midpoint of a first bridge arm in the full-bridge circuit, and a second end of the leakage inductor is connected to a first end of an ideal transformer;

[0018] The second end of the ideal transformer is connected to the midpoint of the second bridge arm in the full-bridge circuit;

[0019] The third end of the ideal transformer is connected to the first end of the excitation inductor and the midpoint of the bridge arm in the half-bridge circuit respectively;

[0020] The fourth end of the ideal transformer is respectively connected to the second end of the excitation inductor, the second end of the first output capacitor, the first end of the second output capacitor, the second end of the first load resistor, and the first end of the second load resistor.

[0021] Further, the full-bridge circuit includes a first switch tube, a second switch tube, a third switch tube, and a fourth switch tube;

[0022] The drain of the first switch tube is respectively connected to the first end of the first capacitor, the positive electrode of the first port, and the drain of the third switch tube;

[0023] The source of the first switch tube is connected to the drain of the second switch tube to form a first bridge arm of the full-bridge circuit, and the midpoint of the first bridge arm is connected to the first end of the leakage inductance;

[0024] The source of the third switch tube is connected to the drain of the fourth switch tube to form a second bridge arm of the full-bridge circuit, and the midpoint of the second bridge arm is connected to the second end of the ideal transformer;

[0025] The source of the fourth switch tube is connected to the source of the second switch tube, the second end of the first capacitor, and the negative electrode of the first port respectively.

[0026] Further, the half-bridge circuit includes a fifth switch tube and a sixth switch tube;

[0027] The source of the fifth switch tube is connected to the drain of the sixth switch tube to form a bridge arm of the half-bridge circuit, and the midpoint of the bridge arm is connected to the third end of the ideal transformer and the first end of the excitation inductor;

[0028] The drain of the fifth switch tube is connected to the first end of the first output capacitor, the first end of the first load resistor, and the positive electrode of the second port;

[0029] The source of the sixth switch tube is connected to the second end of the second output capacitor, the second end of the second load resistor, and the negative electrode of the second port.

[0030] Specifically, the first port is a power input port, and the second port is a load output port.

[0031] The present invention also provides a control method for a bipolar voltage self-balancing converter, which is applied to the bipolar voltage self-balancing converter. The control method includes:

[0032] Step 1, collecting input voltage sampling values, output voltage sampling values, and output power of the bipolar voltage self-balancing converter;

[0033] Step 2, obtaining a voltage gain based on the input voltage sampling value and the output voltage sampling value;

[0034] Step 3, setting the intra-bridge phase shift duty ratio of the full-bridge circuit to a first duty ratio, the inter-bridge phase shift duty ratio of the full-bridge circuit and the half-bridge circuit to a second duty ratio, and dividing the bipolar voltage self-balancing converter into two operating conditions according to the positional relationship between the first duty ratio and the second duty ratio;

[0035] Step 4, calculating the first duty cycle under the two working conditions by using the output power, the second duty cycle, and the voltage gain, and performing linear fitting processing on the first duty cycle under the two working conditions according to the second duty cycle between the starting point and the end point of each working condition to obtain the first duty cycle after linear fitting under the two working conditions;

[0036] Step 5, controlling the bipolar voltage self-balancing converter under the two working conditions by adjusting the first duty cycle after linear fitting under the two working conditions.

[0037] Furthermore, the voltage gain k is calculated as:

[0038]

[0039] Where n represents the transformation ratio of the ideal transformer, V in Indicates the input voltage sampling value, V o Indicates the output voltage sampling value.

[0040] Specifically, the expression for calculating the first duty cycle under the two working conditions by using the output power, the second duty cycle, and the voltage gain is:

[0041]

[0042] in, represents the output power of the bipolar voltage self-balancing converter under working condition 1, represents the output power of the bipolar voltage self-balancing converter under working condition 2, D 11 Indicates the first duty cycle under working condition 1, D 12 represents the first duty cycle under working condition 2, represents the second duty cycle under working condition 1, Indicates the second duty cycle under working condition 2.

[0043] Specifically, a linear fitting process is performed on the first duty cycle under the two working conditions according to the second duty cycle between the starting point and the end point of each working condition, and the expression of the first duty cycle after linear fitting under the two working conditions is obtained as follows:

[0044]

[0045] Among them, D 11_minLINE represents the first duty cycle after linear fitting under working condition 1, D 12_minLINE Represents the second duty cycle after linear fitting under working condition 2.

[0046] The above scheme of the present invention has the following beneficial effects:

[0047] The converter provided by the present invention comprises a first port, a second port, a first capacitor, a first output capacitor, a second output capacitor, a first load resistor, a second load resistor, a coupled inductor, a full-bridge circuit and a half-bridge circuit. Compared with the existing converter, only the magnetic coupling effect of a coupled inductor is used to transfer energy between the primary and secondary sides, thereby reducing the number of magnetic components and supporting the self-balancing of the bipolar output voltage. The control method provided performs linear fitting processing on the intra-bridge phase-shift duty ratio of the full-bridge circuit through the voltage gain and the inter-bridge phase-shift duty ratio of the full-bridge circuit and the half-bridge circuit, and then adjusts the first duty ratio after the linear fitting to control the bipolar voltage self-balancing converter under two working conditions, thereby realizing zero voltage switching of the converter within the full voltage and full load range under unbalanced load conditions, thereby solving the problem that the bipolar voltage balanced output converter is difficult to achieve full load range soft switching, has a large number of devices, and is complex to control.

[0048] Other beneficial effects of the present invention will be described in detail in the subsequent specific implementation section. BRIEF DESCRIPTION OF THE DRAWINGS

[0049] Figure 1 is a schematic diagram of an embodiment of the present invention;

[0050] Figure 2 is a flow chart of a control method according to an embodiment of the present invention;

[0051] Figure 3 is a typical working waveform diagram of an embodiment of the present invention under a balanced load;

[0052] Figure 4 is a current path diagram of an embodiment of the present invention in mode 1;

[0053] Figure 5 is a current path diagram of an embodiment of the present invention in mode 2;

[0054] Figure 6 is a current path diagram of an embodiment of the present invention in mode three;

[0055] Figure 7 is a current path diagram of an embodiment of the present invention in mode 4;

[0056] Figure 8 is a typical working waveform diagram of an embodiment of the present invention under an unbalanced load;

[0057] Fig. 9 This is a linear fitting processing curve diagram of an embodiment of the present invention. DETAILED DESCRIPTION

[0058] In order to make the technical problems, technical solutions and advantages to be solved by the present invention clearer, the following will be described in detail with reference to the accompanying drawings and specific embodiments. Obviously, the described embodiments are part of the embodiments of the present invention, rather than all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0059] In the description of the present invention, it should be noted that the terms "center", "upper", "lower", "left", "right", "vertical", "horizontal", "inner", "outer", etc., indicating the orientation or positional relationship, are based on the orientation or positional relationship shown in the drawings, and are only for the convenience of describing the present invention and simplifying the description, rather than indicating or implying that the device or element referred to must have a specific orientation, be constructed and operated in a specific orientation, and therefore cannot be understood as limiting the present invention. In addition, the terms "first", "second", and "third" are used for descriptive purposes only, and cannot be understood as indicating or implying relative importance.

[0060] In the description of the present invention, it should be noted that, unless otherwise clearly specified and limited, the terms "installation", "connection" and "connection" should be understood in a broad sense, for example, it can be a locking connection, a detachable connection, or an integral connection; it can be a mechanical connection or an electrical connection; it can be a direct connection or an indirect connection through an intermediate medium, or it can be the internal communication of two components. For ordinary technicians in this field, the specific meanings of the above terms in the present invention can be understood according to specific circumstances.

[0061] In addition, the technical features involved in the different embodiments of the present invention described below can be combined with each other as long as they do not conflict with each other.

[0062] The present invention aims at solving the existing problems and provides a bipolar voltage self-balancing converter and a control method.

[0063] like Figure 1 As shown, an embodiment of the present invention provides a bipolar voltage self-balancing converter, such as Figure 1 (a), including:

[0064] First port, second port, first capacitor, first output capacitor C o1 , the second output capacitor C o2 , a first load resistor R1, a second load resistor R2, a coupled inductor, a full-bridge circuit and a half-bridge circuit;

[0065] The positive electrode of the first port is respectively connected to the first end of the first capacitor and the first input end of the full-bridge circuit;

[0066] The negative electrode of the first port is respectively connected to the second end of the first capacitor and the second input end of the full-bridge circuit;

[0067] The first end of the coupled inductor is connected to the midpoint of the first bridge arm in the full-bridge circuit;

[0068] The second end of the coupled inductor is connected to the midpoint of the second bridge arm in the full-bridge circuit;

[0069] The third end of the coupled inductor is connected to the midpoint of the bridge arm in the half-bridge circuit;

[0070] The fourth end of the coupled inductor is connected to the first output capacitor C o1 The second end of the second output capacitor C o2 The first end of the first load resistor R1, the second end of the second load resistor R2 are connected;

[0071] The positive electrode of the second port is respectively connected to the first end of the first load resistor R1 and the first output capacitor C o1 connected to a first end of the half-bridge circuit and a first output end of the half-bridge circuit;

[0072] The negative electrode of the second port is connected to the second end of the second load resistor R2 and the second output capacitor C o2 and a second output end of the half-bridge circuit.

[0073] In the embodiment of the present invention, the first port is a power input terminal, the second port is a load output terminal, and the coupling coefficient of the coupling inductor is K.

[0074] Specifically, if Figure 1 As shown in (b), the coupled inductor is equivalent to the leakage inductance L k , Excitation inductance L m and ideal transformer;

[0075] Leakage inductance L k The first end of is connected to the midpoint of the first bridge arm in the full-bridge circuit, and the leakage inductance L k The second end of is connected to the first end of the ideal transformer;

[0076] The second end of the ideal transformer is connected to the midpoint of the second bridge arm in the full-bridge circuit;

[0077] The third terminal of the ideal transformer is connected to the excitation inductance L m The first end of the half-bridge circuit is connected to the midpoint of the bridge arm;

[0078] The fourth terminal of the ideal transformer is connected to the excitation inductance L m The second end of the first output capacitor C o1 The second end of the second output capacitor C o2 The first end of the first load resistor R1, the second end of the first load resistor R1, and the first end of the second load resistor R2 are connected.

[0079] Specifically, the full-bridge circuit includes a first switch tube S1, a second switch tube S2, a third switch tube S3, and a fourth switch tube S4;

[0080] The drain of the first switch tube S1 is connected to the first capacitor C in The first end of the first switch tube S3 is connected to the positive electrode of the first port and the drain of the third switch tube S3;

[0081] The source of the first switch tube S1 is connected to the drain of the second switch tube S2 to form a first bridge arm of the full-bridge circuit. The midpoint of the first bridge arm is point A. The midpoint of the first bridge arm is connected to the leakage inductance L k A first end is connected;

[0082] The source of the third switch tube S3 is connected to the drain of the fourth switch tube S4 to form a second bridge arm of the full-bridge circuit. The midpoint of the second bridge arm is point B, and the midpoint of the second bridge arm is connected to the second end of the ideal transformer.

[0083] The source of the fourth switch tube S4 is connected to the source of the second switch tube S2 and the first capacitor C in The second end of the first port is connected to the negative electrode of the first port.

[0084] Specifically, the half-bridge circuit includes a fifth switch tube Q1 and a sixth switch tube Q2;

[0085] The source of the fifth switch tube Q1 is connected to the drain of the sixth switch tube Q2 to form a bridge arm of a half-bridge circuit. The midpoint of the bridge arm of the half-bridge circuit is point C. The midpoint of the bridge arm is connected to the third end of the ideal transformer and the excitation inductor L. m A first end is connected;

[0086] The drain of the fifth switch tube Q1 and the first output capacitor C o1 The first end of the first load resistor R1 and the positive electrode of the second port are connected;

[0087] The source of the sixth switch tube Q2 and the second output capacitor C o2 The second end of the second load resistor R2, and the negative electrode of the second port are connected.

[0088] In the embodiment of the present invention, the first output capacitor C o1 and the second output capacitor C o2 It also forms a set of bridge arms, with the midpoint of the bridge arm being point D. The four terminals of the coupled inductor are respectively connected to the midpoints of the four bridge arms A, B, C, and D. No additional voltage balancer is introduced. The bipolar side only includes two switching tubes and two capacitors. Regardless of whether the load is balanced or not, the converter has equal bipolar output voltage by relying on the magnetizing effect of the coupled inductor.

[0089] like Figure 2As shown, the present invention also provides a control method for a bipolar voltage self-balancing converter, which is applied to a bipolar voltage self-balancing converter. The control method includes:

[0090] Step 1, collecting input voltage sampling values, output voltage sampling values, and output power of the bipolar voltage self-balancing converter;

[0091] Step 2, obtaining a voltage gain based on the input voltage sampling value and the output voltage sampling value;

[0092] Step 3, setting the intra-bridge phase shift duty ratio of the full-bridge circuit to a first duty ratio, the inter-bridge phase shift duty ratio of the full-bridge circuit and the half-bridge circuit to a second duty ratio, and dividing the bipolar voltage self-balancing converter into two operating conditions according to the positional relationship between the first duty ratio and the second duty ratio;

[0093] Step 4, calculating the first duty cycle under the two working conditions by using the output power, the second duty cycle, and the voltage gain, and performing linear fitting processing on the first duty cycle under the two working conditions according to the second duty cycle between the starting point and the end point of each working condition to obtain the first duty cycle after linear fitting under the two working conditions;

[0094] Step 5, controlling the bipolar voltage self-balancing converter under the two working conditions by adjusting the first duty cycle after linear fitting under the two working conditions.

[0095] Most preferably, the voltage gain k is calculated as:

[0096]

[0097] Where n represents the transformation ratio of the ideal transformer, V in Indicates the input voltage sampling value, V o Indicates the output voltage sampling value.

[0098] Specifically, the embodiment of the present invention adopts an extended phase shift control strategy (Extended Phase Shift, EPS), that is, on the basis of the single phase shift control strategy, the bridge phase shift duty ratio D1 is added as the control quantity, so that the reflux power of the converter is reduced and the working performance is improved. That is, in the embodiment of the present invention, the bridge phase shift duty ratio D1 of the full-bridge circuit and the bridge phase shift duty ratios of the full-bridge circuit and the half-bridge circuit are set. As a control variable to enhance the controllability of the converter.

[0099] According to the phase shift duty ratio D1 within the bridge and the phase shift duty ratio between bridges The converter can be divided into three working conditions according to the position relationship of When it is less than 0.5, the return current is small. Therefore, in practical applications, only two working conditions are used, namely working condition (1) and working condition (2). The corresponding typical working waveforms are shown in the figure below. Figure 3 shown.

[0100] The embodiment of the present invention analyzes each period of working condition (1) and working condition (2) by analogical reasoning to avoid repeated analysis. Specifically, the modal analysis is performed by taking working condition (1) as an example. Figure 3 (a) It can be seen that t0-t6 is a complete switching cycle, and a switching cycle includes six switching modes. Due to the symmetry of the topology and the working process, the embodiment of the present invention only analyzes half of the switching cycle, which is as follows:

[0101] Mode zero (before time t0): The circuit path of this mode is as follows Figure 4 As shown, in this mode, the first switch tube and the third switch tube are in the on state, and the leakage inductance L k Stored energy, leakage current Continue to increase until it reaches a peak value, at which point v AB =0, v CD =V o2 , the secondary current flows through the sixth switch tube;

[0102] Mode 1 (t0~t1): The circuit path of this mode is as follows Figure 5 As shown, when the first switch tube, the third switch tube and the fifth switch tube are in the on state, this mode begins, the secondary current flows through the fifth switch tube, and the leakage inductance L k Release the stored energy, v AB =0, v CD =V o1 , leakage current Linearly decreases, at this time, the leakage current and the magnetizing inductor current The change in is expressed as follows:

[0103]

[0104] Mode 2 (t1~t2): The circuit path of this mode is as follows Figure 6 As shown, this mode starts when the first switch tube, the fourth switch tube and the fifth switch tube are in the on state. In this mode, the full-bridge circuit is connected to the power supply, the leakage inductance stores energy, the leakage inductance current begins to increase, and the secondary current continues to flow through the fifth switch tube. At this time, v AB =V in 、v CD =V o1 , leakage current and the magnetizing inductor current The change in is expressed as follows:

[0105]

[0106] Mode 3 (t2~t3): The current path of this mode is as follows Figure 7 As shown, this mode starts when the first switch is turned off, and the second, fourth and fifth switches are in the on state. In this mode, the full-bridge circuit is disconnected from the power supply, the leakage inductance releases energy, the leakage inductance current begins to decrease, and the secondary current continues to flow through the fifth switch. AB =0, v CD =V o1 , leakage current and the magnetizing inductor current The change in is expressed as follows:

[0107]

[0108] Among them, n E Represents the primary-to-secondary turns ratio of an ideal transformer, V in Indicates the input voltage sampling value.

[0109] The above formula can be used to calculate the leakage current value at each moment and obtain the expression of the average output power. It can be seen from the expression that the transmission power is only related to the phase-shift duty cycle within the bridge and the phase-shift duty cycle between bridges, that is, it is only related to the first duty cycle and the second duty cycle.

[0110] In order to facilitate subsequent analysis, the embodiment of the present invention uses the basic power to normalize the output power of the converter, where:

[0111] Basic power P base The expression is:

[0112] P base =V in 2 / 8L k ·f s

[0113] Among them, f s Indicates frequency, T s Indicates the cycle.

[0114] The normalized expression of the first duty cycle under two working conditions is calculated by using the output power, the second duty cycle, and the voltage gain:

[0115]

[0116] in, represents the output power of the bipolar voltage self-balancing converter under working condition 1, represents the output power of the bipolar voltage self-balancing converter under working condition 2, D 11Indicates the first duty cycle under working condition 1, D 12 represents the first duty cycle under working condition 2, represents the second duty cycle under working condition 1, Indicates the second duty cycle under working condition 2.

[0117] The above analysis is based on the case of balanced load. Under this condition, there is no DC bias on the secondary side. The circuit works similarly to the traditional DAB converter under EPS control. However, under unbalanced load, the operating conditions of the converter secondary side will change. However, even when the bipolar port is connected to an unbalanced load, the converter can also self-balance the bipolar voltage, such as Figure 8 As shown, this illustrates that the operating waveform of the proposed converter under unbalanced load conditions is not affected by the load.

[0118] In the embodiment of the present invention, the total power is represented by P, and the bipolar output power is represented by P o1 and P o2 Indicates that through Figure 8 P o1 <P o2 The load condition is analyzed. At this time, V o2 The power demand on the side increases, resulting in the excitation inductance L m Generates a positive DC bias current; when the first load resistance is greater than the second load resistance, the initial bipolar voltage V o1 >V o2 , this voltage difference in the excitation inductance L m The DC bias current then flows into V o2 side, this current compensates for the V o2 The power on the two sides is increased, and the voltage on the two ports is symmetrical;

[0119] According to the excitation inductance L m The volt-second area balance principle gives the following formula:

[0120] V o1 D = V o2 (1-D)

[0121] Wherein, D represents the duty cycle of the fifth switch tube.

[0122] In the embodiment of the present invention, as long as the switch tubes on the secondary side are complementary and turned on and the duty cycle D is set to 0.5, bipolar voltage balance can be automatically achieved. In addition, for the provided converter, a half-bridge circuit is adopted, and the secondary side can automatically adjust the secondary voltage balance of the transformer to avoid transformer saturation caused by problems such as mismatch of driving signals.

[0123] In order to prove that the linear fitting process can simplify the calculation, the embodiment of the present invention first lists the effective value representation of the leakage inductance under two working conditions and performs standardization processing, which is recorded as The expression is as follows:

[0124] Working condition (1):

[0125]

[0126] Working condition (2):

[0127]

[0128] Taking minimizing the effective value of leakage inductance current as the optimization goal, the first duty cycle D under two working conditions is 11 , D 12 By taking partial derivatives, we can find that when the first duty cycle reaches the following values ​​under different conditions, the normalized leakage inductance has a minimum effective value, and the expression is as follows:

[0129] Working condition (1):

[0130]

[0131] Working condition (2):

[0132]

[0133] From the above two equations, we can know that the minimization function D corresponding to the effective value of the leakage inductance current is 11_minRMS , D 12_minRMS is the second duty cycle under two working conditions This makes the calculation process complicated. Therefore, in order to simplify the calculation, the embodiment of the present invention linearizes the first duty cycle, specifically:

[0134] Between the starting point and the end point of each working condition, a linear fitting process is performed on the first duty cycle under the two working conditions according to the second duty cycle. The linear fitting process curve is as follows: Fig. 9 As shown, the expression of the first duty cycle after linear fitting under two working conditions is:

[0135]

[0136] Among them, D 11_minLINE represents the first duty cycle after linear fitting under working condition 1, D 12_minLINE Represents the second duty cycle after linear fitting under working condition 2.

[0137] In the embodiment of the present invention, the linear fitting process replaces the leakage current minimization control, and the leakage current effective value I after minimization is LK_RMS Defined as I LkRMS_min , linear fitting processing corresponds to ILkLINE_min , and then define the error as the ratio of the difference between the two and the minimum RMS value of the leakage inductance current, calculated as:

[0138]

[0139] According to the change of voltage gain, different error values ​​are calculated, and the maximum value is recorded as the maximum error value ΔE max , and obtain the statistical list shown in Table 1 below:

[0140] Table 1

[0141] k 1.2 1.4 1.6 1.8 1.9 <![CDATA[Operating condition (1) ΔE max (%)]]> 4.1 2.8 2 1.6 1.4 <![CDATA[Operating condition (2) ΔE max (%)]]> 2.4 1.8 1.3 1 0.8

[0142] It can be seen from Table 1 above that the ΔE corresponding to different voltage gains k under the two working conditions max It can be statistically obtained that the maximum error is about 4.1%; obviously, the current effective values ​​of the two methods are highly similar at any k; therefore, the use of linear fitting is a feasible method to replace the current leakage inductance current effective value minimum control and simplify the calculation; and combining these two situations, the linear fitting line is as follows Fig. 9 As shown by the horizontal dotted line in , the area filled with diagonal lines is the range in which the switch tube can achieve zero voltage turn-on, indicating that the zero voltage turn-on of the switch tube can be achieved under linear fitting processing in all cases.

[0143] The converter provided by the embodiment of the present invention includes a first port, a second port, a first capacitor, a first output capacitor, a second output capacitor, a first load resistor, a second load resistor, a coupled inductor, a full-bridge circuit and a half-bridge circuit; compared with the existing converter, the embodiment of the present invention only uses the magnetic coupling effect of a coupled inductor to transfer energy between the primary and secondary sides, reduces the number of magnetic components and supports the self-balancing of the bipolar output voltage; the control method provided performs linear fitting processing on the intra-bridge phase-shift duty cycle of the full-bridge circuit through the voltage gain, the inter-bridge phase-shift duty cycle of the full-bridge circuit and the half-bridge circuit, and adjusts the first duty cycle after the linear fitting to achieve control of the bipolar voltage self-balancing converter under two working conditions, thereby achieving zero voltage switching of the converter within the full voltage and full load range under unbalanced load conditions, thereby solving the problem that the bipolar voltage balanced output converter is difficult to achieve full load range soft switching and has a large number of devices and complex control.

[0144] The above is a preferred embodiment of the present invention. It should be pointed out that for ordinary technicians in this technical field, several improvements and modifications can be made without departing from the principles of the present invention. These improvements and modifications should also be regarded as the scope of protection of the present invention.

Claims

1. A bipolar voltage self-balancing converter, characterized in that: include: A first port, a second port, a first capacitor, a first output capacitor, a second output capacitor, a first load resistor, a second load resistor, a coupled inductor, a full-bridge circuit, and a half-bridge circuit; The positive electrode of the first port is connected to the first end of the first capacitor and the first input end of the full-bridge circuit respectively; The negative electrode of the first port is connected to the second end of the first capacitor and the second input end of the full-bridge circuit respectively; The first end of the coupled inductor is connected to the midpoint of the first bridge arm in the full-bridge circuit; The second end of the coupled inductor is connected to the midpoint of the second bridge arm in the full-bridge circuit; The third end of the coupled inductor is connected to the midpoint of the bridge arm in the half-bridge circuit; The fourth end of the coupled inductor is respectively connected to the second end of the first output capacitor, the first end of the second output capacitor, the second end of the first load resistor, and the first end of the second load resistor; The positive electrode of the second port is respectively connected to the first end of the first load resistor, the first end of the first output capacitor, and the first output end of the half-bridge circuit; The cathode of the second port is respectively connected to the second end of the second load resistor, the second end of the second output capacitor, and the second output end of the half-bridge circuit.

2. The bipolar voltage self-balancing converter according to claim 1, characterized in that: The coupled inductor is equivalent to leakage inductance, magnetizing inductance and ideal transformer; The first end of the leakage inductor is connected to the midpoint of the first bridge arm in the full-bridge circuit, and the second end of the leakage inductor is connected to the first end of the ideal transformer; The second end of the ideal transformer is connected to the midpoint of the second bridge arm in the full-bridge circuit; The third end of the ideal transformer is connected to the first end of the excitation inductor and the midpoint of the bridge arm in the half-bridge circuit respectively; The fourth end of the ideal transformer is respectively connected to the second end of the excitation inductor, the second end of the first output capacitor, the first end of the second output capacitor, the second end of the first load resistor, and the first end of the second load resistor.

3. The bipolar voltage self-balancing converter according to claim 2, characterized in that: The full-bridge circuit includes a first switch tube, a second switch tube, a third switch tube, and a fourth switch tube; The drain of the first switch tube is respectively connected to the first end of the first capacitor, the positive electrode of the first port, and the drain of the third switch tube; The source of the first switch tube is connected to the drain of the second switch tube to form a first bridge arm of the full-bridge circuit, and the midpoint of the first bridge arm is connected to the first end of the leakage inductance; The source of the third switch tube is connected to the drain of the fourth switch tube to form a second bridge arm of the full-bridge circuit, and the midpoint of the second bridge arm is connected to the second end of the ideal transformer; The source of the fourth switch tube is connected to the source of the second switch tube, the second end of the first capacitor, and the negative electrode of the first port respectively.

4. The bipolar voltage self-balancing converter according to claim 3, characterized in that: The half-bridge circuit includes a fifth switch tube and a sixth switch tube; The source of the fifth switch tube is connected to the drain of the sixth switch tube to form a bridge arm of the half-bridge circuit, and the midpoint of the bridge arm is connected to the third end of the ideal transformer and the first end of the excitation inductor; The drain of the fifth switch tube is connected to the first end of the first output capacitor, the first end of the first load resistor, and the positive electrode of the second port; The source of the sixth switch tube is connected to the second end of the second output capacitor, the second end of the second load resistor, and the negative electrode of the second port.

5. The bipolar voltage self-balancing converter according to claim 4, characterized in that: The first port is a power input terminal; The second port is a load output terminal.

6. A control method for a bipolar voltage self-balancing converter, characterized in that: Applied to the bipolar voltage self-balancing converter according to any one of claims 1 to 5, the control method comprises: Step 1, collecting input voltage sampling values, output voltage sampling values, and output power of the bipolar voltage self-balancing converter; Step 2, obtaining a voltage gain based on the input voltage sampling value and the output voltage sampling value; Step 3, setting the intra-bridge phase shift duty cycle of the full-bridge circuit to a first duty cycle, setting the inter-bridge phase shift duty cycle of the full-bridge circuit and the half-bridge circuit to a second duty cycle, and dividing the bipolar voltage self-balancing converter into two operating conditions according to the positional relationship between the first duty cycle and the second duty cycle; Step 4, calculating the first duty cycle under the two working conditions by using the output power, the second duty cycle, and the voltage gain, and performing linear fitting processing on the first duty cycle under the two working conditions according to the second duty cycle between the starting point and the end point of each working condition to obtain the first duty cycle after linear fitting under the two working conditions; Step 5, controlling the bipolar voltage self-balancing converter under the two working conditions by adjusting the first duty cycle after linear fitting under the two working conditions.

7. The control method of the bipolar voltage self-balancing converter according to claim 6, characterized in that: The calculation expression of the voltage gain k is: Where n represents the transformation ratio of the ideal transformer, V in Indicates the input voltage sampling value, V o Indicates the output voltage sampling value.

8. The control method of the bipolar voltage self-balancing converter according to claim 7, characterized in that: The expression for calculating the first duty cycle under two working conditions by using the output power, the second duty cycle, and the voltage gain is: in, represents the output power of the bipolar voltage self-balancing converter under working condition 1, represents the output power of the bipolar voltage self-balancing converter under working condition 2, D 11 Indicates the first duty cycle under working condition 1, D 12 represents the first duty cycle under working condition 2, represents the second duty cycle under working condition 1, Represents the second duty cycle under working condition 2.

9. The control method of the bipolar voltage self-balancing converter according to claim 8, characterized in that: Between the starting point and the end point of each working condition, a linear fitting process is performed on the first duty cycle under the two working conditions according to the second duty cycle, and the expression of the first duty cycle after linear fitting under the two working conditions is obtained as follows: Among them, D 11_minLINE represents the first duty cycle after linear fitting under working condition 1, D 12_minLINE Represents the second duty cycle after linear fitting under working condition 2.

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