Harmonic suppression method and device for three-phase three-wire cascade power conversion device

Abstract

The invention provides a harmonic suppression method and device for a three-phase three-wire cascade power conversion device, and the method comprises the steps: calculating a first current outputted by a full-bridge rectification module to a floating DC side, the first current being composed of a first current DC component and a first current double frequency component; generating a first injection voltage of triple frequency to inject the first injection voltage to a floating direct current side, transmitting at least one part of a double-frequency component of the first current to a low-voltage direct current side, and enabling the total current of the floating direct current side to be composed of the first current and a second current after the first injection voltage is injected, the second current is composed of a second current double frequency component and a second current quadruplicated frequency component, and the amplitude and phase angle of the first injection voltage are configured such that the first and second current double frequency components are at least partially cancelled out. According to the method, the requirement for floating direct current side capacitance of the power conversion device is reduced, so that the power density is greatly improved, and the power conversion efficiency is high.

Description

Technical Field

[0001] The present invention relates to the field of power management and control, and in particular to a harmonic suppression method and device for a three-phase three-wire cascade power conversion device. Background Art

[0002] The cascade H-bridge power conversion device using modular cascade topology is used to convert the power of the medium-voltage power grid into DC power on the low-voltage DC side. The cascade H-bridge power conversion device not only has the characteristics of high power density and small size, but also has outstanding advantages such as a large number of levels, good harmonic characteristics, and fast adjustment speed. It is very suitable for ultra-large-scale power supply.

[0003] Figure 1 The topological structure diagram of the modular cascade H-bridge power conversion device is shown. The structure of each phase of the R, S and T phases of the cascade H-bridge power conversion device is consistent. The structure of any one stage of each phase mainly includes a rectifier module ( Figure 1 The left AC / DC in the figure), a floating DC side for receiving the rectified power output by the H-bridge rectifier module, and isolating and converting the power on the floating DC side to Figure 1 The dual active bridge (D-AB) module on the low voltage DC side on the right (including Figure 1 The middle DC / AC, isolation transformer and right AC / DC in the figure), where the full-bridge rectifier modules of each stage of each phase are connected in series at the input end, and all the dual-active bridge modules of the three phases are connected in parallel on the low-voltage DC side except for the bypass stage. The typical medium-voltage AC input side voltage is about 10KV AC voltage, the floating DC side voltage is about 9600V DC voltage, and the low-voltage DC side is greatly reduced to 800V DC voltage. Therefore, it is convenient to cascade the H-bridge power conversion device to convert the medium-voltage grid power to the low-voltage DC side and invert the output to the load. In order to more clearly show the circuit connection relationship of the cascaded H-bridge power conversion device, Figure 1 Circuit modules such as a DC / AC inverter part, a load, a control device, and a detection device for detecting voltage connected to the low-voltage DC side are not shown.

[0004] The cascaded H-bridge power conversion device in the prior art has voltage harmonics of the second order or higher on the floating DC side, which must be eliminated before it enters the AC input grid. Therefore, the floating DC side is usually equipped with a capacitor ( Figure 1 The capacitance value depends on the scale of the high-order voltage harmonics. This capacitor must be arranged in each stage, resulting in the current cascaded H-bridge power conversion device often having low power density and bulky size. Summary of the invention

[0005] In view of the above problems existing in the prior art, a first aspect of the present invention provides a method for controlling a three-phase three-wire cascade H-bridge power conversion device, wherein the cascade H-bridge power conversion device has a plurality of stages with the same structure, each stage comprising a full-bridge rectifier module, a floating DC side, a dual-active bridge converter and a low-voltage DC side connected in sequence, wherein the floating DC side is connected in parallel with a capacitor, the full-bridge rectifier module rectifies an AC input voltage with a base frequency into a DC voltage and outputs it to the floating DC side, and the dual-active bridge converter receives the DC voltage and converts it and outputs it to the low-voltage DC side;

[0006] Wherein, the method comprises:

[0007] Calculating a first current outputted from the full-bridge rectifier module to the floating DC side when the dual-active bridge converter performs DC-DC conversion, wherein the first current is composed of a first current DC component and a first current double frequency component;

[0008] Generate a first injection voltage having a triple frequency of the fundamental frequency to inject it into the floating DC side, and transfer at least a portion of the double frequency component of the first current to the low voltage DC side, wherein,

[0009] The total current of the floating DC side is composed of the first current and the second current after the first injection voltage is injected, wherein the second current is composed of a second current double frequency component and a second current quadruple frequency component, and wherein the amplitude and phase angle of the first injection voltage are configured so that the first current double frequency component and the second current double frequency component at least partially cancel each other out.

[0010] Preferably, in the method for controlling a three-phase three-wire cascade H-bridge power conversion device, when the first injection voltage is not injected, the voltage of the full-bridge rectifier module is expressed as:

[0011] U′ in =U f U dc sin(ωt)

[0012] The first injection voltage is expressed as:

[0013] U in j sin(3ωt)=U 3rd U dc sin(3ωt), and

[0014] When the first injection voltage is injected, the voltage of the full-bridge rectifier module is expressed as:

[0015] U in =U′ in +U in j-3 sin(3ωt)=Uf U dc sin(ωt)+U 3rd U dc sin(3ωt)

[0016] Among them U f is the conversion coefficient of the full-bridge rectifier module, U dc is the voltage of the floating DC side, ω is the AC input frequency, and wherein the amplitude of the first injection voltage is configured to satisfy the following first reference wave amplitude constraint to avoid waveform distortion of the full-bridge rectifier module when injecting the first injection voltage:

[0017] |U f sin(ωt)+U 3rd sin(3ωt)|≤1.

[0018] Preferably, in the method for controlling the three-phase three-wire cascade H-bridge power conversion device, the current of the single-phase AC input is I m sin(ωt), the total current of the floating DC side when the first injection voltage is injected is expressed as

[0019]

[0020] Among them, A=-(U f -U 3rd ), B=-U 3rd ,as well as

[0021] The amplitude of the first injection voltage is configured to satisfy the first reference wave amplitude constraint. Minimum.

[0022] Preferably, in the method for controlling a three-phase three-line cascaded H-bridge power conversion device, the first injection voltage is configured to satisfy:

[0023] U 3rd =U f / 2.

[0024] Preferably, each phase of the three-phase three-wire cascaded H-bridge power conversion device includes at least one bypass stage and one redundant stage; the method also includes: for each phase, in addition to the first injection voltage, generating a second injection voltage having a frequency five times the fundamental frequency and a third injection voltage having the same frequency but opposite phase; controlling the full-bridge rectifier module to superimpose the second injection voltage, the AC input voltage, and the first injection voltage and output them to the floating DC side of the redundant stage, wherein the total current of the floating DC side is composed of the first, second and third currents after the first and second injection voltages are injected, wherein the third current is composed of a third current four-fold frequency component and a third current six-fold frequency component, and wherein the amplitude and phase angle of the second injection voltage are configured so that the second current four-fold frequency component and the third current four-fold frequency component at least partially offset each other; controlling the full-bridge rectifier module to inject the third injection voltage into the floating DC side of the bypass stage.

[0025] Preferably, in the method for controlling a three-phase three-wire cascade H-bridge power conversion device, when the first injection voltage is not injected, the voltage of the full-bridge rectifier module is expressed as:

[0026] U in =U f U dc sin(ωt)

[0027] The first injection voltage is expressed as:

[0028] U in j-3 sin(3ωt)=U 3rd U dc sin(3ωt),

[0029] The second injection voltage is expressed as:

[0030] U in j-5th sin(5ωt)=U 5th U dc sin(5ωt),

[0031] Among them U f is the conversion coefficient of the full-bridge rectifier module, U dc is the voltage of the floating DC side, ω is the AC input frequency, and wherein the amplitudes of the first and second injection voltages are configured to satisfy the following second reference wave amplitude constraint to reduce the waveform distortion of the full-bridge rectifier module when injecting the first and second injection voltages:

[0032] |U f sin(ωt)+U 3rd sin(3ωt)+U 5th sin(5ωt)|≤1.

[0033] Preferably, in the method for controlling the three-phase three-wire cascade H-bridge power conversion device, the current of the single-phase AC input is expressed as I m sin(ωt), the total current of the floating DC side when the first injection voltage is injected is expressed as

[0034]

[0035] Among them, A=-(U f -U 3rd ), B=-(U 3rd -U 5th ), C=-U 5th ,as well as

[0036] The first and second injection voltages are configured to satisfy the second reference wave amplitude constraint. Minimum.

[0037] Preferably, in the method for controlling a three-phase three-wire cascaded H-bridge power converter, at least one of the three phases of the cascaded H-bridge power converter includes a redundant stage, and the method further includes:

[0038] After the other stages except the redundant stage are replaced, the redundant stage is controlled to inject an injection voltage having a frequency three times and five times of the AC input, and the other stages are controlled to inject an injection voltage having a frequency three times of the AC input.

[0039] Preferably, in the method of controlling a three-phase three-line cascaded H-bridge power conversion device, at least 90% of the doubled frequency component of the first current is transmitted to the low-voltage DC side through the dual-active bridge converter; or all of the doubled frequency component of the first current is transmitted to the low-voltage DC side through the dual-active bridge converter so that the floating DC side retains only the DC component.

[0040] Preferably, in the method for controlling a three-phase three-line cascaded H-bridge power conversion device, the first injection voltage amplitude and phase angle are configured so that the first and second doubled frequency current components are completely offset.

[0041] Another aspect of the present invention provides a three-phase three-wire cascade H-bridge power conversion device, which is configured to implement the control method described in the first aspect of the present invention.

[0042] Preferably, each phase of the three-phase three-wire cascade H-bridge power conversion device includes at least one bypass stage and one redundant stage; the method further includes: for each phase, in addition to the first injection voltage, generating a second injection voltage with a frequency five times the fundamental frequency and a third injection voltage with the same frequency but opposite phase; controlling the full-bridge rectifier module to superimpose the second injection voltage, the AC input voltage, and the first injection voltage and output them to the floating DC side of the redundant stage, wherein the total current of the floating DC side is composed of the first, second, and third currents after the first and second injection voltages are injected, wherein the third current is composed of a third current quadruple frequency component and a third current sextuple frequency component, and wherein the amplitude and phase angle of the second injection voltage are configured so that the second current quadruple frequency component and the third current quadruple frequency component at least partially offset each other; controlling the full-bridge rectifier module to inject the third injection voltage into the floating DC side of the bypass stage. The redundant stage and the bypass stage are configured as follows:

[0043] When the total factory working time of the redundant stage is longer than that of other stages in the phase,

[0044] injecting the first injection voltage into the full-bridge rectifier modules of other stages except the redundant stage,

[0045] The first and second injection voltages are injected into the full-bridge rectifier module of the redundant stage, and the third injection voltage is injected into the full-bridge rectifier module of the bypass stage, wherein the bypass stage does not participate in the DC-DC conversion from the floating DC side to the low-voltage DC side.

[0046] The three-phase three-wire cascade H-bridge power conversion device and the control method thereof according to the present invention can efficiently transfer the undesired ripples on the floating DC side of each level to the low-voltage DC side for elimination, while also ensuring a low power loss of the DC-DC conversion unit. Therefore, it can realize the miniaturization of the floating DC side capacitor, so that the power density of the three-phase three-wire cascade H-bridge power conversion device is significantly improved and the power conversion efficiency is high.

[0047] The floating DC side is usually equipped with a capacitor ( Figure 1 The capacitance value depends on the scale of the high-order voltage harmonics. This capacitor must be arranged in each stage, resulting in the current cascaded H-bridge power conversion device often having low power density and bulky size. BRIEF DESCRIPTION OF THE DRAWINGS

[0048] The embodiments of the present invention are further described below with reference to the accompanying drawings, in which:

[0049] Figure 1 A schematic diagram of the topological structure of a modular cascaded H-bridge power conversion device is shown;

[0050] Figure 2 Shows Figure 1 One stage of one of the phases of the H-bridge cascade bridge power conversion device shown;

[0051] Figure 3 The figure shows the current waveform flowing through the dual active bridge (D-AB) module when the DC-DC conversion is performed using the pulse width modulation method in normal operation;

[0052] Figure 4 The figure shows the current waveform flowing through the dual active bridge (D-AB) module when the double frequency waveform of the floating DC side is transferred to the low voltage DC side by using the pulse width modulation method. DETAILED DESCRIPTION

[0053] In order to make the purpose, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below through specific embodiments in conjunction with the accompanying drawings.

[0054] Figure 2 Shows Figure 1 One stage of one of the phases of the H-bridge cascade bridge power conversion device shown. Figure 2 , this stage is configured from left to right in sequence as follows: a full-bridge rectifier module 11, which is used to receive single-phase AC power; the full-bridge rectifier module 11 outputs to a floating DC side 12, wherein a capacitor 13 is connected in parallel to the floating DC side; the dual active bridge module 14 includes a DC / AC module 141, an inductor and an isolation transformer, and an AC / DC module 142; the dual active bridge module 14 outputs the low-voltage DC power after DC-DC conversion to a low-voltage DC side 15.

[0055] If the grid input current I g for

[0056] I g =I m sin(ωt),

[0057] Voltage U' in the full-bridge rectifier module in for

[0058] U′ in =U m sin(ωt)=U f U dc sin(ωt),

[0059] Then the output power of the full-bridge rectifier module is

[0060] P in =I g U′ in =U f U dc I m sin 2(ωt),

[0061] The output current of the full-bridge rectifier module is

[0062]

[0063] Among them, I dc is the current output from the full-bridge rectifier module to the floating DC side, U f is the conversion coefficient of the full-bridge rectifier module, U dc is the voltage across the capacitor connected in parallel on the floating DC side, I m is the AC input current amplitude. It can be seen that the current I output to the floating DC side dc Contains DC component and the double frequency AC component Table 1 shows a set of typical values ​​of the above parameters in an application.

[0064] Table 1

[0065] Im 162A Um 680V Udc 800V Idc 70ADC+49.5A100Hz

[0066] As mentioned above, in order to make the waveform outputted by the dual active bridge (D-AB) converter connected to the floating DC side to the low voltage DC side closer to the DC waveform, a larger capacitor is usually connected in parallel to the floating DC side, and the capacitor is used to dc However, such a layout makes the capacitance of the capacitor too large, so it is difficult to reduce the volume of the H-bridge cascade bridge power conversion device.

[0067] In the technical solution of the present application, the inventor proposes to avoid absorbing at least a part of the double frequency AC component of the current on the floating DC side by the capacitor, but to transfer it to the low-voltage DC side for further elimination. The principle is that since the double frequency waveform components of the three phases in the three-phase power system are 120° out of phase with each other, the double frequency waveforms can be superimposed and offset on the low-voltage DC side, thereby reducing the pressure on the capacitor on the floating DC side and reducing its capacitance. However, if the double frequency AC component in the three phases is simply transferred to the low-voltage DC side without waveform adjustment and matching, the waveform changes transmitted in the dual active bridge may cause changes in the losses of various switching transistors, transformers, inductors and other devices. In order to find the optimal solution after combining capacitance reduction and loss control, the inventor further analyzed the loss of the H-bridge cascade bridge power conversion device under various states.

[0068] Figure 3 Figure 2 shows the current waveform output to the low voltage DC side when the dual active bridge (D-AB) module uses the pulse width modulation method to perform DC-DC conversion in normal operation. Figure 3As shown, due to the DC-DC conversion, the dual active bridge module outputs a square wave with substantially the same amplitude, and the current I transmitted in the dual active bridge module is p The corresponding power can be approximately considered as DC power. The dual active bridge module includes a bridge DC-AC module on the primary side, an inductor, an isolation transformer and a bridge AC-DC module on the secondary side. Therefore, during the power transmission process, the switching transistors in each module will have conduction loss and switching loss, and there will also be winding loss on the primary winding and secondary winding of the isolation transformer. Table 2 shows an example of the conduction loss, switching loss and total loss of the primary side DC-AC module and the secondary side AC-DC module of a typical dual active bridge module when the load is 100%. The total loss on the primary side is 154.84 watts, the total loss on the secondary side is 157.92 watts, and the total loss is 312.76 watts. Among them, the winding ratio between the primary side and the secondary side of the isolation transformer used in the test is 1:1, so the winding losses on both sides are close to the same.

[0069] Table 2

[0070]

[0071]

[0072] Figure 3 The square wave shown can be expanded into the superposition of odd multiples of the fundamental frequency by Fourier expansion. In the Fourier expansion, the loss of the isolation transformer winding part of the dual active bridge caused by the first five square wave frequency components of the fundamental frequency, triple frequency, quintuple frequency, septuple frequency and ninuple frequency in the square wave accounts for a relatively high proportion of the total loss. For the convenience of analysis, the sum of the losses of these odd multiple frequency components can be approximately regarded as the total loss of the isolation transformer winding. In an exemplary simulation, the fundamental frequency (the pulse width modulation frequency used in the dual active bridge power conversion) is set to 30KHz. Table 3 shows the total loss of the isolation transformer winding caused by the five multiple frequency component waveforms of the fundamental frequency (30KHz), triple frequency (90KHz), quintuple frequency (150KHz), septuple frequency (210KHz) and ninuple frequency (270KHz) and the accumulated value of the loss of these five items. It can be seen that the total loss of the isolation transformer winding is about 147.75 watts.

[0073] Table 3

[0074]

[0075] After obtaining the energy transfer loss values ​​of the dual active bridge modules in a single stage as determined in Tables 2 and 3, the inventors further tested the effect of floating the current I on the DC side. dcThe doubled frequency AC component is superimposed on the original DC component by appropriate pulse width modulation and transmitted to the low-voltage DC side by the dual active bridge module. The energy transfer loss values ​​of each part of the dual active bridge module in this scenario are obtained.

[0076] Figure 4 The figure shows the current waveform flowing through the dual active bridge (D-AB) module when the double frequency waveform of the floating DC side is transferred to the low voltage DC side by the pulse width modulation method. As mentioned above, the current I on the floating DC side dc The double frequency AC component in is If at least a portion of the doubled frequency AC component is transmitted to the output end (i.e., the low voltage DC side) of the dual active bridge module via pulse width modulation, then the current flowing through the dual active bridge module, for example, the current I transmitted in the DC-AC module on the primary side p Will have Figure 4 The saddle-shaped form shown. Figure 3 Different from the conventional high-frequency switch, the amplitude of the rectangular ripple current changes continuously rather than being constant, so that its envelope is close to a trigonometric function curve, which is a superimposed waveform of the DC component transmitted to the low-voltage DC side in conventional operation and the newly introduced double-frequency AC component transmitted to the low-voltage DC side. Figure 4 The current I shown p Only 90% of the double frequency waveform of the floating DC side and all DC components are included, but the 90% ratio is only an example and not a limitation. It can be selected between 0 and 100% according to the needs, for example, it can also be no more than 50%, 60%, 70%, 80%, 90% or 95%. Tables 4 and 5 show the loss of the dual active bridge module caused by 90% of the double frequency current being transferred from the floating DC side to the low voltage DC side. Also taking 30KHz as the fundamental frequency, Table 4 shows the conduction loss, switching loss and total loss of the primary side DC-AC module and the secondary side AC-DC module of the dual active bridge module when the load is 100%. The total loss on the primary side is 216.615 watts, the total loss on the secondary side is 220.681 watts, and the total loss on the transistor is 437.296 watts.

[0077] Table 4

[0078]

[0079] Table 5 shows the total loss of the isolation transformer winding caused by each of the five frequency-multiplied waveforms, namely the base frequency (30KHz), triple frequency (90KHz), quintuple frequency (150KHz), septuple frequency (210KHz), and ninefold frequency (270KHz), as well as the cumulative loss of these five items. It can be seen that the total loss of the isolation transformer winding is approximately 210.08 watts.

[0080] Table 5

[0081]

[0082] Compared to Figure 3 Compared with the scenario of transmitting the DC waveform in the dual active bridge module shown in FIG, once the double frequency waveform of the floating DC side is superimposed on the DC waveform and transmitted to the low voltage DC side, the transistor loss and transformer winding loss in the dual active bridge module are significantly increased.

[0083] In order to determine the source of this loss increase, the current I in the dual active bridge module when delivering 90% (i.e. 0.9) of the double frequency waveform is P Further Fourier expansion is performed. Current I P It can be expressed as the superposition of a square wave power waveform Squarwave and a double frequency waveform:

[0084]

[0085]

[0086] The first part is:

[0087]

[0088] The second part is:

[0089]

[0090] I P It can be expressed as the sum of the first and second parts. The first part is the Fourier expansion of the square wave in the frequency domain, that is, the main frequency band; and the second part is the sideband added by the transmission of the double frequency waveform. The sideband drifts by 100Hz on the left and right based on each odd frequency component of the square wave waveform in the frequency domain, that is, the drift amount is the double frequency (100Hz) of the AC power frequency (50Hz). Since the total loss can be equivalent to the sum of the loss caused by the first part and the loss caused by the second part, it can be known that the source of the loss is mainly in the second part representing the sideband. This sideband is generated by the double frequency waveform on the floating DC side.

[0091] Based on the above analysis, the technical solution of the present application adopts an active control method to reduce or even eliminate the current I in the dual active bridge. P The proportion of the above sideband current waveform, thereby achieving at least a part of the floating DC side double frequency transmission to the low voltage DC side for elimination, while also controlling the losses of power electronic devices and transformers to an ideal state lower than the values ​​shown in Tables 4 and 5. Based on this idea, the present application intends to realize a high power density current conversion device with high current quality, energy saving and low consumption, and greatly reduced floating DC side capacitance and volume.

[0092] In order to efficiently eliminate or reduce the double frequency component on the floating DC side to reduce the specification requirements of the corresponding capacitor, for the three-phase three-wire system, the inventor proposes to inject zero-sequence components (zero-sequence components refer to the components in the zero-sequence symmetrical system after the electric quantity of the unbalanced three-phase system is decomposed according to the symmetrical component method) on the rectifier side of each phase of the cascaded H-bridge current conversion device to offset the sideband of the floating DC side or reduce its proportion. The most typical method is to inject a triple frequency voltage waveform (whose frequency is three times the AC input power frequency) and the phase angle is 120° different from each other. As long as the third harmonic injected into each phase of the three-phase three-wire system is controlled to have a phase angle difference of 120°, the total superposition of the third harmonics of the three phases at the virtual neutral point is equivalent to 0, and the third harmonics injected into the three-phase three-wire system are mutually offset. However, such a third harmonic injection method is difficult to apply to a three-phase four-wire system because the zero-sequence component of the third harmonic (such as the third component or the ninth component) will produce a path. The following three specific embodiments are used to illustrate the injection method of the zero-sequence component, but the following embodiments are merely examples and are not limitations of the present application.

[0093] Example 1

[0094] In this embodiment, Figure 1 In the three-phase three-wire cascade H-bridge power conversion device shown, three-fold voltage harmonics with equal phase angle and amplitude are injected into each H-bridge module of each phase, and the voltage harmonics between the three phases are 120° out of phase with each other, so all the injected three-fold voltage harmonics cancel each other out. It should be noted that in a single-phase cascade H-bridge power conversion device, it is necessary to avoid injecting a three-fold voltage harmonic of a first phase angle into one stage, and injecting another three-fold voltage harmonic with an opposite phase angle to cancel the three-fold voltage harmonic into another stage, because although this can avoid artificially created harmonic pollution on the AC input grid side, the double-frequency voltage component cannot be reduced in the power conversion process of the dual active bridge, but may increase instead.

[0095] In addition to satisfying the phase angle constraint between the three phases, the following waveform superposition mechanism will occur when the voltage waveform is injected into the cascaded H-bridge module of each phase: the triple frequency voltage injected into a full-bridge rectifier module at a certain level is agreed to be the first injection voltage, which is expressed as:

[0096] U in j-3rd sin(3ωt)=U 3rd U dc sin(3ωt), and

[0097] When the first injection voltage is injected, the voltage of the full-bridge rectifier module is expressed as:

[0098] U in =U′in +U in j-3rd sin(3ωt)=U f U dc sin(ωt)+U 3rd U dc sin(3ωt)

[0099] The grid input current is expressed as:

[0100] I g =I m sin(ωt)

[0101] The power of the current in the full-bridge rectifier module is:

[0102] P in =I g U in =U f U dc I m sin 2 (ωt)+I m U 3rd U dc sin(3ωt)I m sin(ωt)

[0103] Then the current on the floating DC side is:

[0104]

[0105] Among them, the use of P in Computation I dc The above equation can be obtained by using the sum and difference method of trigonometric functions in the process. This equation converts the current I dc Expressed as DC component Double frequency component and the fourth frequency component Therefore, by adjusting the amplitude and phase angle of the first injection voltage, U f and U 3r at least partially cancel each other, thereby making the double frequency component The total amount of also at least partially offsets each other. Although a quadruple frequency component is also introduced in the above formula when injecting the first injection voltage of triple frequency, the energy loss of each power electronic device in the dual active bridge conversion unit caused by the quadruple frequency component is significantly lower than the loss caused by the double frequency component. Therefore, the total energy consumption is greatly reduced in theory.

[0106] However, after the above waveform control principle is determined in theory, it is necessary to further consider the possible waveform expression distortion (overmodulation) risk in the full-bridge rectifier module. For each stage, the voltage waveform in the full-bridge rectifier module after the expected first injection voltage is superimposed on the single-phase AC input waveform is:

[0107] U in =U dc (U f sin(ωt)+U 3rd sin(3ωt))

[0108] When using positive and negative U dc As the amplitude of the triangular wave carrier, and using U in When used as a reference waveform, the amplitude of the reference waveform needs to be limited to no greater than the amplitude of the carrier waveform to ensure that the carrier can accurately approach the true waveform of the reference wave. Otherwise, once the amplitude of the reference waveform is greater than the amplitude of the carrier waveform, the result of the comparison between the two will no longer effectively reflect the reference waveform, and will generate undesirable fifth and / or seventh harmonics at the output of the full-bridge rectifier due to overmodulation. Therefore, the injected first injection voltage needs to meet the following first reference wave amplitude constraint:

[0109] |U f sin(ωt)+U 3rd sin(3ωt)|≤1

[0110] Under this limited premise, it is also necessary to analyze the current waveform of the full-bridge rectifier module:

[0111] make

[0112] A=-(U f -U 3r ), B=-U 3rd ,

[0113] but

[0114] A+B=-U f

[0115] at this time,

[0116]

[0117]

[0118] It can be seen that the coefficients A and B should make the total current waveform of the second harmonic and the fourth harmonic in the above formula as small as possible. That is, when A+B=-U f Under the limitation of If the coefficients A and B corresponding to the minimum value also satisfy the first reference wave amplitude constraint as mentioned above, then such coefficients A and B are the best available values; if not, it is necessary to further calculate the total current waveform of the second harmonic and the fourth harmonic in the above formula according to the actual circuit configuration parameters so that the amplitude is as small as possible and the first reference wave amplitude constraint is satisfied, for example, by using an exhaustive method in the computational simulation.

[0119] In this embodiment, The minimum value of U 3rd Value U f / 2 satisfies the first reference wave amplitude constraint, therefore, U 3r The value should be U f / 2. The first injection voltage is:

[0120]

[0121] According to the above calculation results, the loss on the dual active bridge module after injecting the triple frequency voltage can be determined through simulation.

[0122] Tables 6 and 7 show the energy loss of the dual active bridge module when the full bridge rectifier injects triple frequency voltage and 90% of the double frequency voltage waveform is transferred from the floating DC side to the low voltage DC side by the dual active bridge module. With 30KHz as the fundamental frequency, Table 6 shows the conduction loss, switching loss and total loss of the primary side DC-AC module and the secondary side AC-DC module of the dual active bridge module when the load is 100%. The total loss on the primary side is 178.248 watts, the total loss on the secondary side is 182.083 watts, and the total loss on the switching transistor is 360.331 watts, which is significantly reduced compared to the loss of 437.296 watts when the triple frequency voltage is not injected.

[0123] Table 6

[0124]

[0125] Table 7 shows the total loss of the isolation transformer winding caused by each of the five frequency-multiplied component waveforms, namely the base frequency (30KHz), triple frequency (90KHz), quintuple frequency (150KHz), septuple frequency (210KHz), and ninefold frequency (270KHz), as well as the cumulative loss of these five items. It can be seen that the total loss of the isolation transformer winding is approximately 171.85 watts, which is also significantly smaller than the isolation transformer winding loss of 210.08 watts when the triple frequency voltage is not injected.

[0126] Table 7

[0127]

[0128] Example 2

[0129] In this embodiment, Figure 1 In the three-phase three-wire cascade H-bridge power conversion device shown, three-fold frequency voltage harmonics with equal phase angle and amplitude are injected into each H-bridge module of each phase, and the voltage harmonics between the three phases are staggered by 120°. Although the introduced three-fold frequency voltage harmonics reduce the sideband formed by the double-frequency voltage component, they also introduce four-fold frequency voltage components, which will also produce smaller but still non-negligible energy loss than the double-frequency voltage component. In order to further reduce the energy loss caused by the four-fold frequency voltage component, it is necessary to further introduce a five-fold frequency second injection voltage, and the voltage U is converted into a voltage U by a sum-difference product method similar to that in Example 1. in New quadruple frequency components and sextuple frequency components are introduced, and the total quadruple frequency components are eliminated as much as possible. However, the operation of injecting quintuple frequency is much more difficult than injecting triple frequency voltage harmonics in Example 1. Injecting triple frequency voltage harmonics in a three-phase three-wire system only requires constraining the phase angles between the three phases to be staggered by 120° to cancel each other out, but the quintuple frequency voltage harmonics themselves do not have such an advantage, and they cannot cancel out between the three phases by staggering each other. It is well known in the art that if a quintuple frequency voltage component is artificially injected into a power conversion system but cannot be effectively offset, such harmonic pollution input into the AC input power grid will have a very adverse effect.

[0130] With regard to this technical problem, the inventors approached the problem from another direction and analyzed the configuration features of each level of the three-phase three-wire cascaded H-bridge power conversion device. The cascaded H-bridge power conversion device usually has at least one redundant level in hot standby and one bypass level for bypassing the failed level. When a normally working H-bridge module of a certain level fails, the redundant level will replace it and be put into use, and the failed H-bridge module of the first level will be bypassed by the bypass level. In such a cascaded H-bridge power conversion device, if the redundant level is in hot standby state, it can be used to generate a second injection voltage expected to offset the quadruple frequency voltage. Since this five-fold frequency voltage is artificially injected into the rectifier side, it is necessary to select one level from the remaining levels cascaded with it to inject a negative five-fold frequency voltage with the opposite phase and the same amplitude as the second injection voltage. In order to prevent this negative five-fold frequency voltage from being transmitted to the low-voltage DC side through the dual active bridge module, thereby preventing the first injection voltage from effectively reducing the four-fold frequency waveform in each dual active bridge module in this phase, the bypass stage can generate a third injection voltage with a phase opposite to the second injection voltage but the same amplitude, because the bypass stage does not output any power to the low-voltage DC side.

[0131] Specifically, it is agreed that the triple frequency voltage injected into a full-bridge rectifier module of a certain level is the first injection voltage, which is expressed as:

[0132] U in j sin(3ωt)=U 3r Udc sin(3ωt);

[0133] It is agreed that one stage is selected from all n stages of the phase, and a second injection voltage is injected into its full-bridge rectifier module, which has a five-fold frequency form of the AC input frequency (base frequency), which is expressed as:

[0134] U in j-5th sin(5ωt)=U 5th U dc sin(5ωt)

[0135] Among them U f is the conversion coefficient of the full-bridge rectifier module, Udc is the voltage of the floating DC side, ω is the AC input frequency, and U 5th is the coefficient to be determined; and

[0136] A bypass stage that does not participate in transferring power from the floating DC side to the low-voltage DC side is selected from all n stages of the phase, and a third injection waveform having the same amplitude as the second injection voltage and a phase angle difference of 180° is injected into it.

[0137] When the first injection voltage and the second injection voltage are injected into a full-bridge rectifier module at a certain level at the same time,

[0138] U in =U′ in +U in j-3rd sin(3ωt)+U inj-5th sin(5ωt)

[0139] =U f U dc sin(ωt)+U 3rd U dc sin(3ωt)+U 5th U dc sin(5ωt)

[0140] The mains current of this phase:

[0141] I g =I m sin(ωt)

[0142] The power of the current in the full-bridge rectifier module is:

[0143] P in =I g U in =U f U dc I m sin 2 (ωt)+I m U 3rd Udc sin(3ωt)I m sin(ωt)+I m U 5th U dc sin(5ωt)I m sin(ωt)

[0144] Then the current on the floating DC side is expressed by the sum and difference product:

[0145]

[0146] Rewrite it as:

[0147]

[0148] Among them, A=-(U f -U 3rd ), B=-(U 3rd -U 5th ), C=-U 5th , so A+B+C=-U f ,

[0149] In order to prevent the corresponding reference waveform from being partially distorted due to overmodulation when generated by the pulse width modulation method, the second reference wave amplitude constraint should be satisfied:

[0150] U f sin(ωt)+U 3rd sin(3ωt)+U 5th sin(5ωt)≤1

[0151] The first and second injection voltages are configured such that Minimum to reduce the waveform distortion of the full-bridge rectifier module when injecting the first and second injection voltages. Similar to Example 1, if the coefficients A, B and C corresponding to the minimum value also satisfy the second reference wave amplitude constraint as described above, then such coefficients A, B and C are the best available values; if not, it is necessary to further calculate the total current waveform of the second harmonic and the fourth harmonic in the above formula according to the actual circuit configuration parameters to minimize the amplitude as much as possible and satisfy the second reference wave constraint, for example, using an exhaustive method in the computational simulation.

[0152] Example 3

[0153] For a three-phase three-wire cascade H-bridge power conversion device, assuming that the total number of stages is n, two of which are redundant stages and one is a bypass stage, the number of stages in normal operation is n-3. Since the frequency of use of the redundant stage and the bypass stage is relatively low compared to the remaining stages in normal operation, the remaining stages in normal operation are usually replaced after a certain number of years of use. However, after replacing new power electronic devices with the remaining stages in normal operation, the power electronic devices on the redundant stage and the bypass stage will have a relatively short lifespan, and are more likely to be damaged before other replaced power electronic devices. In order to avoid such a barrel effect, in another technical solution according to the present invention for transmitting the second harmonic of the floating DC side to the low-voltage DC side, a feasible method is to inject only the third harmonic into other stages except the redundant stage after replacing them so as to reduce the second harmonic loss in the dual active bridge module in the stage; at the same time, inject the third and fifth harmonics into the redundant stage with relatively aged power electronic devices at the same time so as to reduce the second and fourth harmonic losses in the dual active bridge module in the stage, and introduce only the sixth harmonic with significantly reduced contribution to the loss into the redundant stage, so that the temperature of the redundant stage is reduced relative to the temperature of the remaining stages, which can effectively shorten the subsequent aging degree of the remaining stages with newer power electronic devices and the redundant stage with relatively aged power electronic devices, and avoid the failure of the entire system due to the early scrapping of the redundant stages which account for a very small proportion of the total number of stages as much as possible.

[0154] Although the present invention has been described through preferred embodiments, the present invention is not limited to the embodiments described herein but includes various changes and modifications that may be made without departing from the scope of the present invention.

Claims

1. A method for controlling a three-phase three-wire cascade H-bridge power conversion device, wherein the cascade H-bridge power conversion device has a plurality of stages with the same structure, each stage comprising a full-bridge rectifier module, a floating DC side, a dual-active bridge converter and a low-voltage DC side connected in sequence, wherein the floating DC side is connected in parallel with a capacitor, the full-bridge rectifier module rectifies an AC input voltage with a base frequency into a DC voltage and outputs the voltage to the floating DC side, and the dual-active bridge converter receives the DC voltage and converts the DC voltage and outputs the DC voltage to the low-voltage DC side; in, The method comprises: Calculating a first current outputted from the full-bridge rectifier module to the floating DC side when the dual-active bridge converter performs DC-DC conversion, wherein the first current is composed of a first current DC component and a first current double frequency component; Generate a first injection voltage having a triple frequency of the fundamental frequency to inject it into the floating DC side, and transfer at least a portion of the double frequency component of the first current to the low voltage DC side, wherein, The total current of the floating DC side is composed of the first current and the second current after the first injection voltage is injected, wherein the second current is composed of a second current double frequency component and a second current quadruple frequency component, and wherein the amplitude and phase angle of the first injection voltage are configured so that the first current double frequency component and the second current double frequency component at least partially cancel each other out.

2. The method according to claim 1, wherein: When the first injection voltage is not injected, the voltage of the full-bridge rectifier module is expressed as: IN' in =U f IN dc sin(ωt) The first injection voltage is expressed as: U inj-3rd sin(3ωt)=U 3rd U dc sin(3ωt), hereafter When the first injection voltage is injected, the voltage of the full-bridge rectifier module is expressed as: IN in =U′ in +U inj_3rd sin(3ωt)=U f IN dc sin(ωt)+U 3r IN dc sin(3ωt) Among them U f is the conversion coefficient of the full-bridge rectifier module, U dc is the voltage of the floating DC side, ω is the AC input frequency, and wherein the amplitude of the first injection voltage is configured to satisfy the following first reference wave amplitude constraint to avoid waveform distortion of the full-bridge rectifier module when injecting the first injection voltage: |U f sin(ωt)+U 3rd sin(3ωt)|≤1。 3. The method according to claim 2, wherein: The single-phase AC input current is I m sin(ωt), the total current of the floating DC side when the first injection voltage is injected is expressed as Where A = -(Ux-U 3rd ), B=-U 3rd ,as well as The amplitude of the first injection voltage is configured to satisfy the first reference wave amplitude constraint. Minimum.

4. The method according to claim 3, wherein: The first injection voltage is configured to satisfy: IN 3r =U f / 2.

5. The method according to claim 1, wherein: Each phase of the three-phase three-wire cascade H-bridge power conversion device includes at least one bypass stage and one redundant stage; The method further comprises: For each phase, In addition to the first injection voltage, generating a second injection voltage having a frequency five times the fundamental frequency and a third injection voltage having the same frequency but opposite phase; The full-bridge rectifier module is controlled to superimpose the second injection voltage, the AC input voltage, and the first injection voltage and output them to the floating DC side of the redundant stage, wherein The floating DC side total current is composed of the first, second and third currents after the first and second injection voltages are injected, wherein the third current is composed of a third current quadruple frequency component and a third current sextuple frequency component, and wherein the amplitude and phase angle of the second injection voltage are configured such that the second current quadruple frequency component and the third current quadruple frequency component at least partially cancel each other out; The full-bridge rectifier module is controlled to inject the third injection voltage into the floating DC side of the bypass stage.

6. The method according to claim 5, wherein: When the first injection voltage is not injected, the voltage of the full-bridge rectifier module is expressed as: IN in =U f IN dc sin(ωt) The first injection voltage is expressed as: U inj sin(3ωt)=U 3rd U dc sin(3ωt), The second injection voltage is expressed as: The inj-5th sin(5ωt)=U 5th The dc sin(5ωt), Among them U f is the conversion coefficient of the full-bridge rectifier module, U dc is the voltage of the floating DC side, ω is the AC input frequency, and wherein the amplitudes of the first and second injection voltages are configured to satisfy the following second reference wave amplitude constraint to reduce the waveform distortion of the full-bridge rectifier module when injecting the first and second injection voltages: |U f sin(ωt)+U 3rd sin(3ωt)+U 5th sin(5ωt)|≤1.

7. The method according to claim 6, wherein: The current of single-phase AC input is expressed as I m sin(ωt), the total current of the floating DC side when the first injection voltage is injected is expressed as Wherein, A=-(U f -U 3rd ), B=-(U 3rd -U 5th ), C=-U 5th , and onward The first and second injection voltages are configured to satisfy the second reference wave amplitude constraint. Minimum.

8. The method according to any one of claims 1 to 7, wherein: At least one of the three phases of the cascaded H-bridge power conversion device includes a redundant stage, and the method further includes: After the other stages except the redundant stage are replaced, the redundant stage is controlled to inject an injection voltage having a frequency three times and five times of the AC input, and the other stages are controlled to inject an injection voltage having a frequency three times of the AC input.

9. The method according to any one of claims 1 to 7, wherein: Transferring at least 90% of the double frequency component of the first current to the low voltage DC side through the dual active bridge converter; Alternatively, all of the first current double frequency components are transferred to the low voltage DC side through the dual active bridge converter so that the floating DC side only retains the DC component.

10. The method according to any one of claims 1 to 7, wherein: The first injection voltage amplitude and phase angle are configured so that the first and second double frequency current components are completely offset.

11. A three-phase three-wire cascade H-bridge power conversion device, configured to implement the control method according to any one of claims 1 to 10.

12. The cascaded H-bridge power conversion device according to claim 11, characterized in that: It is configured to implement the control method according to any one of claims 5 to 7, wherein Each phase of the cascaded H-bridge power conversion device further includes a redundancy stage and a bypass stage, wherein the redundancy stage and the bypass stage are configured as follows: When the total factory working time of the redundant stage is longer than that of other stages in the phase, injecting the first injection voltage into the full-bridge rectifier modules of other stages except the redundant stage, The first and second injection voltages are injected into the full-bridge rectifier module of the redundant stage, and the third injection voltage is injected into the full-bridge rectifier module of the bypass stage, wherein the bypass stage does not participate in the DC-DC conversion from the floating DC side to the low-voltage DC side.

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