Double-sideband FSI ranging system and method based on MZM-PM cascade modulation

Abstract

The present invention discloses a double-sideband FSI ranging system and method based on MZM-PM cascade modulation, which mainly solves the problem in the prior art that the measurement of absolute distance and relative displacement can only be performed separately and the ranging accuracy is not ideal. It includes: 1) constructing an optical frequency scanning interferometer ranging system based on a cascade of a Mach-Zehnder modulator and a phase modulator and outputting an interference signal; 2) using the interference signal, a carrier and a double-frequency carrier to obtain a set of orthogonal components; 3) using the orthogonal components to obtain the sum of the upper and lower sideband phases of the interference signal and calculate the phase change to obtain the relative displacement of the object; 4) using the orthogonal components to construct a complex signal, and using the full-phase FFT to obtain the phase of the interference signal containing absolute distance information; 5) by solving the phase difference of the interference signal, the absolute distance is measured. The present invention realizes the integration of high-precision absolute distance measurement and relative displacement measurement, while improving the real-time performance of displacement measurement.

Description

Technical Field

[0001] The present invention belongs to the field of optical precision measurement and further relates to optical frequency scanning interferometry ranging technology. Specifically, it provides a double-sideband optical frequency scanning interferometry (FSI) ranging system and method based on cascade modulation of a Mach Zwedel modulator and a phase modulator (MZM-PM). The system realizes the integration of high-precision absolute distance measurement and relative displacement measurement, while improving the real-time performance of displacement measurement. Background Art

[0002] In recent years, the requirements for range and accuracy of distance measurement have become increasingly stringent in fields such as autonomous driving, high-end equipment manufacturing, and aerospace measurement. Advances in advanced optical, electronic, and computing technologies have provided new opportunities for distance measurement. FSI laser ranging technology, which utilizes advanced sensors and signal processing methods, enables accurate absolute distance measurement in large, trackless, and high-precision environments. However, in many of these fields, relative displacement measurement is equally important. For example, in automotive lidar, micro-moving targets must be identified; in lithography machine manufacturing, micro-displacements of components must be continuously monitored to ensure machining accuracy; and in gravitational wave detection, systems must monitor minute spatial deformations with extremely high precision to capture gravitational wave signals. These application scenarios demonstrate that single measurement methods often fail to meet practical requirements, as they focus on either absolute distance or relative displacement. Therefore, an integrated distance measurement system is needed that can simultaneously measure both absolute distance and relative displacement to meet the diverse demands of diverse application scenarios. Furthermore, achieving high precision while integrating distance and displacement measurement requires new measurement methods.

[0003] To achieve high-precision integration of absolute distance and relative displacement measurement, several common technologies exist: 1) Dual-wavelength laser interferometry, which utilizes two lasers of different wavelengths for interference measurement, can achieve wide-range absolute distance measurement while also providing high-precision relative displacement measurement capabilities. However, this method requires a highly stable dual-wavelength light source, resulting in a complex system architecture and high cost. 2) Optical coherence tomography (OCT), which uses a low-coherence light source reflected within a sample to form an interference signal and utilizes the depth information of the interference fringes to measure absolute distance and achieve high-precision micro-displacement measurement. However, this method has a limited measurement range and is typically used for measurements in the millimeter to centimeter range. 3) Self-mixing interferometry (SMI), which utilizes a wavelength-modulated laser source and the principle of self-mixing interferometry (SMI), is a compact, low-cost distance measurement method suitable for a variety of high-precision absolute distance and relative displacement measurement applications. However, SMI requires a certain reflectivity of the object being measured; measurement results for objects with low reflectivity are poor. Due to the limitations of the wavelength modulation frequency, the system's measurement range may not be suitable for very long distances. Furthermore, the displacement measurement accuracy of this method is lower than that of absolute distance measurement. Summary of the Invention

[0004] The present invention aims to address the above-mentioned defects of the prior art and propose a double-sideband FSI ranging system and method based on MZM-PM cascade modulation to achieve high-precision integration of absolute distance measurement and relative displacement measurement.

[0005] The basic idea of ​​implementing the present invention is as follows: first, an FSI ranging system based on a cascaded Mach-Zehnder modulator and a phase modulator (MZM-PM) is constructed to generate an interference signal; then, a set of orthogonal components are obtained by using the interference signal, a carrier, and a doubled frequency carrier. In distance measurement, this set of orthogonal components is used to construct a complex signal and obtain the interference signal phase containing absolute distance information through an all-phase fast Fourier transform (APFFT). The distance measurement result is obtained by solving the phase difference of the interference signal that continuously changes at two different times within a fixed optical frequency variation range; in displacement measurement, the phase of the interference signal containing relative displacement information is obtained by dividing it and taking the inverse tangent, and the relative displacement is calculated using the phase change.

[0006] To achieve the above-mentioned objectives, the present invention proposes a double-sideband FSI ranging system based on MZM-PM cascade modulation, comprising: a frequency-stabilized laser, a Mach-Zehnder modulator (MZM), a fiber beam splitter (FS), a phase modulator (PM), an erbium-doped fiber amplifier (EDFA), a circulator, a collimator (COL), a corner cube prism, a fiber beam combiner (FC), a photodetector (PD), and a data acquisition device. The frequency-stabilized laser is used to generate an optical signal, and the optical signal is sequentially passed through the MZM and FS before being split into two optical beams for output: the first optical beam passes through the PM and enters the FC, and the second optical beam passes through the EDFA and enters the circulator.

[0007] The second beam entering the circulator is sent to the COL and emitted by the COL, reaches the corner cube prism, is reflected, and is received by the COL to obtain a laser echo; the laser echo is returned to the circulator and output to the FC through the circulator;

[0008] The FC is used to re-converge the interference of the two incoming light beams and output them to the PD, and the interference signal is converted by the PD detection and data acquisition device;

[0009] Furthermore, the Mach-Zehnder modulator MZM is used to modulate the intensity of the optical signal generated by the frequency-stabilized laser and output it to the fiber optic splitter FS; the fiber optic splitter FS is used to split the optical signal to obtain a first light beam and a second light beam after decomposition; the phase modulator PM is used to phase modulate the first light beam and output it to the fiber optic combiner FC; the erbium-doped fiber amplifier EDFA is used to power amplify the second light beam and output it to the circulator.

[0010] At the same time, the present invention also proposes a double-sideband FSI ranging method based on MZM-PM cascade modulation, and the implementation steps include:

[0011] (1) The frequency-stabilized laser generates an optical signal S L (t) and output to the Mach-Zehnder modulator MZM;

[0012] (2) MZM realizes the optical signal S L (t) double-sideband linear frequency modulation, optical signal S L (t) After modulation, the optical signal S containing upper and lower sidebands is obtained. DSB (t);

[0013] (3) Optical signal S DSB (t) The optical signal L1 and L2 are split into two optical signals L1 and L2 by the fiber optic splitter FS, where L1 is directed to the phase modulator PM and L2 is directed to the circulator via the erbium-doped fiber amplifier EDFA; the PM phase modulates L1 to obtain the reference optical signal S RS (t); EDFA performs power amplification on L2;

[0014] (4) The power-amplified L2 is used as the measurement light signal and is emitted through the collimator COL. After reaching the corner cube prism, it is reflected and received by the collimator to obtain the laser echo signal S MS (t);

[0015] (5) The reference optical signal S RS (t) and laser echo signal S MS (t) The optical fibers are reconverged at the fiber combiner FC and interfere with each other. Finally, the interference signal I is obtained by the photodetector PD and the data acquisition device.M (t):

[0016]

[0017] Where D represents the DC offset; A represents the optical signal S L (t) AC amplitude; represents the phase shift of the light beam L1 after passing through the phase modulator; For S RS (t) and S MS The upper sideband phase obtained by the upper sideband interference of (t) is For S RS (t) and S MS (t) The lower sideband phase obtained by the lower sideband interferometry; f L is the center frequency of the narrow linewidth laser; f0 is the initial frequency of the sweep signal source, K is the sweep rate of the sweep signal source, and π is the circumference of the circle; τ(t)=(L OPD +2νt)n / c is S RS (t) and S MS (t) is the time delay corresponding to the optical path difference, v represents the speed of the object, n represents the refractive index, and c is the speed of light; L OPD =2L means S RS (t) and S MS (t), L represents the optical path difference between the two, and t represents the signal duration;

[0018] (6) Using the interference signal I M (t) and the fundamental wave cos(ω c t) and the second harmonic cos(2ω c t) Calculate the quadrature component S1(t) and the in-phase component S2(t);

[0019] (7) Using S1(t) and S2(t) to obtain the interference signal I M (t) Upper and lower sideband phase and The sum of And calculate the phase change and the relative displacement L of the object re :

[0020] (8) Using S1(t) and S2(t), and using full-phase FFT to achieve the absolute distance L ab The measurement steps are as follows:

[0021] (8.1) Using S1(t) and S2(t), construct the complex signal E1(t);

[0022] (8.2) Using the complex signal E1(t) through full-phase FFT, we can get the phase variation of the upper and lower sidebands. and And calculate the absolute distance L ab .

[0023] Compared with the prior art, the present invention has the following advantages:

[0024] First, the present invention uses a single detector to collect one interference signal in the optical path part of the ranging system that generates the interference signal, thereby avoiding the problem of unequal or non-ideal orthogonality of the two detectors in the traditional ranging system in the dual-detector orthogonal detection.

[0025] Second, compared to existing integrated systems for absolute distance and relative displacement measurement, this invention proposes a ranging system architecture that uses a cascaded intensity modulator (MZM) and phase modulator (PM) to generate an electro-optical frequency comb. This architecture then achieves both absolute distance and relative displacement measurement by retaining the positive and negative first-order sidebands. This effectively simplifies the ranging system structure and reduces construction costs. Furthermore, the relative displacement measurement accuracy of this invention is higher than that of absolute distance measurement, making it more practical for applications involving integrated absolute and relative ranging.

[0026] Third, the present invention generates two pairs of orthogonal components by mixing the interference signal with the carrier signal and the fundamental and second harmonics of the sinusoidal carrier signal. The interference signal phase, which contains relative displacement information, is extracted using the PGC-Atan demodulation method. The instantaneous phase calculation in the phase demodulation algorithm does not rely on subsequent sampling points, improving the real-time performance of displacement measurement.

[0027] Fourth, the present invention constructs a complex signal using two pairs of orthogonal components. It then uses a full-phase FFT to perform phase discrimination on the complex signal, extracting the phase of the interference signal containing absolute distance information. Compared to direct spectral analysis of real signals, which can result in a double-sided spectrum, the present invention uses complex signal spectral analysis to effectively obtain a single-sided spectrum, reducing the risk of spectral aliasing and improving phase discrimination accuracy, thereby significantly enhancing the system's absolute distance measurement accuracy. BRIEF DESCRIPTION OF THE DRAWINGS

[0028] Figure 1 It is a structural diagram of the double-sideband FSI system based on MZM-PM cascade modulation in the present invention;

[0029] Figure 2 It is a flowchart of the algorithm for distance measurement and relative displacement measurement in the present invention;

[0030] Figure 3 1 is a diagram showing simulation results of relative displacement measurement using the method of the present invention;

[0031] Figure 4This is a simulation result diagram of absolute distance measurement using the method of the present invention. DETAILED DESCRIPTION

[0032] The present invention will be further described below with reference to the accompanying drawings.

[0033] Example 1: Reference Figure 1 This embodiment further describes the specific structure of the double-sideband FSI ranging system based on MZM-PM cascade modulation proposed in the present invention as follows:

[0034] The FSI ranging system in the present invention is as follows: Figure 1 As shown; specifically includes a frequency-stabilized laser, a Mach-Zehnder modulator MZM, a fiber beam splitter FS, a phase modulator PM, an erbium-doped fiber amplifier EDFA, a circulator, a collimator COL, a corner cube prism, a fiber combiner FC, a photodetector PD and a data acquisition device;

[0035] The frequency-stabilized laser is used to generate an optical signal and output it to a Mach-Zehnder modulator (MZM). The MZM modulates the intensity of the light beam passing through it and outputs it to a fiber beam splitter, which splits the light to obtain a first light beam and a second light beam. The two decomposed light beams are respectively output to a phase modulator (PM) and an erbium-doped fiber amplifier (EDFA). The PM performs phase modulation on the light beam passing through it and outputs it to a fiber combiner (FC). The EDFA amplifies the power of the light signal passing through it and outputs it to a circulator. The circulator outputs the light signal to a collimator (COL), where it is emitted, reflected by a corner cube, received by the COL, and then transmitted back to the circulator. The circulator outputs the laser echo to the fiber combiner (FC). The two light beams converge again in the FC, interfere with each other, and are output to a photodetector, which detects and converts the interference signal into an interference signal through a data acquisition device.

[0036] Example 2: Reference Figure 1-2 The present invention proposes a double-sideband FSI ranging method based on MZM-PM cascade modulation, which is implemented by a double-sideband FSI ranging system based on cascade modulation of intensity modulator and phase modulator. Figure 1 The system includes a frequency-stabilized laser, a Mach-Zehnder modulator (MZM), a fiber beam splitter (FS), a phase modulator (PM), an erbium-doped fiber amplifier (EDFA), a circulator, a collimator (COL), a corner cube prism, a fiber beam combiner (FC), a photodetector (PD), and a data acquisition device. A method for distance measurement using the system specifically includes the following steps:

[0037] Step 1. The stabilized laser generates an optical signal S L (t) and output to the Mach-Zehnder modulator MZM; in this embodiment, the optical signal output by the laser is expressed as:

[0038] SL (t) = Acos(2πf L t),

[0039] Where A is the optical signal S L (t) AC amplitude, f L The center frequency of the laser emitted by a narrow linewidth laser.

[0040] Step 2. MZM realizes the optical signal S L (t) double-sideband linear frequency modulation, optical signal S L (t) After modulation, the optical signal S containing upper and lower sidebands is obtained. DSB (t):

[0041] S DSB (t) = a dsb [cos(2π(f L +f0)t+πKt 2 )+cos(2π(f L -f0)t-πKt 2 )],

[0042] Among them, a dsb is the power of the modulated optical signal.

[0043] Step 3. Optical signal S DSB (t) The optical signal L1 and L2 are split into two optical signals L1 and L2 by the fiber optic splitter FS, where L1 is directed to the phase modulator PM and L2 is directed to the circulator via the erbium-doped fiber amplifier EDFA; the PM phase modulates L1 to obtain the reference optical signal S RS (t); EDFA amplifies the power of L2.

[0044] The reference optical signal S RS (t), is obtained by the following steps:

[0045] (3.1) Apply a high-frequency carrier signal V to the PM PM (t):

[0046]

[0047] Among them, α represents the amplification factor of the high-voltage amplifier, V0 and ω C is the modulated signal The amplitude and angular frequency of

[0048] (3.2)PM modulates the phase of L1, causing the beam to shift by The phase shift of the phase modulated reference optical signal S is obtained RS (t):

[0049]

[0050] Among them, V π is the half-wave voltage of the phase modulator, m=παV0 / V π is the phase modulation depth.

[0051] Step 4. The power-amplified L2 is used as the measurement light signal and is emitted through the collimator COL. After reaching the corner cube prism, it is reflected and received by the collimator to obtain the laser echo signal S. MS (t):

[0052]

[0053] Step 5. Transform the reference optical signal S RS (t) and laser echo signal S MS (t) The optical fibers are reconverged at the fiber combiner FC and interfere with each other. Finally, the interference signal I is obtained by the photodetector PD and the data acquisition device. M (t):

[0054]

[0055] Where D represents the DC offset; A represents the optical signal S L (t) AC amplitude; represents the phase shift of the light beam L1 after passing through the phase modulator; For S RS (t) and S MS The upper sideband phase obtained by the upper sideband interference of (t) is For S RS (t) and S MS (t) The lower sideband phase obtained by the lower sideband interferometry; f L is the center frequency of the narrow linewidth laser; f0 is the initial frequency of the sweep signal source, K is the sweep rate of the sweep signal source, and π is the circumference of the circle; τ(t)=(L OPD +2νt)n / c is S RS (t) and S MS (t) is the time delay corresponding to the optical path difference, v represents the speed of the object, n represents the refractive index, and c is the speed of light; L OPD =2L means S RS (t) and S MS (t), L represents the optical path difference between the two, and t represents the signal duration;

[0056] Step 6. Using the interference signal I M (t) and the fundamental wave cos(ω c t) and the second harmonic cos(2ω ct) Calculate the quadrature component S1(t) and the in-phase component S2(t):

[0057]

[0058] Wherein, J1(m) and J2(m) are first-order and second-order Bessel functions of the first kind, respectively.

[0059] Step 7. Use S1(t) and S2(t) to get the interference signal I M (t) Upper and lower sideband phase and The sum of And calculate the phase change and the relative displacement L of the object re :

[0060] (7.1) Using S1(t) and S2(t), and setting the modulation depth, we can get the phase demodulation result. The implementation is as follows:

[0061] (7.1.1) The output signal S of the demodulation scheme is obtained according to the following formula out :

[0062]

[0063] (7.1.2) Set the modulation depth m to minimize the change in the value of J1(m) / J2(m), that is, set J1(m) / J2(m) = 1, and obtain the demodulation result

[0064]

[0065] (7.2) Utilization Calculate the phase change at different instantaneous values And calculate the relative displacement L of the object re , implemented as follows:

[0066] (7.2.1) Yes The instantaneous phase values ​​corresponding to times t1 and t2 and Do the difference and get the phase change

[0067]

[0068] in, and They represent the phase changes of the upper and lower sidebands from time t1 to t2 respectively.

[0069] (7.2.2) Using phase change Calculate the relative displacement L of the objectre :

[0070]

[0071] Step 8. Use S1(t) and S2(t) and obtain the absolute distance L through full-phase FFT ab The measurement steps are as follows:

[0072] (8.1) Using S1(t) and S2(t), construct the complex signal E1(t):

[0073]

[0074] Where j represents the imaginary unit in the complex number, J1(m) and J2(m) are the first-order and second-order Bessel functions of the first kind, respectively.

[0075] (8.2) Using the complex signal E1(t) through full-phase FFT, we can get the phase variation of the upper and lower sidebands. and And calculate the absolute distance L ab :

[0076] (8.2.1) Using the complex signal E1(t) and the full-phase FFT, we can obtain the phase variation of the upper and lower sidebands. and The implementation is as follows:

[0077] (8.2.1.1) The convolution window W of length 2N-1 conv The weighted signal E is obtained by weighting the complex signal consisting of the (N-1)·i+1th sample point of the complex signal E1(t) and its first N-1 sample points and the last N-1 sample points, a total of 2N-1 sample points. 1w (i), which is expressed as follows:

[0078] E 1w (i) = W conv ·[E1((N-1)·(i-1)+1),…,E1((N-1)·(i+1)+1)],

[0079] Where N is the number of sampling points of the full-phase FFT algorithm, i = 1, 2, …, T·Fs / (N-1), T is a frequency sweep period, and Fs is the sampling rate of the analog-to-digital converter;

[0080] (8.2.1.2) After weighting, the signal E 1w In (i), the sampling points with an interval of N are added in pairs to obtain the preprocessed complex signal E1(i):

[0081] E1(i)=E 1w (i)+E 1w(i+N);

[0082] (8.2.1.3) Perform a fast Fourier transform on the complex signal E1(i) to obtain its full phase spectrum. From the full phase spectrum of E1(i), obtain the instantaneous phases of the upper and lower sidebands corresponding to the two spectral peaks. and

[0083] (8.2.1.4) Yes and The instantaneous phase value corresponding to the starting time t1 and the ending time t2 within time T and Do the difference and get the phase change and

[0084] (8.2.2) Calculate the absolute distance L according to the following formula ab (i)

[0085]

[0086] Example 3: The overall implementation steps of the ranging method proposed in this embodiment are the same as those in Example 2. Figure 2 The acquisition of the interference signal in the present invention is further described as follows:

[0087] The phase modulator modulates the phase of the optical signal passing through, causing the optical signal to shift by The phase shift of the reference optical signal S after phase modulation is obtained. RS (t):

[0088]

[0089]

[0090] Among them, V π is the half-wave voltage of the phase modulator, m=παV0 / V π is the phase modulation depth.

[0091] Expanding the above formula, we can get:

[0092]

[0093] The part pointing to the circulator is used for distance measurement, thereby obtaining a laser echo signal S containing distance measurement information. MS (t):

[0094]

[0095] Where, τ(t)=(L OPD +2νt)n / c is the reference optical signal SRS (t) and the measured optical signal S MS (t) is the time delay corresponding to the optical path difference, v represents the speed of the object, n represents the refractive index, and c is the speed of light; L OPD =2L represents the reference optical signal S RS (t) and the measured light signal (laser echo signal) S MS (t) is the optical path difference between the two, and L is the distance measurement value.

[0096] The reference optical signal S output by the phase modulator RS (t) and the circulator output S MS (t) The optical fibers are reconverged at the fiber combiner FC and interfere with each other. Finally, the interference signal I is obtained by the photodetector and the analog-to-digital converter. M (t):

[0097]

[0098] Where D is the interference signal I M DC offset of (t); is the reference optical signal S RS (t) and the upper sideband of the measured optical signal S MS The interference signal I obtained by the upper sideband interference of M The upper sideband phase of (t), is the reference optical signal S RS (t) and the lower sideband of the measured optical signal S MS The interference signal I obtained by interfering the lower sideband of M (t) is the lower sideband phase. Expanding the above equation yields:

[0099]

[0100] Example 4: The overall implementation steps of the ranging method proposed in this example are the same as those in Example 2. Figure 2 The present invention utilizes S1(t) and S2(t) and realizes the absolute distance L by full phase FFT (APFFT) ab The measurement process is described in further detail as follows:

[0101] Set the modulation depth m to minimize the change in the value of J1(m) / J2(m), that is, set J1(m) / J2(m) = 1; construct the complex signal E1(t):

[0102]

[0103] Here, j represents the imaginary unit in the complex number.

[0104] Use the complex signal E1(t) to obtain the instantaneous phase of the upper and lower sidebands through APFFT and Here are the steps:

[0105] The convolution window W of length 2N-1 conv The weighted signal E is obtained by weighting the complex signal consisting of the (N-1)·i+1th sample point of the complex signal E1(t) and its first N-1 sample points and the last N-1 sample points, a total of 2N-1 sample points. 1w (i), which is expressed as follows:

[0106] E 1w (i) = W conv ·[E1((N-1)·(i-1)+1),…,E1((N-1)·(i+1)+1)],

[0107] Where N is the number of sampling points of the full-phase FFT algorithm, i = 1, 2, …, T·Fs / (N-1), T is a frequency sweep period, and Fs is the sampling rate of the analog-to-digital converter;

[0108] After weighting, the signal E 1w In (i), the sampling points with an interval of N are added in pairs to obtain the preprocessed complex signal E1(i):

[0109] E1(i)=E 1w (i)+E 1w (i+N);

[0110] Perform fast Fourier transform on the complex signal E1(i) to obtain its full phase spectrum. From the full phase spectrum of E1(i), the instantaneous phases of the upper and lower sidebands corresponding to the two peaks are obtained. and The implementation is as follows:

[0111] From the full phase spectrum of E1(i), the discrete phase information of the upper and lower sidebands corresponding to the two peaks is obtained. and

[0112] Perform phase unwrapping on the discrete phase information to obtain the instantaneous phase corresponding to the two peaks in the full phase spectrum of E1(i) and

[0113] Discrete phase information As an example, perform phase unwrapping to obtain the absolute phase sequence P1 of each sampling point starting from the phase value of the initial sampling point + (i), P1 - (i) Similarly:

[0114]

[0115] Instantaneous phase and It is expressed as follows:

[0116]

[0117] right and The instantaneous phase value corresponding to the starting time t1 and the ending time t2 within time T and Do the difference and get the phase change and

[0118]

[0119] The absolute distance L is calculated according to the following formula ab :

[0120]

[0121] The effects of the present invention can be further illustrated by the following simulation results.

[0122] 1. Simulation conditions

[0123] Using MATLAB simulation software, the wavelength of the laser is set to 1550nm, the sweep frequency range of the sweep frequency signal source is 5GHZ to 10GHZ, the sweep frequency signal source has a scanning period of 1ms, the sampling rate of the analog-to-digital converter AD is 100MHZ, the signal-to-noise ratio is 40dB, the number of sampling points of the full-phase FFT algorithm is set to 512, the absolute distance of the object to be measured is 10m, and it vibrates in a sinusoidal form with an amplitude of 10 nanometers and a frequency of 2000HZ. The sampling resolution is 10bit, and the simulation time is 0.5s.

[0124] 2. Simulation content

[0125] Simulation 1: Under the above simulation conditions, the relative displacement measurement accuracy diagram is realized by the method of the present invention. The results are as follows: Figure 3 , where the horizontal axis represents the simulation time and the vertical axis represents the ranging accuracy. Figure 3 It can be seen that the relative displacement measurement accuracy achieved by the method of the present invention is 0.53 nanometers.

[0126] Simulation 2: Under the above simulation conditions, the absolute distance measurement accuracy diagram is achieved using the method of the present invention. The results are as follows: Figure 4 , where the horizontal axis represents the simulation time and the vertical axis represents the ranging accuracy. Figure 4 It can be seen that the method of the present invention achieves an absolute distance measurement standard deviation of 7.60 microns.

[0127] The above simulation analysis proves the correctness and effectiveness of the method proposed in the present invention.

[0128] Parts of the present invention that are not described in detail belong to common knowledge among those skilled in the art.

[0129] The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Obviously, for professionals in this field, after understanding the content and principles of the present invention, they may make various modifications and changes in form and details without departing from the principles and structure of the present invention. However, these modifications and changes based on the ideas of the present invention are still within the scope of protection of the claims of the present invention.

Claims

1. A double-sideband FSI ranging method based on MZM-PM cascade modulation is implemented by a double-sideband FSI ranging system based on cascade modulation of an intensity modulator and a phase modulator, the system comprising a frequency-stabilized laser, a Mach-Zehnder modulator (MZM), a fiber beam splitter (FS), a phase modulator (PM), an erbium-doped fiber amplifier (EDFA), a circulator, a collimator (COL), a corner cube, a fiber combiner (FC), a photodetector (PD), and a data acquisition device; characterized in that: The steps include: (1) The frequency-stabilized laser generates an optical signal S L (t) and output to the Mach-Zehnder modulator MZM; (2) MZM realizes the optical signal S L (t) double-sideband linear frequency modulation, optical signal S L (t) After modulation, the optical signal S containing upper and lower sidebands is obtained. DSB (t); (3) Optical signal S DSB (t) The optical signal L1 and L2 are split into two optical signals L1 and L2 by the fiber optic splitter FS, where L1 is directed to the phase modulator PM and L2 is directed to the circulator via the erbium-doped fiber amplifier EDFA; the PM phase modulates L1 to obtain the reference optical signal S RS (t); EDFA performs power amplification on L2; (4) The power-amplified L2 is used as the measurement light signal and is emitted through the collimator COL. After reaching the corner cube prism, it is reflected and received by the collimator to obtain the laser echo signal S MS (t); (5) The reference optical signal S RS (t) and laser echo signal S MS (t) The optical fibers are reconverged at the fiber combiner FC and interfere with each other. Finally, the interference signal I is obtained by the photodetector PD and the data acquisition device. M (t): Where D represents the DC offset; A represents the optical signal S L (t) AC amplitude; represents the phase shift of the light beam L1 after passing through the phase modulator; For S RS (t) and S MS The upper sideband phase obtained by the upper sideband interference of (t) is For S RS (t) and S MS (t) The lower sideband phase obtained by the lower sideband interferometry; f L is the center frequency of the narrow linewidth laser; f0 is the initial frequency of the sweep signal source, K is the sweep rate of the sweep signal source, and π is the circumference of the circle; τ(t)=(L OPD +2νt)n / c is S RS (t) and S MS (t) is the time delay corresponding to the optical path difference, v represents the speed of the object, n represents the refractive index, and c is the speed of light; L OPD =2L means S RS (t) and S MS (t), L represents the optical path difference between the two, and t represents the signal duration; (6) Using the interference signal I M (t) and the fundamental wave cos(ω c t) and the second harmonic cos(2ω c t) Calculate the quadrature component S1(t) and the in-phase component S2(t); (7) Using S1(t) and S2(t) to obtain the interference signal I M (t) Upper and lower sideband phase and The sum of And calculate the phase change and the relative displacement L of the object re : (8) Using S1(t) and S2(t), and using full-phase FFT to achieve the absolute distance L ab The measurement steps are as follows: (8.1) Using S1(t) and S2(t), construct the complex signal E1(t); (8.2) Using the complex signal E1(t) through full-phase FFT, we can get the phase variation of the upper and lower sidebands. and And calculate the absolute distance L ab .

2. The method according to claim 1, wherein: The optical signal S in step (1) L (t) and the optical signal S containing upper and lower sidebands in step (2) DSB (t), are represented as follows: S L (t)=Acos(2πf L t), S DSB (t)=a dsb [cos(2π(f L +f0)t+πKt 2 )+cos(2π(f L -f0)t-πKt 2 )], Among them, a dsb is the power of the modulated optical signal.

3. The method according to claim 1, wherein: The reference optical signal S in step (3) RS (t), is obtained by the following steps: (3.1) Apply a high-frequency carrier signal V to the PM PM (t): Among them, α represents the amplification factor of the high-voltage amplifier, V0 and ω C is the modulated signal The amplitude and angular frequency of (3.2)PM modulates the phase of L1, causing the beam to shift by The phase shift of the phase modulated reference optical signal S is obtained RS (t): Among them, V π is the half-wave voltage of the phase modulator, m=παV0 / V π is the phase modulation depth.

4. The method according to claim 1, wherein: The laser echo signal S in step (4) MS (t), is expressed as follows:

5. The method according to claim 1, wherein: The quadrature component S1(t) and the in-phase component S2(t) in step (6) are expressed as follows: Wherein, J1(m) and J2(m) are first-order and second-order Bessel functions of the first kind, respectively.

6. The method according to claim 5, characterized in that: Step (7) calculates the phase change and the relative displacement L of the object re , as follows: (7.1) Using S1(t) and S2(t), and setting the modulation depth, we can get the phase demodulation result. The implementation is as follows: (7.1.1) The output signal S of the demodulation scheme is obtained according to the following formula out : (7.1.2) Set the modulation depth m to minimize the change in the value of J1(m) / J2(m), that is, set J1(m) / J2(m) = 1, and obtain the demodulation result (7.2) Utilization Calculate the phase change at different instantaneous values And calculate the relative displacement L of the object re , implemented as follows: (7.2.1) Yes The instantaneous phase values ​​corresponding to times t1 and t2 and Do the difference and get the phase change in, and Respectively represent the phase changes of the upper and lower sidebands from time t1 to t2; (7.2.2) Using phase change Calculate the relative displacement L of the object re :

7. The method according to claim 1, wherein: Using S1(t) and S2(t) as described in step (8.1), the complex signal E1(t) is constructed as follows: Where j represents the imaginary unit in the complex number, J1(m) and J2(m) are the first-order and second-order Bessel functions of the first kind, respectively.

8. The method according to claim 1, wherein: The absolute distance L mentioned in step (8.2) ab , calculated according to the following steps: (8.2.1) Using the complex signal E1(t) and the full-phase FFT, we can obtain the phase variation of the upper and lower sidebands. and The implementation is as follows: (8.2.1.1) The convolution window W with a length of 2N-1 conv The weighted signal E is obtained by weighting the complex signal consisting of the (N-1)·i+1th sample point of the complex signal E1(t) and its first N-1 sample points and the last N-1 sample points, a total of 2N-1 sample points. 1w (i), which is expressed as follows: HAVE BEEN 1w (i)=W conv ·[E1((N-1)·(i-1)+1),…,E1((N-1)·(i+1)+1)], Where N is the number of sampling points of the full-phase FFT algorithm, i = 1, 2, …, T·Fs / (N-1), T is a frequency sweep period, and Fs is the sampling rate of the analog-to-digital converter; (8.2.1.2) After weighting, the signal E 1w In (i), the sampling points with an interval of N are added in pairs to obtain the preprocessed complex signal E1(i): E1(i)=E 1w (i)+E 1w (i+N); (8.2.1.3) Perform a fast Fourier transform on the complex signal E1(i) to obtain its full phase spectrum. From the full phase spectrum of E1(i), obtain the instantaneous phases of the upper and lower sidebands corresponding to the two spectral peaks. and (8.2.1.4) Yes and The instantaneous phase value corresponding to the starting time t1 and the ending time t2 within time T and Do the difference and get the phase change and (8.2.2) Calculate the absolute distance L according to the following formula ab (i) 9. A double-sideband FSI ranging system implemented according to the method of claim 1, characterized in that: include: A frequency-stabilized laser, a Mach-Zehnder modulator (MZM), a fiber beam splitter (FS), a phase modulator (PM), an erbium-doped fiber amplifier (EDFA), a circulator, a collimator (COL), a corner cube prism, a fiber beam combiner (FC), a photodetector (PD), and a data acquisition device. The frequency-stabilized laser generates an optical signal, which is then split into two output beams after passing through the MZM and FS. The first beam passes through the PM and enters the FC, while the second beam passes through the EDFA and enters the circulator. The second beam entering the circulator is sent to the COL and emitted by the COL, reaches the corner cube prism, is reflected, and is received by the COL to obtain a laser echo; the laser echo is returned to the circulator and output to the FC through the circulator; The FC is used to reconverge the interference of the two incoming light beams and output them to the PD, and the interference signal is obtained by conversion through the PD detection and data acquisition device.

10. The system according to claim 9, characterized in that: The Mach-Zehnder modulator (MZM) is used to modulate the intensity of the optical signal generated by the frequency-stabilized laser and output it to the fiber optic splitter (FS). The fiber optic splitter (FS) is used to split the optical signal to obtain a first light beam and a second light beam after decomposition. The phase modulator (PM) is used to phase-modulate the first light beam and output it to the fiber optic combiner (FC). The erbium-doped fiber amplifier (EDFA) is used to power-amplify the second light beam and output it to the circulator.

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