MMC type power electronic transformer control method and system for bidirectional active full-bridge parallel system

Abstract

The invention discloses an MMC type power electronic transformer control method and system for a bidirectional active full-bridge parallel system. The voltage difference between the modulation wave correction component of each MMC sub-module and the reference value of the negative sequence 2 times frequency component of the updated circulating current and the AC output voltage reference value is used as the modulation voltage of each MMC sub-module to obtain a trigger pulse. The difference between the secondary DC side voltage of the bidirectional active full-bridge and the corresponding reference value is used to generate the phase shift angle reference value of all the bidirectional active full-bridges based on a phase shift proportional loop. The difference between the DC input voltage of each bidirectional active full-bridge and the average value of the DC voltage of all the MMC sub-modules is used to generate the phase shift angle correction value of each bidirectional active full-bridge based on a phase shift proportional loop. The difference between the phase shift angle reference value and the phase shift angle correction value is used as the phase shift angle of each bidirectional active full-bridge. The invention realizes the capacitor voltage balance of each MMC sub-module and reduces the influence of the 2 times frequency circulating current.

Description

Technical Field

[0001] The present invention belongs to the technical field of power electronic transformer control technology, and in particular to an overall control method and system of an MMC type power electronic transformer of a bidirectional active full-bridge output parallel system. Background Art

[0002] With the modernization of power systems and the large-scale access to renewable energy, the application of power electronic transformers has gradually attracted attention. Traditional power transformers face problems of low efficiency, complex control and large size in high-voltage transmission and energy conversion, especially when they need to handle both AC and DC grids. To meet these challenges, the Modular Multilevel Converter (MMC) structure has become one of the main architectural choices for power electronic transformers (PETs) due to its high voltage control flexibility and reliability. However, the limitations of a single MMC topology in handling high-power and multi-port applications have prompted the introduction of a dual active bridge (DAB) output parallel system to optimize power transmission and improve system flexibility.

[0003] The combination of a bidirectional active full-bridge output parallel system and an MMC-type power electronic transformer enables PET to provide multifunctional outputs of four ports: high voltage AC, high voltage DC, low voltage AC, and low voltage DC. This topology not only simplifies the device structure, but also optimizes system performance through sub-module capacitor voltage balance control, circulating current suppression, and coordinated control of DAB output. This technology is particularly suitable for new energy power generation systems (such as photovoltaic power generation), distribution links in smart grids, and other scenarios that require high voltage and high power power conversion.

[0004] The MMC-type power electronic transformer of the bidirectional active full-bridge output parallel system in the prior art has many shortcomings in multi-port applications. First, the sub-module capacitor voltage is prone to imbalance, resulting in voltage deviation accumulation, which affects the power quality and stability of the system output. In addition, the circulating current in the MMC contains a negative-sequence 2-fold frequency component, which has a negative impact on the balance of the bridge arm voltage and system efficiency, and increases energy loss. The existing control strategy is complex and requires separate control and coordination of each DAB module, resulting in high system hardware costs and complex control algorithms. At the same time, the traditional topology has limited capabilities in power fault recovery and cannot achieve fast pre-charging, resulting in a long fault recovery time, affecting the reliability and continuous operation of the system. Summary of the invention

[0005] In order to solve the deficiencies in the prior art, the present invention provides a MMC type power electronic transformer control method and system for a bidirectional active full-bridge parallel system. According to the topological structure characteristics of the MMC type PET, the submodule capacitor voltage balance control is used to improve the balance of the submodule capacitor voltage, ensure the consistency of the voltage of each submodule during the multi-port output process, and enhance the system stability. By introducing a negative sequence circulating current suppression controller, the influence of the 2-fold frequency circulating current on the system output is reduced, and the voltage and current quality is improved.

[0006] The present invention adopts the following technical solution.

[0007] The present invention proposes an MMC type power electronic transformer control method of a bidirectional active full-bridge parallel system, wherein the MMC type power electronic transformer comprises a plurality of MMC sub-modules cascaded on the AC side and a plurality of bidirectional active full-bridges connected in parallel on the DC side, wherein the bidirectional active full-bridges are connected in parallel to the DC side capacitors of the MMC sub-modules, and comprises:

[0008] Obtain the actual capacitor voltage and rated capacitor voltage of each MMC submodule, and the current and circulating current of the upper and lower bridge arms of each phase of the MMC power electronic transformer;

[0009] Using the actual capacitor voltage, rated capacitor voltage and current, a modulation wave correction component of each MMC submodule is generated based on a voltage proportional control loop;

[0010] Using the difference between the negative-sequence double frequency component of the circulating current and the corresponding reference value, a compensation voltage is generated based on the PI loop, and the compensation voltage compensates for the voltage drop caused by the negative-sequence double frequency component of the circulating current to update the reference value of the negative-sequence double frequency component of the circulating current;

[0011] The voltage control outer loop of each MMC submodule outputs the AC output voltage reference value of each MMC submodule; the voltage difference between the modulation wave correction component of each MMC submodule and the reference value of the negative sequence double frequency component of the updated circulating current and the AC output voltage reference value is used as the modulation voltage of each MMC submodule, and the trigger pulse of the power device is obtained according to the modulation voltage of each MMC submodule;

[0012] The phase shift angle reference value of all bidirectional active full bridges is generated based on the phase shift proportional loop by utilizing the difference between the secondary DC side voltage of the bidirectional active full bridge and the corresponding reference value; the phase shift angle correction value of each bidirectional active full bridge is generated based on the phase shift proportional loop by utilizing the difference between the DC input voltage of each bidirectional active full bridge and the average value of the DC voltage of all MMC sub-modules, and the difference between the phase shift angle reference value and the phase shift angle correction value is used as the phase shift angle of each bidirectional active full bridge.

[0013] Preferably, the modulation wave correction component of the MMC submodule in the upper bridge arm of each phase of the MMC type power electronic transformer and the modulation wave correction component of the MMC submodule in the lower bridge arm respectively satisfy the following relationship:

[0014]

[0015] In the formula, , They are The first The modulation wave correction component of each MMC submodule, is the proportional coefficient of the voltage proportional control loop, is the rated capacitance voltage of the MMC submodule, , They are The first The actual capacitor voltage of each MMC submodule, , They are The upper arm current and lower arm current of the phase, where , is the total number of MMC submodules, , indicating different phases.

[0016] Preferably, the even harmonic components in the circulating current include: a 6k+2 harmonic component in a negative sequence distribution, a 6k+4 harmonic component in a positive sequence distribution, and a 6k+6 harmonic component in a zero sequence distribution, wherein k is a non-negative integer;

[0017] When k is 0, the AC component in the second harmonic component with negative sequence distribution serves as the negative sequence double frequency component of the circulating current.

[0018] Preferably, in the stationary coordinate system, the negative sequence 2 times frequency component of the circulating current of the upper and lower bridge arms of each phase is , , , which is converted into the negative sequence 2-fold frequency component of the circulating current in the dq coordinate system by the coordinate conversion unit , ;

[0019] Reference value of d-axis component of negative sequence 2 times frequency component of circulating current The d-axis component of the negative sequence 2nd frequency component of the circulating current The difference between them is obtained by the PI regulator to obtain the d-axis component of the compensation voltage, and the q-axis component of the negative-sequence double frequency component of the circulating current. Inductance in the bridge arm The q-axis component of the voltage drop caused by the upper current is compensated; the q-axis component reference value of the negative sequence 2 times frequency component of the circulating current The q-axis component of the negative-sequence 2nd frequency component of the circulating current The difference between them is obtained by the PI regulator to obtain the q-axis component of the compensation voltage, and the d-axis component of the negative-sequence double frequency component of the circulating current Inductance in the bridge arm The voltage drop d-axis component caused by the upper side is compensated to satisfy the following relationship:

[0020]

[0021] In the formula, is the voltage vector after the voltage drop caused by the negative sequence double frequency component of the circulating current in the dq coordinate system is compensated, , is the voltage after the voltage drop caused by the negative sequence 2 times frequency component of the circulating current in the dq coordinate system is compensated, , is the negative sequence 2-fold frequency component of the circulating current in the dq coordinate system, is the system angular frequency, , They are The resistance and inductance of the phase, , indicating different phases;

[0022] The voltage d-axis component after the voltage drop caused by the negative sequence 2 times frequency component of the circulating current is compensated and the q-axis component The reference value of the negative sequence 2 times frequency component of the updated circulating current in the upper and lower bridge arms of the three-phase stationary coordinate system is obtained by the coordinate conversion unit. , , .

[0023] Preferably, the modulation voltage of each MMC sub-module in the upper bridge arm and the lower bridge arm of each phase generates a trigger pulse of a power device in the MMC sub-module through a CPS-PWM modulator.

[0024] Preferably, the voltage control outer loop of each MMC submodule includes: a positive sequence current decoupling control loop; a rated capacitor voltage of each MMC submodule The average value of the actual capacitor voltage of all MMC submodules The input current obtained by the PI loop after the difference between , q-axis component , q-axis component reference value As the input signal of the positive sequence current decoupling control loop, the potential difference d-axis component output by the positive sequence current decoupling control loop is related to the d-axis component of the system three-phase potential The sum of the potential difference of the positive sequence current decoupling control loop output and the q-axis component of the system three-phase potential The sum of the AC output voltage reference values ​​of all MMC sub-modules is obtained through coordinate transformation. , , ; Set the AC output voltage reference value of all MMC submodules , , According to the total number of MMC submodules Evenly distribute to each MMC submodule to obtain the AC output voltage reference value of each MMC submodule , , .

[0025] Preferably, in the positive sequence current decoupling control loop, the d-axis component of the input current and the system three-phase current The difference between the two is passed through the PI loop to obtain the d-axis component of the intermediate potential difference and the q-axis component of the system three-phase current. Equivalent reactance of the system The product of is used to compensate the d-axis component of the intermediate potential difference, and the d-axis component of the potential difference output by the positive sequence current decoupling control loop is obtained; the q-axis component reference value of the system three-phase current is With q-axis component The difference between the two is passed through the PI loop to obtain the q-axis component of the intermediate potential difference and the d-axis component of the system three-phase current. Equivalent reactance of the system The product of is used to compensate the q-axis component of the intermediate potential difference to obtain the q-axis component of the potential difference output by the positive-sequence current decoupling control loop.

[0026] Preferably, the transfer function of the phase-shift proportional loop is ,in, and are the coefficients of the phase-shift proportional loop, is the Laplace operator.

[0027] Preferably, the shift ratio of the bidirectional active full bridge is calculated using the following relationship:

[0028]

[0029] In the formula, For the shift ratio of the bidirectional active full bridge, The ratio of the medium frequency transformer of the bidirectional active full bridge is , They are the primary DC side voltage and the secondary DC side voltage of DAB respectively;

[0030] The phase shift angle of each bidirectional active full bridge is limited by the following relationship to control the bidirectional active full bridge to operate in the soft switching mode:

[0031]

[0032] In the formula, is the phase shift angle of the bidirectional active full bridge.

[0033] The present invention also proposes an MMC type power electronic transformer control system of a bidirectional active full-bridge parallel system, comprising:

[0034] The acquisition module is used to obtain the actual capacitor voltage and rated capacitor voltage of each MMC submodule, and the current and circulating current of the upper and lower bridge arms of each phase of the MMC power electronic transformer;

[0035] The MMC submodule capacitor voltage control module is used to generate a modulation wave correction component of each MMC submodule based on a voltage proportional control loop using the actual capacitor voltage, the rated capacitor voltage and the current;

[0036] The MMC submodule negative-sequence 2 times frequency circulating current suppression module is used to generate a compensation voltage based on the PI loop using the difference between the negative-sequence 2 times frequency component of the circulating current and the corresponding reference value. The compensation voltage compensates for the voltage drop caused by the negative-sequence 2 times frequency component of the circulating current to update the reference value of the negative-sequence 2 times frequency component of the circulating current;

[0037] The MMC submodule modulation voltage module is used for the voltage control outer loop of each MMC submodule to output the AC output voltage reference value of each MMC submodule; the voltage difference between the modulation wave correction component of each MMC submodule and the reference value of the negative sequence 2 times frequency component of the updated circulating current and the AC output voltage reference value is used as the modulation voltage of each MMC submodule;

[0038] MMC submodule trigger pulse module, used to obtain the trigger pulse of the power device according to the modulation voltage of each MMC submodule;

[0039] The bidirectional active full-bridge output parallel system control module is used to generate the phase shift angle reference value of all bidirectional active full bridges based on the phase shift proportional loop by using the difference between the secondary DC side voltage of the bidirectional active full bridge and the corresponding reference value; and to generate the phase shift angle correction value of each bidirectional active full bridge based on the phase shift proportional loop by using the difference between the DC input voltage of each bidirectional active full bridge and the average value of the DC voltage of all MMC sub-modules, and to use the difference between the phase shift angle reference value and the phase shift angle correction value as the phase shift angle of each bidirectional active full bridge.

[0040] The beneficial effect of the present invention is that, compared with the prior art, at least the method proposed by the present invention improves the balance of the submodule capacitor voltage, ensures the consistency of the voltage of each submodule during the multi-port output process, and enhances the stability of the system. In addition, by introducing a negative sequence circulating current suppression controller, the impact of the 2-fold frequency circulating current on the system output is reduced, and the voltage and current quality are improved. The optimization of the overall control strategy simplifies the coordination requirements for multiple DAB modules, reduces hardware costs and algorithm complexity, and realizes the MMC-type PET overall control of the submodule capacitor voltage balance in the bidirectional active full-bridge output parallel system, providing support for the simplification of hardware and software of high-voltage and large-capacity power electronic transformers with a large number of devices.

[0041] The pre-charging scheme proposed in the present invention can quickly restore the DC bus voltage after a fault, improve the fault ride-through capability, enable the system to achieve rapid recovery during power fluctuations, and ensure stable operation. BRIEF DESCRIPTION OF THE DRAWINGS

[0042] Figure 1 1 is a basic main circuit topology diagram of the MMC type power electronic transformer in the embodiment of the present invention;

[0043] Figure 2 Schematic diagram of the MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system proposed by the present invention;

[0044] Figure 3 It is a schematic diagram of the MMC submodule capacitor voltage control logic proposed by the present invention;

[0045] Figure 4 It is a logic diagram of negative sequence 2-fold frequency circulating current suppression of the MMC submodule proposed by the present invention;

[0046] Figure 5 It is a schematic diagram of the logic of generating modulation voltage of the MMC submodule proposed in the present invention;

[0047] Figure 6 is a single-phase equivalent circuit diagram of the MMC submodule in an embodiment of the present invention;

[0048] Figure 7 It is a logic schematic diagram of the positive sequence current decoupling control loop before the input signal is improved in the present invention;

[0049] Figure 8 It is a control logic diagram of the bidirectional active full-bridge output parallel system proposed by the present invention;

[0050] Fig. 9 is a schematic diagram of the single phase shift control logic of DAB in an embodiment of the present invention;

[0051] Fig.10 It is a structural diagram of an MMC type power electronic transformer control system of a bidirectional active full-bridge output parallel system in an embodiment of the present invention. DETAILED DESCRIPTION

[0052] In order to make the purpose, technical scheme and advantages of the present invention clearer, the technical scheme of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. The embodiments described in this application are only embodiments of a part of the present invention, rather than all embodiments. Based on the spirit of the present invention, all other embodiments obtained by ordinary technicians in this field without making creative work belong to the protection scope of the present invention.

[0053] like Figure 1 As shown, each phase of the AC side of the MMC power electronic transformer includes an upper bridge arm and a lower bridge arm, and the low-voltage DC side of the MMC power electronic transformer is configured with a low-voltage converter. The upper bridge arm and the lower bridge arm of each phase include multiple units with the same structure, and each unit includes an MMC sub-module and a bidirectional active full-bridge DAB, wherein the AC side of the MMC sub-module is cascaded, and the bidirectional active full-bridge DAB is connected in parallel to the DC side capacitor of the MMC sub-module, and the DC sides of each bidirectional active full-bridge DAB are connected in parallel to form a bidirectional active full-bridge output parallel system; the cascaded MMC sub-modules in the upper bridge arm and the lower bridge arm provide a high-voltage AC port and a high-voltage DC port, and provide a low-voltage DC port for the low-voltage converter; DAB is used to transfer energy between the high-voltage DC port and the low-voltage DC port through the DC side capacitor of the MMC sub-module; the low-voltage converter is used to provide a low-voltage AC port and transfer energy between the low-voltage AC port and the low-voltage DC port. As shown Figure 1 As shown, the MMC type PET includes four ports of different voltage levels and forms: high voltage AC, high voltage DC, low voltage AC and low voltage DC.

[0054] When it comes to distribution network applications, the number of sub-modules in the MMC generally ranges from a dozen to dozens. Carrier phase-shift pulse width modulation is selected as the modulation strategy to generate a pulse signal for each sub-module based on the modulation signal.

[0055] The present invention proposes a MMC type power electronic transformer control method for a bidirectional active full-bridge parallel system. Figure 2 As shown, including:

[0056] Step 1: Obtain the actual capacitor voltage and rated capacitor voltage of each MMC submodule , and obtain the upper bridge arm current and the lower bridge arm current of each phase of the MMC type power electronic transformer; based on the voltage proportional control loop, generate the modulation wave correction component of each MMC sub-module in the upper bridge arm and the lower bridge arm of each phase .

[0057] The present invention proposes that the MMC submodule uses voltage proportional control to balance the capacitor voltage. The parameters of the voltage closed-loop control loop include: proportional coefficient The voltage closed-loop control loop uses the difference between the rated capacitor voltage of the MMC submodule and the actual capacitor voltage of the MMC submodule as the input signal. The input signal and the proportional coefficient The product of the bridge arm current constitutes the modulation wave correction component of the MMC submodule; the modulation wave correction component of each MMC submodule in the upper bridge arm and the lower bridge arm of each phase , including: the modulation wave correction component of the MMC submodule in the upper bridge arm of each phase, and the modulation wave correction component of the MMC submodule in the lower bridge arm of each phase, which respectively satisfy the following relationship:

[0058]

[0059] In the formula, , They are The first The modulation wave correction component of each MMC sub-module, in the embodiment, since the upper bridge arm current and the lower bridge arm current are in opposite directions, The first Modulation wave correction component of each MMC submodule Introduce the coefficient "-1" to achieve the same The first Modulation wave correction component of each MMC submodule The values ​​are the same, is the proportional coefficient of the voltage proportional control loop, is the rated capacitance voltage of the MMC submodule. The rated capacitance voltages of the MMC submodules in the embodiment are the same. , They are The first The actual capacitor voltage of each MMC submodule, , They are The upper arm current and lower arm current of the phase, where , is the total number of MMC submodules, , indicating different phases.

[0060] In the embodiment, the voltage proportional control loop structure of the MMC submodule is as follows: Figure 3 As shown in the figure, the voltage proportional control loop uses the feedback value of the actual capacitor voltage of each MMC sub-module to achieve closed-loop control of the capacitor voltage. Compared with the existing open-loop control or simple hierarchical control, it can respond quickly according to the change of the capacitor voltage and dynamically adjust the output voltage of the sub-module. The closed-loop control achieves more accurate voltage balance through real-time feedback, so that the sub-module capacitor voltage can still maintain balance under multi-port output and complex load conditions. In addition, the adjustment speed of the capacitor voltage is directly adjusted by adjusting the proportional coefficient This can significantly improve the power quality of the system and avoid fluctuations and power losses caused by voltage imbalance.

[0061] Step 2, obtain the circulating current of the upper bridge arm and the circulating current of the lower bridge arm of each phase of the MMC type power electronic transformer; generate a compensation voltage based on the PI loop using the reference value of the negative sequence 2 times frequency component of the circulating current and the difference between the negative sequence 2 times frequency component of the circulating current, and use the compensation voltage to compensate for the voltage drop caused by the negative sequence 2 times frequency component of the circulating current, so as to update the reference value of the negative sequence 2 times frequency component of the circulating current in the upper and lower bridge arms of each phase .

[0062] The circulating current in the MMC submodule is the carrier of energy flowing inside the MMC submodule. If the circulating current contains a non-zero AC component, a voltage drop caused by the circulating current will be generated on the bridge arm inductance, thereby affecting the output voltage.

[0063] Assuming that the three-phase MMC submodule is symmetrical, the DC side current is evenly distributed among the three phases. The upper arm current and lower arm current of each phase satisfy the following relationship:

[0064]

[0065] In the formula, is the DC current, is the amplitude of the fundamental current, is the system angular frequency, for The power angle of the phase, is the initial phase of the fundamental current, For the circulation The amplitude of the subharmonic components, For the circulation The initial phase of the subharmonic components, is an even number;

[0066] The circulating current of the three-phase MMC submodule contains only even-order harmonic components, and as the harmonic order increases, its amplitude decreases accordingly. The even-order harmonic components in the circulating current are:

[0067] 1) 6k+2 harmonic components with negative sequence distribution;

[0068] 2) 6k+4 harmonic components with positive sequence distribution;

[0069] 3) 6k+6th harmonic component with zero-sequence distribution;

[0070] Wherein, k is a non-negative integer.

[0071] When k is 0, the 2nd harmonic component with negative sequence distribution is used as the 2nd frequency component of the circulating current. The 2nd frequency component is the main harmonic in the circulating current, so the controller is designed with the suppression of the 2nd frequency component as the circulating current control target.

[0072] When only the 2-fold frequency component is considered and other components are ignored, the circulating current of the three-phase MMC submodule satisfies the following relationship:

[0073]

[0074] In the formula, , , is the double frequency component of the three-phase circulating current of the MMC submodule, is the amplitude of the 2-fold frequency component current, is the initial phase of the 2-fold frequency component current;

[0075] Each phase circulating current contains a DC component and an AC component, wherein the AC component is the negative sequence 2 times frequency component of the circulating current that needs to be suppressed.

[0076] The voltage drop caused by the negative-sequence 2 times frequency component of the circulating current on the bridge arm inductance is the direct cause of the bridge arm voltage imbalance. According to the three-phase MMC equivalent circuit, the voltage drop caused by the negative-sequence 2 times frequency component of the circulating current in the bridge arm of each phase satisfies the following relationship:

[0077]

[0078] In the formula, for The voltage drop caused by the negative sequence 2 times frequency component of the circulating current in the bridge arm of the phase, , They are The resistance and inductance of the phase, For MMC submodule The negative sequence 2-fold frequency component of the phase circulating current, is the DC voltage, , They are The upper bridge arm voltage and the lower bridge arm voltage of the phase;

[0079] In the stationary coordinate system, in the bridge arm of each phase, the voltage vector after the voltage drop caused by the negative sequence double frequency component of the circulating current is compensated satisfies the following relationship:

[0080]

[0081] In the formula, is the voltage vector after the voltage drop caused by the negative sequence double frequency component of the circulating current in the three-phase bridge arm in the stationary coordinate system is compensated, , , is the voltage after the voltage drop caused by the negative sequence double frequency component of the circulating current in the three-phase bridge arm in the stationary coordinate system is compensated, , , is the negative sequence 2 times frequency component of the circulating current in the three-phase bridge arm in the stationary coordinate system;

[0082] In the dq coordinate system, the voltage drop caused by the negative sequence 2 times frequency component of the circulating current satisfies the following relationship:

[0083]

[0084] In the formula, is the voltage vector after the voltage drop caused by the negative sequence double frequency component of the circulating current in the dq coordinate system is compensated, , is the voltage after the voltage drop caused by the negative sequence 2 times frequency component of the circulating current in the dq coordinate system is compensated, , is the negative sequence 2-fold frequency component of the circulating current in the dq coordinate system, is the system angular frequency, , They are The resistance and inductance of the phase, , indicating different phases;

[0085] The circulating current suppression control strategy proposed in the present invention is mainly aimed at the negative sequence double frequency circulating current in the MMC system, which is the main harmonic component affecting the system stability and output voltage. The traditional circulating current suppression method usually roughly controls the entire circulating current and cannot finely suppress the harmonics of a specific frequency. Through a special double frequency component controller, the harmonics of this frequency are suppressed in a targeted manner, thereby more efficiently reducing the adverse effects on system performance.

[0086] The circulating current suppression control strategy proposed in the present invention converts the circulating current into a stationary coordinate system (dq coordinate system) for processing, which is convenient for separating different harmonic components, especially the negative sequence component of 2 times the frequency. This design based on the dq coordinate system can accurately calculate and control the circulating current components, while the traditional method usually directly suppresses in a stationary coordinate system, resulting in low control accuracy and poor suppression effect.

[0087] The negative sequence double frequency circulating current suppression topology proposed by the present invention is as follows: Figure 4 As shown, including:

[0088] Negative sequence 2-fold frequency component of circulating current in three-phase bridge arm in stationary coordinate system , , , through the first coordinate conversion unit The negative sequence 2-fold frequency component of the circulating current converted to the dq coordinate system , ; Reference value of d-axis component of negative sequence 2 times frequency component of circulating current The d-axis component of the negative sequence 2nd frequency component of the circulating current The difference between them is passed through the first PI regulator to obtain the first signal, the q-axis component of the negative sequence 2 times frequency component of the circulating current and Phase reactance The product of two times is taken as the second signal, and the difference between the first signal and the second signal is taken as the d-axis component of the voltage drop caused by the negative sequence double frequency component of the circulating current. ; Reference value of q-axis component of negative sequence 2 times frequency component of circulating current The q-axis component of the negative-sequence 2nd frequency component of the circulating current The difference between them is passed through the second PI regulator to obtain the third signal, the d-axis component of the negative sequence 2 times frequency component of the circulating current and Phase reactance The product of two times is taken as the fourth signal, and the sum of the third signal and the fourth signal is taken as the q-axis component of the voltage drop caused by the negative sequence double frequency component of the circulating current. ; The d-axis component of the voltage drop caused by the negative-sequence 2-fold frequency component of the circulating current and the q-axis component Through the second coordinate conversion unit Obtain the reference value of the negative sequence 2 times frequency component of the circulating current in the three-phase bridge arm in the stationary coordinate system , , .

[0089] Step 3, the voltage control outer loop of each MMC sub-module outputs the AC output voltage reference value of each MMC sub-module; the voltage difference between the modulation wave correction component of each MMC sub-module and the reference value of the negative-sequence 2 times frequency component of the updated circulating current and the AC output voltage reference value is used as the modulation voltage of each MMC sub-module, and the trigger pulse of the power device is obtained according to the modulation voltage of each MMC sub-module.

[0090] In the embodiment, the difference between the modulation wave correction component of each MMC submodule in the upper bridge arm of each phase and the reference value of the negative sequence 2 times frequency component of the circulating current in the upper bridge arm of each phase is used as the first voltage difference; the difference between the modulation wave correction component of each MMC submodule in the lower bridge arm of each phase and the reference value of the negative sequence 2 times frequency component of the circulating current in the lower bridge arm of each phase is used as the second voltage difference; the voltage control outer loop of each MMC submodule on each phase is based on the capacitor voltage reference value , Active power reference value and reactive power reference Output AC output voltage reference value of each MMC submodule The difference between the first voltage difference and the AC output voltage reference value of each MMC submodule in the upper bridge arm of each phase is used as the modulation voltage of each MMC submodule in the upper bridge arm of each phase The difference between the second voltage difference and the AC output voltage reference value of each MMC submodule in the lower bridge arm of each phase is used as the modulation voltage of each MMC submodule in the lower bridge arm of each phase. ; A trigger pulse of a power device in the MMC submodule is obtained according to the modulation voltage of each MMC submodule in the upper bridge arm and the lower bridge arm of each phase.

[0091] In the embodiment, the modulation voltage of each MMC sub-module in the upper bridge arm and the lower bridge arm of each phase generates a trigger pulse of the power device in the MMC sub-module through a CPS-PWM modulator.

[0092] The topology of the voltage control outer loop of each MMC submodule on each phase is as follows: Figure 5 As shown, it includes a positive sequence current decoupling control loop; the rated capacitor voltage of each MMC submodule on each phase The average value of the capacitor voltage of all MMC submodules The voltage difference between them is used as the input signal of the voltage control outer loop. The current obtained after the voltage difference passes through the PI loop is the d-axis component of the system three-phase current. , q-axis component of the system three-phase current And the reference value of the q-axis component of the system three-phase current As the input signal of the positive sequence current decoupling control loop, the d-axis component of the potential difference output by the positive sequence current decoupling control loop is equal to the d-axis component of the system three-phase potential. The sum of the q-axis component of the potential difference output by the positive sequence current decoupling control loop and the q-axis component of the system three-phase potential The sum of Coordinate transformation obtains the reference value of the three-phase output voltage on the AC side of all MMC submodules , , ; Set the reference value of the three-phase output voltage on the AC side of all MMC submodules , , Evenly distribute to each MMC submodule to obtain the reference value of the three-phase output voltage on the AC side of each MMC submodule , , .

[0093] MMC provides high-voltage AC and high-voltage DC interfaces respectively. In order to transfer energy between ports, the transmission power needs to be controlled, and its essence is the control of the converter current. Therefore, the current controller is designed first.

[0094] based on Figure 6 The MMC submodule shown The equivalent circuit is obtained to obtain the mathematical model of the MMC submodule, which satisfies the following relationship:

[0095]

[0096] In the formula, , are the equivalent resistance and equivalent inductance of the system respectively, , , Separately for the moment The system three-phase current, , , Separately for the moment The system equivalent voltage source , , The corresponding three-phase potential is, , , The AC side of the MMC submodule is The three-phase output voltage;

[0097] Figure 6 In the example, a neutral point O is taken from the DC side of the MMC submodule, and the potential of this point is equal to the neutral point potential of the AC side. The connection point between the upper and lower bridge arms of the phase Voltage , The equivalent voltage source of the upper bridge arm of the phase , equivalent inductance , Current , The equivalent voltage source of the lower bridge arm of the phase , equivalent inductance , Current , the voltage between the positive electrode P and the neutral point O on the DC side of the MMC submodule is , the voltage between the negative pole N and the neutral point O on the DC side of the MMC submodule is .

[0098] The mathematical model of the MMC submodule is converted from the stationary coordinate system to the dq coordinate system to achieve decoupling, and the relationship in the dq coordinate system is obtained:

[0099]

[0100] In the formula, , are the d-axis component and q-axis component of the system three-phase current respectively, , are the d-axis component and q-axis component of the system three-phase potential, , are the d-axis component and q-axis component of the three-phase output voltage on the AC side of the MMC submodule, is the system angular frequency.

[0101] Since the variables of the d-axis and q-axis are coupled to each other, the d-axis component and q-axis component of the system three-phase current are introduced as feedforward, satisfying the following relationship:

[0102]

[0103] In the formula, , are the coefficients of the PI ring, is the Laplace operator, , They are the reference values ​​of the d-axis component and the q-axis component of the system three-phase current, respectively, which are calculated according to the control objective and the designed outer loop controller.

[0104] Based on the above structure Figure 7 The current inner loop decoupling control structure is shown.

[0105] The decoupled control equation satisfies the following relationship:

[0106]

[0107] From the above formula, we can see that the active current and reactive current The decoupling has been fully achieved.

[0108] The capacitor voltage balance control of the MMC submodule can be realized in the control of the MMC submodule, and can also be realized in the control of the bidirectional active full-bridge output parallel system. If the capacitor voltage balance control of the MMC submodule is only realized in the control of the MMC submodule, then the capacitor voltage balance control of the MMC submodule is not considered in the bidirectional active full-bridge output parallel system, and only the output voltage constant value control of the bidirectional active full-bridge output parallel system and the balance control of the transmission power of each DAB need to be realized. If the capacitor voltage balance control of the MMC submodule is only realized in the control of the bidirectional active full-bridge output parallel system, the capacitor voltage balance control of the MMC submodule is no longer considered in the control of the MMC submodule, and the output voltage constant value control still needs to be realized in the control of the bidirectional active full-bridge output parallel system. Therefore, it is proposed to realize the capacitor voltage balance control of the MMC submodule in the control of the bidirectional active full-bridge output parallel system, without measuring the DC current output by each DAB, significantly reducing the hardware cost, reducing the related calculations of the DC current output by each DAB in the control algorithm, thereby reducing the control complexity. Moreover, when the capacitor voltage balance control of the MMC submodule is realized in the bidirectional active full-bridge output parallel system, the input voltage of each DAB can be processed for voltage balancing, and there is no need to perform capacitor voltage balance control in the control of the MMC submodule, but only the average value of the capacitor voltage of all MMC submodules is controlled. Therefore, based on the current inner loop decoupling control structure, the present invention uses the rated capacitor voltage of each MMC submodule on each phase as the The average value of the capacitor voltage of all MMC submodules The voltage difference between the two is passed through the PI loop and the current obtained is used as the reference value of the d-axis component of the system three-phase current; in the obtained positive sequence current decoupling control loop, and The voltage difference between the two is passed through the PI loop to obtain the current and the d-axis component of the system three-phase current. The current difference between the two phases is obtained through the PI loop to obtain the corresponding d-axis component of the intermediate potential difference and the q-axis component of the system three-phase current. Equivalent reactance of the system The difference between the product of and the d-axis component of the intermediate potential difference is used as the d-axis component of the potential difference output by the positive sequence current decoupling control loop; the reference value of the q-axis component of the system three-phase current The q-axis component of the system three-phase current The current difference between the two phases is obtained through the PI loop to obtain the corresponding q-axis component of the intermediate potential difference and the d-axis component of the system three-phase current. Equivalent reactance of the system The difference between the product of and the q-axis component of the intermediate potential difference is taken as the q-axis component of the potential difference output by the positive-sequence current decoupling control loop;

[0109] In the voltage control outer loop, Introducing parameters in coordinate transformation ; The system three-phase current through Coordinate transformation obtains the dq axis components of the system three-phase current , the system three-phase electromotive force through Coordinate transformation obtains the dq axis components of the system's three-phase electromotive force ;The total number of MMC submodules is .

[0110] In the embodiment, each MMC submodule on each phase also adopts power outer loop control to achieve control of active power and reactive power transmitted by the MMC submodule.

[0111] The voltage outer loop controller includes a voltage outer loop, which are essentially the same as those in the embodiment. Figure 7 The voltage outer loop control is shown.

[0112] Based on steps 1 to 3, the control method of the MMC submodule is determined, and the corresponding control structure is as follows: Figure 2 shown.

[0113] Step 4, using the difference between the secondary DC side voltage of the bidirectional active full bridge and the corresponding reference value, based on the phase shift proportional loop, generate the phase shift angle reference value of all bidirectional active full bridges; using the difference between the DC input voltage of each bidirectional active full bridge and the average value of the DC voltage of all MMC sub-modules, based on the phase shift proportional loop, generate the phase shift angle correction value of each bidirectional active full bridge, and use the difference between the phase shift angle reference value and the phase shift angle correction value as the phase shift angle of each bidirectional active full bridge.

[0114] In a bidirectional active full-bridge output parallel system, such as Figure 8 As shown, the DC side voltage of the DAB secondary side is and the reference value of the DC side voltage of the DAB secondary side The difference is passed through the first phase shift proportional loop PI-4 to obtain the phase shift angle reference value of all DABs. ; The first corresponding to each MMC sub-mode on each phase DAB input DC voltage The average value of DC voltage of all MMC submodules The voltage difference is passed through the second phase-shift proportional loop PI-3 to obtain the first Phase shift angle correction of DAB ,in, , is the total number of MMC submodules, which is also the total number of DABs; the phase shift angle reference value Phase shift angle correction value of each DAB The difference is used as the phase shift angle of each DAB .

[0115] When the capacitor voltage balance control of the MMC submodule is realized in the bidirectional active full-bridge output parallel system, the control objectives of the bidirectional active full-bridge output parallel system include: capacitor voltage balance of the MMC submodule and constant DAB output voltage. Therefore, the voltage difference between the average value of the capacitor voltage of all MMC submodules and the actual capacitor voltage of each MMC submodule in the upper and lower bridge arms of each phase is used to replace the first capacitor voltage of each MMC submodule on each phase. DAB input DC voltage The average value of DC voltage of all MMC submodules The voltage difference is used to realize the capacitor voltage balance control of the MMC sub-module in the bidirectional active full-bridge output parallel system.

[0116] The first phase shift ratio loop PI-3 and the second phase shift ratio PI-4 both adopt a single phase shift control method, such as Fig. 9 As shown, according to the output side DC voltage command value and actual value The voltage difference between the two is transferred through the phase proportional loop. To adjust the phase shift angle between the primary and secondary sides of DAB , thereby changing the size of the power flow and achieving the stability of the DC voltage on the output side. and These are the coefficients of the phase-shift proportional loop.

[0117] The shift ratio of the bidirectional active full bridge is calculated using the following relationship:

[0118]

[0119] In the formula, For the shift ratio of the bidirectional active full bridge, The ratio of the medium frequency transformer of the bidirectional active full bridge is , They are the primary DC side voltage and the secondary DC side voltage of DAB respectively;

[0120] By comparing the operating mode of DAB in hard switching mode, it is determined that the constraint condition for DAB to achieve soft switching is the maximum value of DAB primary current. and the maximum value of DAB secondary current All must be greater than 0, and the following relationship is satisfied:

[0121]

[0122] In the formula, is the phase shift angle of the bidirectional active full bridge;

[0123] The phase shift angle of each bidirectional active full bridge is limited to control the bidirectional active full bridge to operate in a soft switching mode.

[0124] The present invention also proposes an MMC type power electronic transformer control system of a bidirectional active full-bridge output parallel system, wherein the upper bridge arm and the lower bridge arm of each phase of the MMC type power electronic transformer include a plurality of units with the same structure, each unit, each unit includes an MMC submodule and a bidirectional active full-bridge DAB, wherein the AC side of the MMC submodule is cascaded, the bidirectional active full-bridge DAB is connected in parallel to the DC side capacitor of the MMC submodule, and the DC sides of the bidirectional active full-bridge DABs are connected in parallel to form a bidirectional active full-bridge output parallel system; the invention is characterized in that it includes:

[0125] The acquisition module is used to obtain the actual capacitor voltage and rated capacitor voltage of each MMC submodule, and the current and circulating current of the upper and lower bridge arms of each phase of the MMC power electronic transformer;

[0126] The MMC submodule capacitor voltage control module is used to generate a modulation wave correction component of each MMC submodule based on a voltage proportional control loop using the actual capacitor voltage, the rated capacitor voltage and the current;

[0127] The MMC submodule negative-sequence 2 times frequency circulating current suppression module is used to generate a compensation voltage based on the PI loop using the difference between the negative-sequence 2 times frequency component of the circulating current and the corresponding reference value. The compensation voltage compensates for the voltage drop caused by the negative-sequence 2 times frequency component of the circulating current to update the reference value of the negative-sequence 2 times frequency component of the circulating current;

[0128] The MMC submodule modulation voltage module is used for the voltage control outer loop of each MMC submodule to output the AC output voltage reference value of each MMC submodule; the voltage difference between the modulation wave correction component of each MMC submodule and the reference value of the negative sequence 2 times frequency component of the updated circulating current and the AC output voltage reference value is used as the modulation voltage of each MMC submodule;

[0129] MMC submodule trigger pulse module, used to obtain the trigger pulse of the power device according to the modulation voltage of each MMC submodule;

[0130] The bidirectional active full-bridge output parallel system control module is used to generate the phase shift angle reference value of all bidirectional active full bridges based on the phase shift proportional loop by using the difference between the secondary DC side voltage of the bidirectional active full bridge and the corresponding reference value; and to generate the phase shift angle correction value of each bidirectional active full bridge based on the phase shift proportional loop by using the difference between the DC input voltage of each bidirectional active full bridge and the average value of the DC voltage of all MMC sub-modules, and to use the difference between the phase shift angle reference value and the phase shift angle correction value as the phase shift angle of each bidirectional active full bridge.

[0131] The MMC type power electronic transformer overall control system of the bidirectional active full-bridge output parallel system proposed by the present invention utilizes the DC side of the MMC submodule as the connection point of the bidirectional active full-bridge output parallel system, which significantly simplifies the measurement complexity. By realizing capacitor voltage balance control in a multi-DAB system, the system reduces the dependence on the DAB output DC current measurement circuit and reduces the hardware cost. This coupling control method enables the capacitor voltage of the submodule to be balanced in the DAB system, ensuring the stability of multiple DAB modules. Under the control of voltage balance and current balance, each DAB output current can be dynamically adjusted by a PI regulator. This method ensures the balance of current distribution, thereby effectively controlling power transmission and fluctuations.

[0132] In traditional control systems, complex current and voltage measurements increase the difficulty of control system design. Fig.10 The control system shown achieves constant output voltage through a multi-DAB parallel system, which greatly reduces the control difficulty and design complexity of the system. Specifically, the voltage balancing problem of the sub-module is transferred from the MMC module to the DAB output module, which simplifies the task allocation of each control module and ensures the stability and efficiency of the overall control system.

[0133] The present disclosure may be a system, a method and / or a computer program product. The computer program product may include a computer-readable storage medium carrying computer-readable program instructions for causing a processor to implement various aspects of the present disclosure.

[0134] A computer-readable storage medium may be a tangible device that can hold and store instructions used by an instruction execution device. A computer-readable storage medium may be, for example, but not limited to, an electrical storage device, a magnetic storage device, an optical storage device, an electromagnetic storage device, a semiconductor storage device, or any suitable combination of the above. More specific examples (a non-exhaustive list) of computer-readable storage media include: a portable computer disk, a hard disk, a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or flash memory), a static random access memory (SRAM), a portable compact disk read-only memory (CD-ROM), a digital versatile disk (DVD), a memory stick, a floppy disk, a mechanical encoding device, such as a punch card or a raised structure in a groove on which instructions are stored, and any suitable combination of the above. The computer-readable storage medium used herein is not to be interpreted as a transient signal itself, such as a radio wave or other freely propagating electromagnetic wave, an electromagnetic wave propagating through a waveguide or other transmission medium (e.g., a light pulse through a fiber optic cable), or an electrical signal transmitted through a wire.

[0135] The computer-readable program instructions described herein can be downloaded from a computer-readable storage medium to each computing / processing device, or downloaded to an external computer or external storage device via a network, such as the Internet, a local area network, a wide area network, and / or a wireless network. The network can include copper transmission cables, optical fiber transmissions, wireless transmissions, routers, firewalls, switches, gateway computers, and / or edge servers. The network adapter card or network interface in each computing / processing device receives the computer-readable program instructions from the network and forwards the computer-readable program instructions for storage in the computer-readable storage medium in each computing / processing device.

[0136] The computer program instructions for performing the operations of the present disclosure may be assembly instructions, instruction set architecture (ISA) instructions, machine instructions, machine-dependent instructions, microcode, firmware instructions, state setting data, or source code or object code written in any combination of one or more programming languages, including object-oriented programming languages, such as Smalltalk, C++, etc., and conventional procedural programming languages, such as "C" language or similar programming languages. The computer-readable program instructions may be executed entirely on the user's computer, partially on the user's computer, as a separate software package, partially on the user's computer, partially on a remote computer, or entirely on a remote computer or server. In the case of a remote computer, the remote computer may be connected to the user's computer through any type of network, including a local area network (LAN) or a wide area network (WAN), or may be connected to an external computer (e.g., using an Internet service provider to connect through the Internet). In some embodiments, by using the state information of the computer-readable program instructions to personalize an electronic circuit, such as a programmable logic circuit, a field programmable gate array (FPGA), or a programmable logic array (PLA), the electronic circuit may execute the computer-readable program instructions, thereby implementing various aspects of the present disclosure.

[0137] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention rather than to limit it. Although the present invention has been described in detail with reference to the above embodiments, ordinary technicians in the relevant field should understand that the specific implementation methods of the present invention can still be modified or replaced by equivalents, and any modifications or equivalent replacements that do not depart from the spirit and scope of the present invention should be covered within the scope of protection of the claims of the present invention.

Claims

1. A control method for an MMC type power electronic transformer of a bidirectional active full-bridge parallel system, wherein the MMC type power electronic transformer comprises a plurality of MMC sub-modules cascaded on the AC side and a plurality of bidirectional active full-bridges in parallel on the DC side, wherein the bidirectional active full-bridges are connected in parallel to the DC side capacitors of the MMC sub-modules, and wherein: include: Obtain the actual capacitor voltage and rated capacitor voltage of each MMC submodule, and the current and circulating current of the upper and lower bridge arms of each phase of the MMC power electronic transformer; Using the actual capacitor voltage, rated capacitor voltage and current, a modulation wave correction component of each MMC submodule is generated based on a voltage proportional control loop; Using the difference between the negative-sequence double frequency component of the circulating current and the corresponding reference value, a compensation voltage is generated based on the PI loop, and the compensation voltage compensates for the voltage drop caused by the negative-sequence double frequency component of the circulating current to update the reference value of the negative-sequence double frequency component of the circulating current; The voltage control outer loop of each MMC submodule outputs the AC output voltage reference value of each MMC submodule; The voltage difference between the modulation wave correction component of each MMC submodule and the reference value of the negative sequence double frequency component of the updated circulating current and the AC output voltage reference value is used as the modulation voltage of each MMC submodule, and the trigger pulse of the power device is obtained according to the modulation voltage of each MMC submodule; The difference between the secondary DC side voltage of the bidirectional active full bridge and the corresponding reference value is used to generate the phase shift angle reference value of all bidirectional active full bridges based on the phase shift proportional loop; The difference between the DC input voltage of each bidirectional active full-bridge and the average DC voltage of all MMC sub-modules is used to generate the phase shift angle correction of each bidirectional active full-bridge based on the phase shift proportional loop, and the difference between the phase shift angle reference value and the phase shift angle correction is used as the phase shift angle of each bidirectional active full-bridge.

2. The MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system according to claim 1 is characterized in that: The modulation wave correction component of the MMC submodule in the upper bridge arm of each phase of the MMC type power electronic transformer and the modulation wave correction component of the MMC submodule in the lower bridge arm respectively satisfy the following relationship: In the formula, , They are The first The modulation wave correction component of each MMC submodule, is the proportional coefficient of the voltage proportional control loop, is the rated capacitance voltage of the MMC submodule, , They are The first The actual capacitor voltage of each MMC submodule, , They are The upper arm current and lower arm current of the phase, where , is the total number of MMC submodules, , indicating different phases.

3. The MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system according to claim 1 is characterized in that: The even harmonic components in the circulating current include: 6k+2 harmonic components with negative sequence distribution, 6k+4 harmonic components with positive sequence distribution, and 6k+6 harmonic components with zero sequence distribution, where k is a non-negative integer; When k is 0, the AC component in the second harmonic component with negative sequence distribution serves as the negative sequence double frequency component of the circulating current.

4. The MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system according to claim 3 is characterized in that: In the stationary coordinate system, the negative sequence 2 times frequency component of the circulating current of the upper and lower bridge arms of each phase , , , which is converted into the negative sequence 2-fold frequency component of the circulating current in the dq coordinate system by the coordinate conversion unit , ; Reference value of d-axis component of negative sequence 2 times frequency component of circulating current The d-axis component of the negative sequence 2nd frequency component of the circulating current The difference between them is obtained by the PI regulator to obtain the d-axis component of the compensation voltage, and the q-axis component of the negative-sequence double frequency component of the circulating current. Inductance in the bridge arm The q-axis component of the voltage drop caused by the upper current is compensated; the q-axis component reference value of the negative sequence 2 times frequency component of the circulating current The q-axis component of the negative-sequence 2nd frequency component of the circulating current The difference between them is obtained by the PI regulator to obtain the q-axis component of the compensation voltage, and the d-axis component of the negative-sequence double frequency component of the circulating current Inductance in the bridge arm The voltage drop d-axis component caused by the upper side is compensated to satisfy the following relationship: In the formula, is the voltage vector after the voltage drop caused by the negative sequence double frequency component of the circulating current in the dq coordinate system is compensated, , is the voltage after the voltage drop caused by the negative sequence 2 times frequency component of the circulating current in the dq coordinate system is compensated, , is the negative sequence 2-fold frequency component of the circulating current in the dq coordinate system, is the system angular frequency, , They are The resistance and inductance of the phase, , indicating different phases; The voltage d-axis component after the voltage drop caused by the negative sequence 2 times frequency component of the circulating current is compensated and the q-axis component The reference value of the negative sequence 2 times frequency component of the updated circulating current in the upper and lower bridge arms of the three-phase stationary coordinate system is obtained by the coordinate conversion unit. , , .

5. The MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system according to claim 1, characterized in that: The modulation voltage of each MMC submodule generates a trigger pulse for the power device in the MMC submodule through a CPS-PWM modulator.

6. The MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system according to claim 1, characterized in that: The voltage control outer loop of each MMC submodule includes: positive sequence current decoupling control loop; rated capacitor voltage of each MMC submodule The average value of the actual capacitor voltage of all MMC submodules The input current obtained by the PI loop after the difference between , q-axis component , q-axis component reference value As the input signal of the positive sequence current decoupling control loop, the potential difference d-axis component output by the positive sequence current decoupling control loop is related to the d-axis component of the system three-phase potential The sum of the potential difference of the positive sequence current decoupling control loop output and the q-axis component of the system three-phase potential The sum of the AC output voltage reference values ​​of all MMC sub-modules is obtained through coordinate transformation. , , ; Set the AC output voltage reference value of all MMC submodules , , According to the total number of MMC submodules Evenly distribute to each MMC submodule to obtain the AC output voltage reference value of each MMC submodule , , .

7. The MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system according to claim 6, characterized in that: In the positive sequence current decoupling control loop, the d-axis component of the input current and the system three-phase current The difference between the two is passed through the PI loop to obtain the d-axis component of the intermediate potential difference and the q-axis component of the system three-phase current. Equivalent reactance of the system The product of is used to compensate the d-axis component of the intermediate potential difference, and the d-axis component of the potential difference output by the positive sequence current decoupling control loop is obtained; the q-axis component reference value of the system three-phase current is With q-axis component The difference between the two is passed through the PI loop to obtain the q-axis component of the intermediate potential difference and the d-axis component of the system three-phase current. Equivalent reactance of the system The product of is used to compensate the q-axis component of the intermediate potential difference to obtain the q-axis component of the potential difference output by the positive-sequence current decoupling control loop.

8. The MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system according to claim 1, characterized in that: The transfer function of the phase-shift proportional loop is: ,in, and are the coefficients of the phase-shift proportional loop, is the Laplace operator.

9. The MMC type power electronic transformer control method of the bidirectional active full-bridge parallel system according to claim 8, characterized in that: The shift ratio of the bidirectional active full bridge is calculated using the following relationship: In the formula, For the shift ratio of the bidirectional active full bridge, is the ratio of the medium frequency transformer of the bidirectional active full bridge. , They are the primary DC side voltage and the secondary DC side voltage of DAB respectively; The phase shift angle of each bidirectional active full bridge is limited by the following relationship to control the bidirectional active full bridge to operate in the soft switching mode: In the formula, is the phase shift angle of the bidirectional active full bridge.

10. MMC type power electronic transformer control system of bidirectional active full-bridge parallel system, characterized in that: include: The acquisition module is used to obtain the actual capacitor voltage and rated capacitor voltage of each MMC submodule, and the current and circulating current of the upper and lower bridge arms of each phase of the MMC power electronic transformer; The MMC submodule capacitor voltage control module is used to generate a modulation wave correction component of each MMC submodule based on a voltage proportional control loop using the actual capacitor voltage, the rated capacitor voltage and the current; The MMC submodule negative-sequence 2 times frequency circulating current suppression module is used to generate a compensation voltage based on the PI loop using the difference between the negative-sequence 2 times frequency component of the circulating current and the corresponding reference value. The compensation voltage compensates for the voltage drop caused by the negative-sequence 2 times frequency component of the circulating current to update the reference value of the negative-sequence 2 times frequency component of the circulating current; MMC submodule voltage modulation module, used for the voltage control outer loop of each MMC submodule to output the AC output voltage reference value of each MMC submodule; The voltage difference between the modulation wave correction component of each MMC submodule and the reference value of the negative sequence double frequency component of the updated circulating current and the AC output voltage reference value is used as the modulation voltage of each MMC submodule; MMC submodule trigger pulse module, used to obtain the trigger pulse of the power device according to the modulation voltage of each MMC submodule; A bidirectional active full-bridge output parallel system control module is used to generate a phase shift angle reference value of all bidirectional active full-bridges based on a phase shift proportional loop by using a difference between a secondary DC side voltage of the bidirectional active full-bridge and a corresponding reference value; The difference between the DC input voltage of each bidirectional active full-bridge and the average DC voltage of all MMC sub-modules is used to generate the phase shift angle correction of each bidirectional active full-bridge based on the phase shift proportional loop, and the difference between the phase shift angle reference value and the phase shift angle correction is used as the phase shift angle of each bidirectional active full-bridge.

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