PWM (Pulse Width Modulation) and phase-shift modulation current type push-pull circuit circulating current suppression method

Abstract

The invention discloses a PWM (Pulse Width Modulation) and phase-shift modulation current type push-pull circuit circulating current suppression method, which is characterized in that PWM and phase-shift modulation are combined with the operation condition of a converter, and the converter works in a low circulating current state by optimally controlling the duty ratio D of power tubes S1 and S2 of a primary side circuit of the current type push-pull circuit and the inner phase-shift angle D2 between bridge arms of a secondary side full-bridge circuit. According to the invention, the reverse current of the power tubes S1 and S2 of the primary side circuit is significantly reduced, and the soft switching performance of the power tubes S1 and S2 of the primary side circuit is retained; the problem of voltage spikes of a power tube of a traditional current type converter is solved, and an active clamping circuit or a passive buffer for suppressing the voltage spikes is omitted; the two control variables D and D2 obtained through modulation control the converter, so that the current matching time is optimized, the circulating current power is remarkably reduced, and the efficiency of the circuit is effectively improved.

Description

Technical Field

[0001] The invention relates to the technical field of direct current converters, and in particular to a method for suppressing circulating current in a current-type push-pull circuit with PWM plus phase-shift modulation. Background Art

[0002] The current-mode push-pull circuit has the advantages of low input current ripple, high step-up ratio, electrical isolation and simple structure, and is widely used in renewable energy power supply systems. However, the converter has an inherent problem, that is, the long-term mismatch between the current flowing through the input inductor and the current of the transformer series inductor will lead to severe voltage spikes and circulating currents. The circulating current causes multiple energy exchanges inside the converter, resulting in additional losses and reducing the converter efficiency. The voltage spike causes the power tube to bear large voltage stress, which can easily lead to device breakdown or thermal damage. To address this problem, the IEEE Transactions on Power Electronics journal reported a single PWM control strategy for the current-mode push-pull circuit in the paper "Naturally Clamped Zero-Current Commutated Soft-Switching Current-Fed Push–Pull DC / DC Converter: Analysis, Design, and Experimental Results" in Volume 30, Issue 3, 2015. It does not require an active clamping circuit or a passive buffer, and solves the voltage spike problem of the power tube shutdown of the current-mode converter. However, the current of the power tube in the primary circuit of the converter using this method flows for a long time before current matching. When the input voltage is reduced or the converter transmission power is small, the reverse current of the power tube in the primary circuit is large, and there is a problem of large circulating current. In order to further reduce the circulating current, the IEEE Transactions on Power Electronics reported a dual PWM plus phase shift control of a current-type dual active full-bridge circuit in the paper "An Improved Modulation Scheme of Current-Fed Bidirectional DC–DC Converters For Loss Reduction" in Volume 33, Issue 5, 2018, which uses more control freedom to obtain better performance, but the modulation becomes complicated. In order to reduce the complexity of modulation, the IEEE Journal of Emerging and Selected Topics in Power Electronics reported a dual PWM control strategy for a current-type dual active full-bridge circuit in the paper "Decoupled Dual-PWM Control for Naturally Commutated Current-Fed Dual-Active-Bridge DC / DC Converter" in Volume 8, Issue 4, 2020, which reduces the reverse current of the power tube in the primary circuit, thereby reducing the circulating current.However, this method will cause discontinuity in the input inductor current, resulting in voltage and current oscillation of the converter, affecting system stability, being unfavorable for achieving soft switching, and increasing the loss of the converter. Summary of the invention

[0003] The present invention aims to address the technical deficiencies described in the background and propose a method for suppressing circulating current in a current-type push-pull circuit using PWM plus phase-shift modulation, so that it has the advantages of being able to solve the voltage spike problem of the power tube and the excessive reverse current problem of the power tube in traditional current-type converters and significantly reduce the circulating current.

[0004] The present invention is achieved by adopting the following technical solutions:

[0005] A method for suppressing circulating current in a current-type push-pull circuit using PWM plus phase-shift modulation. The circulating current suppression method uses PWM plus phase-shift modulation, combined with the operating conditions of a converter, to optimize and control the duty cycle D of power tubes S1 and S2 of the primary circuit of the current-type push-pull circuit, and the internal phase shift angle D2 between the bridge arms of the secondary full-bridge circuit, so that the converter operates in a low circulating current state.

[0006] The duty cycle of the power tubes S1 and S2 of the primary circuit is defined as D, the overlapping part of the duty cycle of the power tubes S1 and S2 in half a cycle is defined as D1, and the mathematical relationship between D and D1 is D=D1+0.5; S3 and S4 of the secondary full-bridge circuit constitute the first bridge arm, S5 and S6 constitute the second bridge arm, the duty cycle of the first bridge arm power tubes S3, S4 and the second bridge arm power tubes S5 and S6 is 0.5, the internal phase shift angle between the first bridge arm and the second bridge arm circuit is defined as D2, and the power tubes of the same bridge arm are complementary; the charging time of the input inductor in half a cycle is defined as T1, and the discharging time is defined as T2.

[0007] The optimization calculation steps of the control variables D, D2 include:

[0008] Step 1: Derive the transformer leakage inductance instantaneous current expression of the converter;

[0009] The current push-pull circuit adopts PWM plus phase shift modulation. Since the circuit is symmetrical in the first half cycle and the second half cycle, only the operating mode of the first half cycle is analyzed, and the half cycle working mode is divided into three modes: the t0-t2 period is defined as mode 1, the t2-t3 period is defined as mode 2, and the t3-t4 period is defined as mode 3; according to Kirchhoff's law and Thevenin equivalent law, the transformer leakage inductance voltage V of the converter under the three modes is derived respectively. LS The expression of (t):

[0010]

[0011] Where: V1, V2 are the voltages across the converter, n is the primary-to-secondary turns ratio of the transformer, L is the input inductance, LS is the transformer leakage inductance;

[0012] According to formula (1), the transformer leakage inductance current i of the converter under the three modes is derived respectively: LS The expression of (t) is:

[0013]

[0014] Where: i LS (t2),i LS (t3) and i LS (t4) are the instantaneous currents of the transformer leakage inductance at t2, t3 and t4 respectively;

[0015] Furthermore, according to formula (2), the transformer leakage inductance instantaneous current i of the converter at time t2, t3 and t4 is calculated respectively: LS (t2),i LS (t3) and i LS The expression of (t4) is:

[0016]

[0017] Where: T S is the switching period of the power tube of the converter;

[0018] Step 2: Based on the principle that the average value of the input inductor current is the same as the converter input current, derive the input inductor current expression at time t0;

[0019] Considering that the input inductor current change rates of mode 1 and mode 2 are similar, the change rates of the inductor currents of the two modes are approximately the same to simplify the calculation; the input inductor current expression of the t0-t3 converter is:

[0020]

[0021] Where: P is the transmission power of the converter;

[0022] The change in input inductor current from t0 to t3 is:

[0023]

[0024] According to equations (4) and (5), the input inductor current expression at time t0 is derived as follows:

[0025]

[0026] Step 3: Based on the symmetry of the transformer leakage inductance current in one cycle and the relationship between the transformer primary and secondary currents, derive the expression of the input inductor current excluding P at time t0;

[0027] The symmetry of the transformer leakage current in one cycle shows that LS (t0) = -i LS (t5), combined with formula (3), derive i LS The expression of (t0) is:

[0028]

[0029] Furthermore, according to the current relationship between the primary and secondary sides of the transformer, L (t0) = -i LS (t5) / n can be derived i L The expression of (t0) is:

[0030]

[0031] Step 4: Solve for the optimal variable combination of D;

[0032] First, we eliminate i from equations (6) and (8) LS (t0), we get a linear equation with two variables including D1 and D2:

[0033]

[0034] Furthermore, the volt-second area balance equation is established based on the input inductor current of mode 1 and mode 2:

[0035]

[0036] Combining equations (9) and (10) yields the expression for D1:

[0037]

[0038] Finally, the mathematical expression of D is calculated from the mathematical relationship between D and D1:

[0039]

[0040] Step 5: Derive the optimal variable combination of D2;

[0041] The discharge time T2 of the input inductor in half a cycle is derived from formula (10):

[0042]

[0043] Furthermore, according to the mathematical relationship between the input inductor current charging and discharging time and cycle:

[0044]

[0045] Combining equations (13) and (14) yields the expression for T1:

[0046]

[0047] Finally, substitute equation (15) into the mathematical relationship D2 = T1 / T S -D1 gives the mathematical expression of D2:

[0048]

[0049] Compared with the prior art, the present invention adopts the above technical solution and has the following beneficial effects:

[0050] (1) The modulation step of the present invention combines the working condition that the power tubes S1 and S2 of the primary circuit of the converter have no reverse current. At the same time, the approximate calculation method is used in the calculation process, which significantly reduces the reverse current of the power tubes S1 and S2 of the primary circuit, and retains the soft switching performance of the power tubes S1 and S2 of the primary circuit.

[0051] (2) The two control variables D and D2 obtained by modulation based on the present invention control the converter, which can solve the voltage spike problem of the power tube of the traditional current-type converter, eliminate the active clamping circuit or passive buffer for suppressing the voltage spike, and can improve the power density of the converter.

[0052] (3) Based on the two control variables D and D2 obtained by modulation of the present invention, the converter is controlled, which can reduce the mismatch time between the input inductor current and the secondary current and reduce the circulating current. BRIEF DESCRIPTION OF THE DRAWINGS

[0053] Figure 1 It is a current-type push-pull circuit topology;

[0054] Figure 2 This is the circuit diagram of mode 1 (t0-t2);

[0055] Figure 3 This is the circuit diagram of mode 2 (t2-t3) stage;

[0056] Figure 4 This is the circuit diagram of mode 3 (t3-t4) stage;

[0057] Figure 5 The two control variables D and D2 modulated by the present invention control the main waveforms of the converter;

[0058] Figure 6 This is the simulation waveform of the current push-pull circuit with traditional PWM plus phase shift;

[0059] Figure 7 This is the simulation waveform of the current push-pull circuit with PWM plus phase shift modulation of the present invention. DETAILED DESCRIPTION

[0060] The following will clearly and completely describe the technical solutions in the embodiments of the present invention in conjunction with the drawings of the specification. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary persons in the art without creative work are within the scope of protection of the present invention.

[0061] The present invention provides a method for suppressing circulating current in a current-type push-pull circuit with PWM plus phase shift modulation. The current-type push-pull circuit is as follows: Figure 1 As shown, the circulating current suppression method adopts PWM plus phase shift modulation, combined with the converter operating conditions, by optimizing the control current type push-pull circuit primary circuit power tube S1, S2 duty cycle D, and the secondary full-bridge circuit bridge arm internal phase shift angle D2, so that the converter works in a low circulating current state;

[0062] The duty cycle of the power tubes S1 and S2 of the primary circuit is defined as D, and the overlapping part of the duty cycle of the power tubes S1 and S2 in half a cycle is defined as D1. The mathematical relationship between D and D1 is D=D1+0.5; S3 and S4 of the secondary full-bridge circuit constitute the first bridge arm, and S5 and S6 constitute the second bridge arm. The duty cycle of the first bridge arm power tubes S3 and S4 and the second bridge arm power tubes S5 and S6 is 0.5. The internal phase shift angle between the first bridge arm and the second bridge arm circuit is defined as D2, and the power tubes of the same bridge arm are complementary. The charging time of the input inductor in half a cycle is defined as T1, and the discharging time is defined as T2;

[0063] The optimization calculation steps of the control variables D, D2 include:

[0064] Step 1: Derive the transformer leakage inductance instantaneous current expression of the converter;

[0065] The current push-pull circuit adopts PWM plus phase shift modulation. Since the circuit is symmetrical in the first half cycle and the second half cycle, only the operation mode of the first half cycle is analyzed, and the half cycle working mode is divided into three modes: the t0-t2 period is defined as mode 1, the t2-t3 period is defined as mode 2, and the t3-t4 period is defined as mode 3;

[0066] like Figure 2 As shown, the circuit of mode 1 (t0-t2) stage: before t0, the power tubes S1, S3 and S6 are turned off, the power tubes S2, S4 and S5 are turned on, the input inductor current and the secondary leakage inductance current match, the power supply V1, the input inductor L and the transformer leakage inductance L S The power tube S1 is turned on at time t0, the input inductor current and the transformer leakage inductance current do not match, the circuits on both sides of the transformer do not transfer energy, the input inductor L is charged, and the transformer leakage inductance L S Discharge first and then charge. The transformer leakage inductance voltage VLS (t) and leakage inductance current i LS The expression of (t):

[0067] V LS (t)=V2,t0≤t≤t2 (1)

[0068]

[0069] like Figure 3 As shown, in the mode 2 (t2-t3) stage circuit: the power tubes S2, S4 and S6 are turned off, the power tubes S1, S3 and S5 are turned on, the input inductor current matches the secondary leakage inductance current, and the power supply V1 supplies current to the input inductor L and the transformer leakage inductance L. S , the load voltage V2 is stabilized by the discharge of capacitor C2. The transformer leakage inductance voltage V LS (t) and leakage inductance current i LS The expression of (t):

[0070]

[0071] like Figure 4 As shown, the circuit in mode 3 (t3-t4) is as follows: the power tubes S2, S4 and S5 are turned off, the power tubes S1, S3 and S6 are turned on, the input inductor current matches the secondary leakage inductance current, the circuits on both sides of the transformer transfer energy, the power supply V1, the input inductor L and the transformer leakage inductance L S Discharge, and transfer energy to the load together. The transformer leakage inductance voltage V LS (t) and leakage inductance current i LS The expression of (t):

[0072]

[0073] Furthermore, according to equations (2), (4) and (6), the transformer leakage inductance instantaneous current i of the converter at time t2, t3 and t4 is calculated respectively: LS (t2),i LS (t3) and i LS The expression of (t4) is:

[0074]

[0075] Where: T S is the switching period of the power tube of the converter;

[0076] Step 2: Based on the principle that the average value of the input inductor current is the same as the converter input current, derive the input inductor current expression at time t0;

[0077] Considering that the input inductor current change rates of mode 1 and mode 2 are similar, the change rates of the inductor currents of the two modes are approximately the same to simplify the calculation; the input inductor current expression of the t0-t3 converter is:

[0078]

[0079] Where: P is the transmission power of the converter;

[0080] The change in input inductor current from t0 to t3 is:

[0081]

[0082] According to equations (8) and (9), the input inductor current expression at time t0 is derived as follows:

[0083]

[0084] Step 3: Based on the symmetry of the transformer leakage inductance current in one cycle and the relationship between the transformer primary and secondary currents, derive the expression of the input inductor current excluding P at time t0;

[0085] The symmetry of the transformer leakage current in one cycle shows that LS (t0) = -i LS (t5), combined with formula (3), derive i LS The expression of (t0) is:

[0086]

[0087] Furthermore, according to the current relationship between the primary and secondary sides of the transformer, L (t0) = -i LS (t5) / n can be derived i L The expression of (t0) is:

[0088]

[0089] Step 4: Solve for the optimal variable combination of D;

[0090] First, we eliminate i from equations (10) and (12). LS (t0), we get a linear equation with two variables including D1 and D2:

[0091]

[0092] Furthermore, the volt-second area balance equation is established based on the input inductor current of mode 1 and mode 2:

[0093]

[0094] Combining equations (13) and (14) yields the expression for D1:

[0095]

[0096] Finally, the mathematical expression of D is calculated from the mathematical relationship between D and D1:

[0097]

[0098] Step 5: Derive the optimal variable combination of D2;

[0099] The discharge time T2 of the input inductor in half a cycle is derived from formula (14):

[0100]

[0101] Furthermore, according to the mathematical relationship between the input inductor current charging and discharging time and cycle:

[0102]

[0103] Combining equations (17) and (18) yields the expression for T1:

[0104]

[0105] Finally, substitute equation (19) into the mathematical relationship D2 = T1 / T S -D1 gives the mathematical expression of D2:

[0106]

[0107] The main waveforms of the converter controlled by the two control variables D and D2 modulated by the present invention are as follows: Figure 5 In this embodiment, Figure 6 This is the simulation waveform of the current push-pull circuit with traditional PWM plus phase shift, such as Figure 6 As shown, the reverse current i of the power tubes S1 and S2 of the traditional PWM plus phase shift ds1 and i ds2 Very large, and the current matching time point of each half cycle is late; Figure 7 The PWM plus phase shift modulation current push-pull circuit simulation waveform of the present invention is as follows: Figure 7 As shown, the reverse current i of power tubes S1 and S2 ds1 and i ds2 The current matching time point in each half cycle is significantly advanced.

[0108] Table 1 Comparison of performance between traditional PWM plus phase shift and PWM plus phase shift modulation

[0109]

[0110] The simulation parameters are shown in Table 1. Under the premise of only changing the duty cycle D of the power tubes S1 and S2 of the primary circuit and the internal phase shift angle D2 between the bridge arms of the secondary full-bridge circuit, the reverse current of the power tubes S1 and S2 of the primary circuit is reduced by 96.73%, and the circulating power is reduced by 99.88%.

[0111] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention, rather than to limit it. Although the present invention has been described in detail with reference to the aforementioned embodiments, those skilled in the art should understand that they can still modify the technical solutions described in the aforementioned embodiments, or replace some or all of the technical features therein with equivalents. However, these modifications or replacements do not cause the essence of the corresponding technical solutions to deviate from the scope of the technical solutions of the embodiments of the present invention.

Claims

1. A method for suppressing circulating current in a current-type push-pull circuit with PWM plus phase shift modulation, characterized in that: The circulating current suppression method adopts PWM plus phase shift modulation, combined with the operating conditions of the converter, by optimizing the duty cycle D of the power tubes S1 and S2 of the primary circuit of the current-type push-pull circuit and the internal phase shift angle D2 between the bridge arms of the secondary full-bridge circuit, so that the converter works in a low circulating current state; The duty cycle of the power tubes S1 and S2 of the primary circuit is defined as D, and the overlapping part of the duty cycle of the power tubes S1 and S2 in half a cycle is defined as D1. The mathematical relationship between D and D1 is D=D1+0.5; S3 and S4 of the secondary full-bridge circuit constitute the first bridge arm, and S5 and S6 constitute the second bridge arm. The duty cycle of the first bridge arm power tubes S3 and S4 and the second bridge arm power tubes S5 and S6 is 0.

5. The internal phase shift angle between the first bridge arm and the second bridge arm circuit is defined as D2, and the power tubes of the same bridge arm are complementary. The charging time of the input inductor in half a cycle is defined as T1, and the discharging time is defined as T2; The optimization calculation steps of the duty cycle D and the inner phase shift angle D2 include: Step 1: Analyze and model the current push-pull circuit with PWM plus phase shift modulation. Using the circuit symmetry of the first half cycle and the second half cycle, divide the half cycle working mode into three modes: define the t0-t2 period as mode 1, the t2-t3 period as mode 2, and the t3-t4 period as mode 3, and derive the transformer leakage inductance instantaneous current expression of the converter; Step 2: Based on the principle that the average value of the input inductor current is the same as the converter input current, derive the expression of the input inductor current including the converter transmission power P at time t0; Step 3: Using the transformer leakage inductance instantaneous current expression of the converter obtained in step 1, derive the expression of the input inductance current excluding the converter transmission power P at time t0 based on the symmetry of the transformer leakage inductance current in one cycle and the relationship between the transformer primary and secondary currents; Step 4: Use the input inductor current obtained in steps 2 and 3 to eliminate the instantaneous current of the transformer leakage inductance at time t0, establish the volt-second area balance equation based on the input inductor current, and solve the optimal variable combination of the duty cycle D; Step 5: Use the volt-second area balance equation in step 4 to derive the discharge time T2 of the input inductor within half a cycle, and derive the optimized variable combination of the internal phase shift angle D2 based on the mathematical relationship between the input inductor current charging and discharging time and cycle.

2. The method for suppressing circulating current in a current-type push-pull circuit with PWM plus phase shift modulation according to claim 1, characterized in that: In step 1, the instantaneous current i of the transformer leakage inductance at t2, t3, and t4 LS (t2),i LS (t3) and i LS The specific calculation process of (t4) expression is as follows: Step 1-1: According to Kirchhoff's law and Thevenin equivalent law, derive the transformer leakage inductance voltage V of the converter in the three modes respectively. LS The expression of (t) is: Where: V1, V2 are the voltages across the converter, n is the primary-to-secondary turns ratio of the transformer, L is the input inductance, L S is the transformer leakage inductance; Step 1-2: According to formula (1), derive the transformer leakage inductance current i of the converter in the three modes respectively: LS The expression of (t) is: Where: i LS (t2),i LS (t3) and i LS (t4) are the instantaneous currents of the transformer leakage inductance at t2, t3 and t4 respectively; Step 1-3: Calculate the transformer leakage inductance instantaneous current i of the converter at t2, t3 and t4 respectively according to formula (2): LS (t2),i LS (t3) and i LS The expression of (t4) is: Where: T S is the switching period of the power tube of the converter.

3. The method for suppressing circulating current in a current-type push-pull circuit with PWM plus phase shift modulation according to claim 1, characterized in that: The specific process of calculating the expression of the input inductor current including the converter transmission power P at time t0 in step 2 is as follows: Step 2-1: Since the input inductor current change rates of mode 1 and mode 2 are similar, the change rates of the inductor currents of the two modes are approximately the same to simplify the calculation; the input inductor current expression of the t0-t3 converter is: Where: P is the transmission power of the converter; Step 2-2: Based on the principle that the average value of the input inductor current is the same as the converter input current, derive the input inductor current expression at time t0; The change in input inductor current from t0 to t3 is: According to equations (4) and (5), the input inductor current expression at time t0 is derived as follows:

4. The method for suppressing circulating current in a current-type push-pull circuit with PWM plus phase shift modulation according to claim 1, characterized in that: The specific calculation process of the expression of the input inductor current excluding the converter transmission power P at time t0 in step 3 is as follows: Step 3-1: From the symmetry of the transformer leakage current in one cycle, we know that LS (t0) = -i LS (t5), combined with formula (3), derive i LS The expression of (t0) is: Step 3-2: According to the relationship between the primary and secondary currents of the transformer L (t0) = -i LS (t5) / n can be derived i L The expression of (t0) is:

5. The method for suppressing circulating current in a current-type push-pull circuit with PWM plus phase shift modulation according to claim 1, characterized in that: The specific process of calculating the optimal variable combination for solving the control variable D in step 4 is as follows: Step 4-1: Eliminate i from equations (6) and (8) LS (t0), we get a linear equation with two variables including D1 and D2: Step 4-2: Establish the volt-second area balance equation based on the input inductor current of mode 1 and mode 2: Step 4-3: Combining equations (9) and (10) yields the expression for D1: Step 4-4: The mathematical expression of D is calculated from the mathematical relationship between D and D1:

6. The method for suppressing circulating current in a current-type push-pull circuit with PWM plus phase shift modulation according to claim 1, characterized in that: The specific process of calculating the optimized variable combination for solving the control variable D2 in step 5 is as follows: Step 5-1: From equation (10), the discharge time T2 of the input inductor in half a cycle is derived as: Step 5-2: Based on the mathematical relationship between the input inductor current charge and discharge time and cycle: Step 5-3: Combining equations (13) and (14) yields the expression for T1: Step 5-4: Substitute equation (15) into the mathematical relationship D2 = T1 / T S -D1 gives the mathematical expression of D2:

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