A PWM plus phase-shift modulation current-mode push-pull circuit circulating current suppression method

Abstract

The application discloses a PWM plus phase shift modulation current mode push-pull circuit circulating current suppression method, adopts PWM plus phase shift modulation, combines the operating condition of a converter, optimizes the duty cycle D of power tubes S1 and S2 of a primary circuit of the current mode push-pull circuit and the internal phase shift angle D2 between bridge arms of a secondary full-bridge circuit, and makes the converter work in a low circulating current state. The application significantly reduces the reverse current of the power tubes S1 and S2 of the primary circuit, retains the soft switching performance of the power tubes S1 and S2 of the primary circuit, solves the voltage peak problem of the power tubes of the traditional current mode converter, saves the active clamping circuit or passive buffer for suppressing the voltage peak, controls the converter through two control variables D and D2 obtained through modulation, optimizes the current matching time, significantly reduces the circulating current power, and effectively improves the efficiency of the circuit.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of direct current converter, and particularly relates to a PWM plus phase-shift modulation current-mode push-pull circuit circulating current suppression method. BACKGROUND

[0002] Current-fed push-pull (CFPP) converters have been widely used in renewable energy power supply systems due to their low input current ripple, high step-up ratio, galvanic isolation, and simple structure. However, there is an inherent problem in this converter, i.e., the current through the input inductor and the transformer series inductor do not match for a long time, which will cause serious voltage spikes and circulating current. Circulating current makes multiple energy exchanges occur in the converter, resulting in additional losses and reducing the efficiency of the converter. Voltage spikes make the power tube bear large voltage stress, which easily leads to device breakdown or thermal damage. To solve this problem, a single PWM control strategy for CFPP converters was reported in the paper "Naturally Clamped Zero-Current Commutated Soft-Switching Current-Fed Push-Pull DC / DC Converter: Analysis, Design, and Experimental Results" in IEEE Transactions on Power Electronics, Vol. 30, No. 3, 2015, which does not require active clamp circuits or passive snubbers, and solves the problem of voltage spikes when the power tube of the primary side circuit is turned off. However, the converter using this method has a long power tube current flow time in the primary side circuit before the current matches. When the input voltage decreases or the converter transmits less power, the reverse current of the power tube in the primary side circuit is large, and there is a problem of large circulating current. To further reduce circulating current, an improved modulation scheme of current-fed bidirectional DC-DC converters for loss reduction was reported in the paper "An Improved Modulation Scheme of Current-Fed Bidirectional DC-DC Converters For Loss Reduction" in IEEE Transactions on Power Electronics, Vol. 33, No. 5, 2018, which uses more control degrees of freedom to obtain better performance, but the modulation becomes complex. To reduce the complexity of modulation, a dual PWM control strategy for current-fed dual-active-bridge (DAB) converters was reported in the paper "Decoupled Dual-PWM Control for Naturally Commutated Current-Fed Dual-Active-Bridge DC / DC Converter" in IEEE Journal of Emerging and Selected Topics in Power Electronics, Vol. 8, No. 4, 2020, which reduces the reverse current of the power tube in the primary side circuit, thereby reducing the circulating current.However, the method causes input inductance current discontinuity, leads to voltage and current oscillation of the converter, affects system stability, is not conducive to realizing soft switching, and increases converter loss. SUMMARY

[0003] The application aims to solve the technical problems described above, and provides a PWM plus phase-shift modulation current-mode push-pull circuit circulating current suppression method, which can solve the problems of voltage peak of power tube and excessive reverse current of power tube in traditional current-mode converters and significantly reduce circulating current.

[0004] The application adopts the following technical solutions to achieve the above-mentioned technical effects:

[0005] The PWM plus phase-shift modulation current-mode push-pull circuit circulating current suppression method adopts PWM plus phase-shift modulation, combines the operating conditions of the converter, optimizes the duty cycle D of the power tubes S1 and S2 of the primary circuit of the current-mode push-pull circuit, and the internal phase-shift angle D2 between the bridge arms of the secondary full-bridge circuit, so that the converter works in a low circulating current state.

[0006] The duty cycle of the power tubes S1 and S2 of the primary circuit is defined as D, the overlap part of the duty cycle of the power tubes S1 and S2 in a half cycle is defined as D1, and the mathematical relationship between D and D1 is D=D1+0.5; the S3 and S4 of the secondary full-bridge circuit form a first bridge arm, the S5 and S6 form a second bridge arm, the duty cycle of the power tubes S3 and S4 of the first bridge arm and the power tubes S5 and S6 of the second bridge arm is 0.5, the internal phase-shift angle between the first bridge arm and the second bridge arm circuit is defined as D2, and the power tubes of the same bridge arm are complementary conduction; the charging time of the input inductance in a half cycle is defined as T1, and the discharging time is defined as T2.

[0007] The optimization calculation steps of the control variables D and D2 include:

[0008] Step 1: Derive the expression of the instantaneous current of the transformer leakage inductance of the converter;

[0009] The current-mode push-pull circuit adopts PWM plus phase-shift modulation, and the circuit is symmetrical in the first half cycle and the second half cycle, so only the operating mode of the first half cycle is analyzed, and the half cycle operating mode is divided into three modes: the t0-t2 period is defined as mode 1, the t2-t3 period is defined as mode 2, and the t3-t4 period is defined as mode 3; according to Kirchhoff's law and Thevenin's equivalent law, the expressions of the transformer leakage inductance voltage V LS (t) of the converter in the three modes are derived respectively:

[0010]

[0011] In the formula, V1 and V2 are the voltages across the converter, n is the turns ratio of the primary and secondary of the transformer, L is the input inductance, LS Transformer leakage inductance;

[0012] According to formula (1), the transformer leakage inductance currents i LS (t) of the three modes are respectively derived as follows:

[0013]

[0014] Wherein, i LS (t2), i LS (t3) and i LS (t4) are the instantaneous currents of the transformer leakage inductance at t2, t3 and t4 respectively.

[0015] Further, according to formula (2), the expressions of the transformer leakage inductance instantaneous currents i LS (t2), i LS (t3) and i LS (t4) at t2, t3 and t4 respectively are as follows:

[0016]

[0017] Wherein, T S is the switching period of the power tube of the converter;

[0018] Step 2: The input inductance current expression at t0 is derived according to the principle that the input inductance current average value is the same as the converter input current.

[0019] Considering that the input inductance current change rates of mode 1 and mode 2 are similar, the input inductance current change rates of the two modes are approximately the same for simplifying calculation. The input inductance current expression from t0 to t3 is as follows:

[0020]

[0021] Wherein, P is the transmission power of the converter.

[0022] The input inductance current change amount from t0 to t3 is as follows:

[0023]

[0024] According to formula (4) and (5), the input inductance current expression at t0 is derived as follows:

[0025]

[0026] Step 3: The input inductance current expression at t0 without P is derived according to the symmetry of the transformer leakage inductance current in a period and the relationship between the primary side and secondary side currents of the transformer.

[0027] The symmetry of the leakage current of the transformer in one cycle is known as i LS (t0) = -i LS (t5), combined with equation (3), to derive i LS (t0) is:

[0028]

[0029] Further, according to the relationship between the primary side and the secondary side current of the transformer i L (t0) = -i LS (t5) / n can be derived i L (t0) is:

[0030]

[0031] Step 4: Solve the optimal variable combination of D;

[0032] First, eliminate i LS (t0) from equations (6) and (8) to get a binary first-order equation containing D1 and D2:

[0033]

[0034] Further, according to the input inductance current of mode 1 and mode 2, the volt-second area balance equation is established:

[0035]

[0036] Solving equations (9) and (10) gives the expression of D1:

[0037]

[0038] Finally, the mathematical expression of D is calculated from the mathematical relationship between D and D1:

[0039]

[0040] Step 5: Derive the optimal variable combination of D2;

[0041] The expression of the discharge time T2 of the input inductance in half a cycle is derived from equation (10):

[0042]

[0043] Further, according to the mathematical relationship between the charging and discharging time of the input inductance current and the period:

[0044]

[0045] Solving equations (13) and (14) gives the expression of T1:

[0046]

[0047] Finally, formula (15) is brought into the mathematical relationship D2=T1 / T S The mathematical expression of D2 derived from D1 is:

[0048]

[0049] Compared with the prior art, the present application has the following beneficial effects:

[0050] (1) In the modulation step of the present application, the power tubes S1 and S2 of the primary side circuit of the converter are combined in the working condition without reverse current, and the approximate calculation method is used in the calculation process, so that the reverse current of the power tubes S1 and S2 of the primary side circuit is significantly reduced, and the soft switching performance of the power tubes S1 and S2 of the primary side circuit is retained.

[0051] (2) Based on the two control variables D and D2 obtained by the modulation of the present application, the converter can be controlled, the problem of voltage peak of the power tube of the traditional current-mode converter can be solved, the active clamp circuit or passive buffer for suppressing the voltage peak is saved, and the power density of the converter can be improved.

[0052] (3) Based on the two control variables D and D2 obtained by the modulation of the present application, the converter can be controlled, the mismatch time of the input inductor current and the secondary side current can be reduced, and the circulating current can be reduced. BRIEF DESCRIPTION OF DRAWINGS

[0053] Figure 1 It is a current-mode push-pull circuit topology;

[0054] Figure 2 It is a circuit diagram of the mode 1 (t0-t2) stage;

[0055] Figure 3 It is a circuit diagram of the mode 2 (t2-t3) stage;

[0056] Figure 4 It is a circuit diagram of the mode 3 (t3-t4) stage;

[0057] Figure 5 It is the main waveform of the converter controlled by the two control variables D and D2 obtained by the modulation of the present application;

[0058] Figure 6 It is a simulation waveform diagram of the current-mode push-pull circuit with traditional PWM plus phase shift;

[0059] Figure 7 It is a simulation waveform of the current-mode push-pull circuit with PWM plus phase shift modulation of the present application. DETAILED DESCRIPTION

[0060] The technical solutions in the embodiments of the present application will be clearly and completely described below with reference to the accompanying drawings of the specification. Obviously, the described embodiments are only a part of the embodiments of the present application, rather than all the embodiments of the present application. Based on the embodiments in the present application, all the other embodiments obtained by those skilled in the art without creative work fall within the protection scope of the present application.

[0061] The present application is a PWM plus phase shift modulation current mode push-pull circuit circulating current suppression method, as shown in the current mode push-pull circuit, Figure 1 The circulating current suppression method adopts PWM plus phase shift modulation, combines the operating conditions of the converter, and optimizes the duty cycle D of the primary side circuit power tubes S1 and S2 of the current mode push-pull circuit and the internal phase shift angle D2 between the bridge arms of the secondary side full-bridge circuit, so that the converter works in a low circulating current state.

[0062] The duty cycle of the primary side circuit power tubes S1 and S2 is defined as D, the overlap part of the duty cycle of the power tubes S1 and S2 in a half cycle is defined as D1, and the mathematical relationship between D and D1 is D=D1+0.5; S3 and S4 of the secondary side full-bridge circuit form a first bridge arm, S5 and S6 form a second bridge arm, the duty cycle of the power tubes S3, S4, S5 and S6 of the first bridge arm and the second bridge arm is 0.5, the internal phase shift angle between the first bridge arm and the second bridge arm circuit is defined as D2, and the power tubes of the same bridge arm are complementary conduction; the charging time of the input inductor in a half cycle is defined as T1, and the discharging time is defined as T2.

[0063] The optimization calculation steps of the control variables D and D2 include:

[0064] Step 1: Derive the instantaneous current expression of the transformer leakage inductance of the converter;

[0065] The current mode push-pull circuit adopts PWM plus phase shift modulation, and because the circuits in the first half cycle and the second half cycle are symmetrical, only the operating mode of the first half cycle is analyzed, and the half cycle operating mode is divided into three modes: the t0-t2 period is defined as mode 1, the t2-t3 period is defined as mode 2, and the t3-t4 period is defined as mode 3.

[0066] As shown in Figure 2 The circuit in mode 1 (t0-t2) stage: before t0, the power tubes S1, S3 and S6 are off, the power tubes S2, S4 and S5 are on, the input inductor current and the secondary side leakage inductance current are matched, the power is transmitted to the load through the input inductor L, the transformer leakage inductance L S discharge, and energy is transmitted to the load together. The power tube S1 is turned on at t0, the input inductor current and the transformer leakage inductance current are not matched, the transformer two-side circuit does not transmit energy, the input inductor L is charged, the transformer leakage inductance L S discharge, and energy is transmitted to the load together. The power tube S1 is turned on at t0, the input inductor current and the transformer leakage inductance current are not matched, the transformer two-side circuit does not transmit energy, the input inductor L is charged, the transformer leakage inductance LLS (t) and leakage inductance current i LS The expression for (t):

[0067] V LS (t)=V2,t0≤t≤t2 (1)

[0068]

[0069] like Figure 3 As shown, in the mode 2 (t2-t3) stage circuit: power transistors S2, S4, and S6 are off, power transistors S1, S3, and S5 are on, the input inductor current and the secondary leakage inductor current are matched, and the power supply V1 supplies power to the input inductor L and the transformer leakage inductance L. S The load voltage V2 is stabilized solely by the discharge of capacitor C2. The transformer leakage inductance voltage Vo of the converter... LS (t) and leakage inductance current i LS The expression for (t):

[0070]

[0071] like Figure 4 As shown, the circuit in mode 3 (t3-t4) is as follows: power transistors S2, S4, and S5 are off, power transistors S1, S3, and S6 are on, the input inductor current and the secondary leakage inductor current are matched, energy is transferred between the circuits on both sides of the transformer, and the power supply V1, input inductor L, and transformer leakage inductance L... S Discharge, together transferring energy to the load. The transformer leakage inductance voltage V of the converter. LS (t) and leakage inductance current i LS The expression for (t):

[0072]

[0073] Furthermore, the instantaneous leakage current i of the transformer at times t2, t3, and t4 is calculated according to equations (2), (4), and (6), respectively. LS (t2), i LS (t3) and i LS The expression for (t4) is:

[0074]

[0075] In the formula: T S The switching cycle of the power transistor in the converter;

[0076] Step 2: Based on the principle that the average value of the input inductor current is the same as the converter input current, derive the expression for the input inductor current at time t0;

[0077] Considering that the input inductance current change rates of the mode 1 and the mode 2 are similar, the input inductance current change rates of the two modes are approximately the same for simplifying calculation; the input inductance current expression of the transformer from t0 to t3 is:

[0078]

[0079] In the formula, P is the transmission power of the transformer;

[0080] The input inductance current change amount from t0 to t3 is:

[0081]

[0082] According to the formula (8) and (9), the input inductance current expression of t0 is:

[0083]

[0084] Step 3: According to the symmetry of the leakage inductance current of the transformer in a period and the relationship between the primary side and the secondary side current of the transformer, the expression of the input inductance current of t0 without P is derived;

[0085] According to the symmetry of the leakage inductance current of the transformer in a period, it is known that LS (t0) = -i LS (t5), and the expression of i LS (t0) is derived by combining the formula (3) as:

[0086]

[0087] Further, according to the relationship between the primary side and the secondary side current of the transformer, L (t0) = -i LS (t5) / n, the expression of i L (t0) is derived as:

[0088]

[0089] Step 4: solving the optimization variable combination of D;

[0090] Firstly, i LS (t0) is eliminated from the formula (10) and (12), and a binary first-order equation containing D1 and D2 is obtained:

[0091]

[0092] Further, the volt-second area balance equation is established according to the input inductance current of the mode 1 and the mode 2:

[0093]

[0094] Simultaneous equations (13) and (14) give the D1 expression:

[0095]

[0096] Finally, the mathematical expression of D is calculated from the mathematical relationship of D and D1 as:

[0097]

[0098] Step 5: Derive the optimal variable combination of D2;

[0099] The expression of the discharge time T2 of the input inductor in half a cycle is derived from equation (14):

[0100]

[0101] Further, according to the mathematical relationship of the input inductor current charging and discharging time and period:

[0102]

[0103] Simultaneous equations (17) and (18) give the T1 expression:

[0104]

[0105] Finally, equation (19) is brought into the mathematical relationship D2=T1 / T S -D1 to derive the mathematical expression of D2:

[0106]

[0107] The two control variables D and D2 obtained by the modulation of the application control the main waveforms of the down converter as shown in Figure 5 In this embodiment, Figure 6 The simulation waveform diagram of the current-mode push-pull circuit with traditional PWM plus phase shift is shown in Figure 6 As shown in the figure, the reverse currents i ds1 and i ds2 of the power tubes S1 and S2 are very large, and the current matching time point of each half cycle is late; Figure 7 The simulation waveform of the current-mode push-pull circuit with PWM plus phase shift modulation of the application is shown in Figure 7 As shown in the figure, the reverse currents i ds1 and i ds2 of the power tubes S1 and S2 are significantly reduced, and the current matching time point of each half cycle is significantly advanced.

[0108] Table 1 Comparison of modulation performance between traditional PWM plus phase shift and PWM plus phase shift

[0109]

[0110] The simulation parameters are shown in Table 1. On the premise of only changing the duty cycle D of the power tubes S1 and S2 of the primary side circuit and the internal phase shift angle D2 between the bridge arms of the secondary side full-bridge circuit, the reverse current of the power tubes S1 and S2 of the primary side circuit is reduced by 96.73%, and the circulating current power is reduced by 99.88%.

[0111] Finally, it should be noted that: the above embodiments are only used to illustrate the technical solutions of the present application, but not to limit them; although the present application has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that: it can still modify the technical solutions recorded in the foregoing embodiments, or make equivalent replacement for part or all of the technical features; and these modifications or replacements do not make the essence of the corresponding technical solutions deviate from the scope of the technical solutions of the embodiments of the present application.

Claims

1. A method for suppressing circulating current in a PWM plus phase-shift modulation current-mode push-pull circuit, characterized in that, The circulating current suppression method adopts PWM plus phase shift modulation, combines with the operation condition of the converter, and optimizes the power tube of the primary circuit of the current-mode push-pull circuit S 1、 S 2duty cycle D , and the inner phase shift angle between the bridge arms of the secondary full-bridge circuit D 2, so that the converter works in a low circulating current state; Definition of power tube of primary side circuit S 1、 S 2, D definition of power tube S 1、 S 2, duty cycle of overlap part of half cycle is D 1, D and D 1, mathematical relationship is D = D 1+0.5; power tube of secondary side full-bridge circuit S 3、 S 4 constitutes first bridge arm, S 5、 S 6 constitutes second bridge arm, duty cycle of power tube S 3、 S 4 of first bridge arm and power tube S 5 and S 6 of second bridge arm is 0.5, definition of internal phase shift angle between first bridge arm and second bridge arm circuit is D 2, power tubes of same bridge arm are complementary conduction; definition of charging time length of input inductor in half cycle is T 1, discharging time length is T 2; The duty cycle D , the inner phase angle D The optimization calculation step 2 includes: Step 1: Analysis and modeling of the current-fed push-pull circuit with phase-shifted modulation on PWM, using the symmetry characteristics of the circuit in the first half cycle and the second half cycle, the half cycle working mode is divided into three modes: define t 0- t 2 period is mode 1, t 2- t 3 period is mode 2, t 3- t 4 period is mode 3, and the transformer leakage inductance instantaneous current expression of the converter is derived; Step 2: Derivation based on the principle that the input inductor current average value is the same as the converter input current t 0 time input inductor current contains the expression of the converter transmission power P ; Step 3: Using the transformer leakage inductance instantaneous current expression of the converter obtained in Step 1, the expression of the transformer leakage inductance current is derived according to the symmetry of the transformer leakage inductance current in a period and the relationship between the primary side and secondary side currents of the transformer t The expression of the input inductance current at time 0 does not contain the transmission power of the converter P ​ Step 4: The instantaneous current at time 0 is eliminated by using the input inductor current obtained in Step 2 and Step 3 to cancel the leakage inductance of the transformer t , and the volt-second area balance equation is established according to the input inductor current to solve the optimal combination of duty cycle D ​ Step 5: Derive the discharge time of input inductor in half cycle by using the volt-second area balance equation in step 4 T 2, Derive the internal phase shift angle by the mathematical relationship of input inductor current charge and discharge time and cycle D 2, Optimization variable combination.

2. The method of claim 1, wherein the PWM plus phase shifting modulation current-fed push-pull circuit circulating current suppression method is characterized by, The transformer leakage inductance in step 1 is t 2、 t 3、 t 4 the moment of instantaneous current i LS ( t 2)、 i LS ( t 3) and i LS ( t 4) expression of the calculation process is as follows: Step 1-1: Derivation of the expression of transformer leakage voltage of the converter under three modes according to Kirchhoff's law and Thevenin's equivalent law V LS ( t ) （1） wherein: V 1、 V 2 is the voltage across the transformer, n is the turns ratio of the transformer, L is the input inductance, L S is the leakage inductance of the transformer; Step 1-2: Derivation of the transformer leakage current of the transformer for the three modalities according to equation (1) i LS ( t ) is given by （2） wherein: i LS t 2), i LS t 3) and i LS t 4) are the instantaneous currents at transformer leakage inductances at t 2, t 3 and t 4 instants, respectively.​​​ Step 1-3: Calculate according to formula (2) respectively t 2、 t 3 and t 4 transformer leakage inductance instantaneous current of the transformer i LS ( t 2)、 i LS ( t 3) and i LS ( t 4) expression: （3） In the formulae: T S is the switching period of the power tube of the inverter.

3. The method of claim 2, wherein the PWM plus phase shifting modulation current-fed push-pull circuit circulating current suppression method is characterized by, The step 2 in t The input inductor current at time 0 contains the converter transmitted power P The calculation process of the expression of the input inductor current at time 0 is as follows: Step 2-1: Since the input inductor current variation rates of the mode 1 and the mode 2 are similar, the input inductor current variation rates of the two modes are approximately the same for simplifying the calculation; t 0- t 3The transformer input inductor current expression is: （4） In the formulae: P is the transmission power of the transponder; Step 2-2: Derivation based on the principle that the input inductor current average value is the same as the converter input current t 0 time input inductor current expression; From t 0- t The amount of change in the input inductor current of the third embodiment is: （5） The expressions of the input inductor current at time 0 are: t The input inductor current expression at time 0 is: （6）。 4. The method of claim 2, wherein the PWM plus phase shifting modulation current-fed push-pull circuit circulating current suppression method is characterized by, The step 3 in t The input inductor current at time 0 does not contain the converter transmitted power P The calculation process of the expression of the input inductor current is as follows: Step 3-1: From the symmetry of the leakage current of the transformer over a cycle i LS ( t 0)=- i LS ( t 5), combined with equation (3), the expression for i LS ( t 0) is （7） Step 3-2: Relationship between primary and secondary side currents of the transformer i L ( t 0)=- i LS ( t 5) / n It can be derived that i L ( t 0) is （8）。 5. The method of claim 2, wherein the PWM plus phase shifting modulation current-fed push-pull circuit circulating current suppression method is characterized by, The step 4 solving the control variables D The calculation process of the optimization variable combination is as follows: Step 4-1: Elimination from formula (6) and (8) i LS ( t 0), to obtain a binary linear equation containing D 1 and D 2 （9） Step 4-2: Establishing the volt-second area balance equation according to the input inductor current of modal 1 and modal 2: （10） Step 4-3: Simultaneous equations (9) and (10) give D 1Expression: （11） Step 4-4: Calculating the mathematical relationship from D and D 1 is calculated as D The mathematical expression for is （12）。 6. The method of claim 2, wherein the PWM plus phase shifting modulation current-fed push-pull circuit circulating current suppression method is characterized by, Solving the control variables in step 5 D The calculation process of the optimization variable combination of step 2 is as follows: Step 5-1: Deriving the discharge time of input inductor over half cycle from equation (10) T 2The expression is: （13） Step 5-2: According to the mathematical relationship of the input inductor current charging and discharging time and period: （14） Step 5-3: Simultaneous equations (13) and (14) give T 1Expression: （15） Step 5-4: Plugging the formula for (15) into the mathematical relationship D 2= T 1 / T S -D 1resulting in D 2 （16）。

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