Bidirectional resonant DC-DC converter and modulation method considering wide voltage range variation

Abstract

This invention discloses a bidirectional resonant DC-DC converter and its modulation method that consider a wide range of voltage variations. The modulation method includes: S1: Considering the wide range of voltage variations, constructing a corresponding operating mode based on the voltage relationship between the power supplies of the bidirectional resonant DC-DC converter; S2: Generating an optimized modulation scheme for the corresponding operating mode based on the equivalent circuit of the bidirectional resonant DC-DC converter; S3: Modulating the bidirectional resonant DC-DC converter based on the optimized modulation scheme to bring it into the corresponding operating mode, while simultaneously utilizing the excitation current flowing through the magnetizing inductor to achieve soft switching of the device. This invention fully considers the wide range of voltage variations in the energy storage battery, utilizes the transformer excitation current to achieve soft switching of the device, enables all switching transistors to achieve zero-voltage turn-on, and only two transistors experience losses during turn-off.

Description

Technical Field

[0001] This invention relates to the field of power electronics technology, and more specifically to a bidirectional resonant DC-DC converter and modulation method that takes into account a wide range of voltage variations. Background Technology

[0002] In recent years, bidirectional isolated DC-DC converters have developed rapidly and are widely used in electric vehicles, DC power supplies, new energy grid connection, and energy storage systems. Currently, the mainstream bidirectional isolated DC-DC converters include dual active bridge DC-DC converters and bidirectional resonant DC-DC converters. Dual active bridge DC-DC converters can achieve flexible energy adjustment, bidirectional wide voltage range gain, and numerous optimized modulation algorithms, but they also have limitations, such as high return power, large turn-off current, and narrow soft-switching range.

[0003] A bidirectional resonant DC-DC converter (BRDC) is a power conversion device capable of bidirectional energy transfer between two DC power sources. Its core lies in utilizing the characteristics of a resonant circuit to achieve energy conversion and transfer by controlling the on / off state of a switching transistor. BRDC converters typically consist of a high-frequency transformer, resonant inductor, resonant capacitor, and switching transistors, offering advantages such as compact structure, high efficiency, and flexible control. However, BRDC converters have some limitations, including the inability to achieve bidirectional step-up / step-down conversion and a narrow voltage gain range, which restricts their application scope.

[0004] When operating in reverse, bidirectional resonant DC-DC converters typically only perform a buck function, not a boost function. This limitation restricts their application in situations with large voltage variations. For example, in bidirectional interaction between an electric vehicle and the power grid, if the grid voltage fluctuates significantly while the electric vehicle battery voltage remains relatively stable, the bidirectional resonant DC-DC converter may be unable to boost the low grid voltage to the high voltage required by the battery when transferring power from the grid to the electric vehicle battery, resulting in low energy transfer efficiency or failure to complete the transfer.

[0005] Bidirectional resonant DC-DC converters typically have a narrow voltage gain range, which limits their application in wide output voltage ranges. A narrow voltage gain range means that the output voltage of a bidirectional resonant DC-DC converter can only vary within a limited range when the input voltage changes, making it unsuitable for applications with high output voltage requirements. For example, in a DC microgrid, the voltage of a supercapacitor may fluctuate over a wide range, and the narrow voltage gain range of the bidirectional resonant DC-DC converter may prevent it from stably converting the supercapacitor's voltage to the voltage range required by the DC microgrid bus.

[0006] Therefore, how to design a modulation method for a bidirectional resonant DC-DC converter that can achieve bidirectional step-up and step-down and has a wider voltage gain range is an urgent technical problem to be solved. Summary of the Invention

[0007] To address the shortcomings of the existing technologies, the technical problem this invention aims to solve is: how to provide a bidirectional resonant DC-DC converter and modulation method that considers a wide range of voltage variations. This method fully considers the wide range of energy storage battery voltage variations and formulates different optimized modulation schemes under different primary and secondary power supply voltage conditions. Simultaneously, it utilizes the transformer's magnetizing current to achieve soft switching of the devices, and expands the soft switching range through optimized modulation schemes, reducing converter operating losses and improving overall efficiency. Furthermore, all switching transistors can achieve zero-voltage turn-on, and only two transistors experience losses during turn-off, while the others achieve zero-current turn-off, resulting in low overall losses.

[0008] To solve the above-mentioned technical problems, the present invention adopts the following technical solution:

[0009] Considering bidirectional resonant DC-DC converters with a wide voltage range, including:

[0010] The primary-side half-bridge circuit is set at power supply U. s The primary side includes a half-bridge consisting of a primary upper switch S1 and a primary lower switch S2 connected in series.

[0011] The secondary-side full-bridge circuit is located at power supply U. p The secondary side includes an H-bridge consisting of a series-connected upper secondary switch Q1 and lower secondary switch Q2, and a series-connected upper secondary switch Q3 and lower secondary switch Q4 connected in parallel.

[0012] A transformer is connected between the primary half-bridge circuit and the secondary full-bridge circuit; the magnetizing inductance of the transformer is L. m ;

[0013] A resonant circuit, connected between the primary half-bridge circuit and the transformer, includes a resonant inductor L connected in series. r and resonant capacitor C r .

[0014] A modulation method for a bidirectional resonant DC-DC converter considering a wide voltage range variation, implemented based on the bidirectional resonant DC-DC converter of this invention, includes:

[0015] S1: Considering a wide range of voltage variations, based on the power supply U of the bidirectional resonant DC-DC converter. s and U p The voltage relationship between them establishes the corresponding operating mode;

[0016] S2: Based on the equivalent circuit of the bidirectional resonant DC-DC converter, generate an optimized modulation scheme for the corresponding operating mode;

[0017] S3: Modulate the bidirectional resonant DC-DC converter based on the optimized modulation scheme to enable it to enter the corresponding working mode, and at the same time use the excitation current flowing through the excitation inductor to realize the soft switching of the device.

[0018] Preferably, in step S1, the operating mode includes operating mode one, in which the primary half-bridge circuit forward boosts and transmits power to the secondary full-bridge circuit.

[0019] Preferably, in step S2, the optimized modulation scheme for working mode one is as follows:

[0020] By using phase shifting, the falling edge of the PWM drive signal of the primary side switch S1 is aligned with the falling edge of the PWM drive signal of the second secondary side switch Q3, and the falling edge of the PWM drive signal of the primary side lower switch S2 is aligned with the falling edge of the PWM drive signal of the second secondary side lower switch Q4; the rising edge of the PWM drive signal of the first secondary side lower switch Q2 lags behind the rising edge of the PWM drive signal of the primary side switch S1, and the rising edge of the PWM drive signal of the first secondary side switch Q1 lags behind the rising edge of the PWM drive signal of the primary side lower switch S2.

[0021] After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on.

[0022] After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on.

[0023] When the upper switch Q1 on the first secondary side is turned off, a turn-off loss is generated, and the lower switch Q2 on the first secondary side achieves zero-voltage turn-on; when the lower switch Q2 on the first secondary side is turned off, a turn-off loss is generated, and the upper switch Q1 on the first secondary side achieves zero-voltage turn-on.

[0024] Preferably, in step S1, the operating mode includes operating mode two, in which the primary half-bridge circuit transmits power to the secondary full-bridge circuit via forward step-down.

[0025] Preferably, in step S2, the optimized modulation scheme for working mode two is as follows:

[0026] By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1; the falling edge of the PWM drive signal of the second secondary side upper switch Q3 lags behind the falling edge of the PWM drive signal of the primary side upper switch S1, and the falling edge of the PWM drive signal of the second secondary side lower switch Q4 lags behind the falling edge of the PWM drive signal of the primary side lower switch S2.

[0027] After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on.

[0028] After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on.

[0029] After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on.

[0030] After the first secondary switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary switch Q2 is turned on in advance, achieving zero-voltage turn-on.

[0031] Preferably, in step S1, the operating mode includes operating mode three, in which the secondary-side full-bridge circuit reverses the voltage to transfer power to the primary-side half-bridge circuit.

[0032] Preferably, in step S2, the optimized modulation scheme for operating mode three is as follows:

[0033] By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1; the falling edge of the PWM drive signal of the second secondary side upper switch Q3 leads the falling edge of the PWM drive signal of the primary side upper switch S1, and the falling edge of the PWM drive signal of the second secondary side lower switch Q4 leads the falling edge of the PWM drive signal of the primary side lower switch S2.

[0034] After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on.

[0035] After the first secondary side upper switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side lower switch Q2 is turned on in advance, achieving zero-voltage turn-on;

[0036] After the upper switch Q3 on the second auxiliary side is turned off, a turn-off loss is generated, and the lower switch Q4 on the second auxiliary side achieves zero-voltage turn-on; after the lower switch Q4 on the second auxiliary side is turned off, a turn-off loss is generated, and the upper switch Q3 on the second auxiliary side achieves zero-voltage turn-on.

[0037] Preferably, in step S1, the operating mode includes operating mode four, in which the secondary-side full-bridge circuit reverses the voltage to transfer power to the primary-side half-bridge circuit.

[0038] Preferably, in step S2, the optimized modulation scheme for operating mode four is as follows:

[0039] By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2; the falling edge of the PWM drive signal of the second secondary side lower switch Q4 lags behind the falling edge of the PWM drive signal of the primary side upper switch S1, and the falling edge of the PWM drive signal of the second secondary side upper switch Q3 lags behind the falling edge of the PWM drive signal of the primary side lower switch S2.

[0040] After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on.

[0041] After the first secondary side upper switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side lower switch Q2 is turned on in advance, achieving zero-voltage turn-on;

[0042] After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on.

[0043] After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on.

[0044] Compared with existing technologies, the bidirectional resonant DC-DC converter and modulation method considering a wide range of voltage variations in this invention have the following advantages:

[0045] The modulation method of this invention fully considers the wide range of voltage variations in the energy storage battery and formulates different optimized modulation schemes under different primary and secondary power supply voltage conditions. Simultaneously, it utilizes the transformer excitation current to achieve soft switching of the devices. By optimizing the modulation scheme, the soft switching range is expanded, reducing converter operating losses and improving overall efficiency. Furthermore, all switches can achieve zero-voltage turn-on, and only two switches experience losses during turn-off, while the others achieve zero-current turn-off, resulting in low overall losses. In addition, the modulation algorithm is similar to ordinary buck / boost modulation, without complex multi-phase shift control. The algorithm and control method are simple. By changing the pulse width of the primary-side switch and the phase shift of the secondary-side switch, a bidirectional resonant DC-DC converter achieves bidirectional wide-range voltage gain and flexible bidirectional energy flow without relying on the high-frequency transformer turns ratio. It also features no backflow power, and the low transformer turns ratio reduces the effective value of the resonant current, resulting in high converter operating efficiency. Attached Figure Description

[0046] To make the objectives, technical solutions, and advantages of the invention clearer, the invention will now be described in further detail with reference to the accompanying drawings, wherein:

[0047] Figure 1 This is a circuit diagram of a bidirectional resonant DC-DC converter.

[0048] Figure 2 This is a timing diagram of the drive state waveform under the working mode.

[0049] Figure 3 The current flow diagram for time t0 to t1 under the working mode is shown.

[0050] Figure 4 The current flow diagram at times t1 to t2 is shown for the working mode.

[0051] Figure 5 The current flow diagram at times t2 to t3 is shown for the working mode.

[0052] Figure 6 The current flow diagram at times t3 to t4 is shown for the working mode.

[0053] Figure 7 This is the timing diagram of the drive state waveform under working mode two.

[0054] Figure 8 This is a diagram showing the current flow direction from time t0 to t1 under operating mode 2.

[0055] Figure 9 This is a diagram showing the current flow direction at times t1 to t2 under operating mode 2.

[0056] Figure 10 This is a diagram showing the current flow direction at times t2 to t3 under operating mode 2.

[0057] Figure 11 This is a diagram showing the current flow direction at times t3 to t4 under operating mode 2.

[0058] Figure 12 This is a diagram showing the current flow direction at times t4 to t5 under operating mode 2.

[0059] Figure 13 This is the timing diagram of the drive state waveform under working mode three.

[0060] Figure 14 This is a diagram showing the current flow direction from time t0 to t1 under operating mode 3.

[0061] Figure 15 This is a diagram showing the current flow direction at times t1 to t2 under operating mode 3.

[0062] Figure 16 This is a diagram showing the current flow direction at times t2 to t3 under operating mode 3.

[0063] Figure 17 This is a diagram showing the current flow direction at times t3 to t4 under operating mode 3.

[0064] Figure 18 This is a timing diagram of the drive state waveform under working mode four.

[0065] Figure 19 The diagram shows the current flow direction at times t0 to t1 under operating mode 4.

[0066] Figure 20 The diagram shows the current flow direction at times t1 to t2 under operating mode 4.

[0067] Figure 21 The diagram shows the current flow direction at times t2 to t3 under operating mode 4.

[0068] Figure 22 The diagram shows the current flow direction at times t3 to t4 under operating mode 4.

[0069] Figure 23 The diagram shows the current flow direction at times t4 to t5 under operating mode 4. Detailed Implementation

[0070] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some, not all, of the embodiments of the present invention. The components of the embodiments of the present invention described and shown in the accompanying drawings can generally be arranged and designed in various different configurations. Therefore, the following detailed description of the embodiments of the present invention provided in the accompanying drawings is not intended to limit the scope of the claimed invention, but only to illustrate selected embodiments of the invention. All other embodiments obtained by those skilled in the art based on the embodiments of the present invention without inventive effort are within the scope of protection of the present invention.

[0071] It should be noted that similar reference numerals and letters in the following figures indicate similar items. Therefore, once an item is defined in one figure, it does not need to be further defined and explained in subsequent figures. In the description of this invention, it should be noted that the terms "center," "upper," "lower," "left," "right," "vertical," "horizontal," "inner," and "outer," etc., indicate the orientation or positional relationship based on the orientation or positional relationship shown in the figures, or the orientation or positional relationship commonly used when the product is in use. They are only for the convenience of describing the invention and simplifying the description, and do not indicate or imply that the device or element referred to must have a specific orientation, or be constructed and operated in a specific orientation. Therefore, they should not be construed as limitations on the invention. Furthermore, the terms "first," "second," and "third," etc., are only used to distinguish descriptions and should not be construed as indicating or implying relative importance. In addition, the terms "horizontal," "vertical," etc., do not mean that the component is required to be absolutely horizontal or suspended, but can be slightly tilted. For example, "horizontal" only means that its direction is more horizontal than "vertical," and does not mean that the structure must be completely horizontal, but can be slightly tilted. In the description of this invention, it should also be noted that, unless otherwise explicitly specified and limited, the terms "set," "install," "connect," and "link" should be interpreted broadly. For example, they can refer to a fixed connection, a detachable connection, or an integral connection; they can refer to a mechanical connection or an electrical connection; they can refer to a direct connection or an indirect connection through an intermediate medium; and they can refer to the internal connection of two components. Those skilled in the art can understand the specific meaning of the above terms in this invention based on the specific circumstances.

[0072] The following detailed explanation illustrates the specific implementation methods:

[0073] Example 1:

[0074] This embodiment discloses a bidirectional resonant DC-DC converter that considers a wide range of voltage variations.

[0075] like Figure 1 As shown, a bidirectional resonant DC-DC converter considering a wide voltage range variation is characterized by comprising:

[0076] The primary-side half-bridge circuit is set at the primary-side power supply U. s The primary side includes a half-bridge consisting of a primary upper switch S1 and a primary lower switch S2 connected in series.

[0077] The secondary-side full-bridge circuit is located at the secondary-side power supply U. p The secondary side includes an H-bridge consisting of a series-connected upper secondary switch Q1 and lower secondary switch Q2, and a series-connected upper secondary switch Q3 and lower secondary switch Q4 connected in parallel.

[0078] A transformer is connected between the primary half-bridge circuit and the secondary full-bridge circuit. The magnetizing inductance of the transformer is L. m The transformer's turns ratio is n (n is a natural number);

[0079] A resonant circuit, connected between the primary half-bridge circuit and the transformer, includes a resonant inductor L connected in series. r and resonant capacitor C r .

[0080] in:

[0081] Power supply U s A voltage stabilizing capacitor C is installed on the original side. S Power supply U p A voltage regulator capacitor C is installed on the secondary side. p .

[0082] The primary-side upper switch S1, primary-side lower switch S2, first secondary-side upper switch Q1, first secondary-side lower switch Q2, second secondary-side upper switch Q3, and second secondary-side lower switch Q4 are respectively equipped with corresponding anti-parallel diodes and parasitic capacitances: D S1 and C S1 D S2 and C S2 D Q1 and C Q1 D Q2 and C S2 D Q3 and C Q3 D Q4 and C Q4 .

[0083] In this embodiment, the power supply U s To power supply U p The transmission is in the forward direction.

[0084] Example 2:

[0085] This embodiment discloses a modulation method for a bidirectional resonant DC-DC converter that considers a wide range of voltage variations, based on the bidirectional resonant DC-DC converter of Embodiment 1.

[0086] Modulation methods for bidirectional resonant DC-DC converters considering wide voltage range variations include:

[0087] S1: Considering a wide range of voltage variations, based on the external power supply U of the bidirectional resonant DC-DC converter... s and U p The voltage relationship between them establishes the corresponding operating mode;

[0088] S2: Based on the equivalent circuit of the bidirectional resonant DC-DC converter, generate an optimized modulation scheme for the corresponding operating mode;

[0089] S3: Modulate the bidirectional resonant DC-DC converter based on the optimized modulation scheme to enable it to enter the corresponding working mode, and at the same time use the excitation current flowing through the excitation inductor to realize the soft switching of the device.

[0090] The modulation method of this invention fully considers the wide range of voltage variations in the energy storage battery and formulates different optimized modulation schemes under different primary and secondary power supply voltage conditions. Simultaneously, it utilizes the transformer excitation current to achieve soft switching of the devices. By optimizing the modulation scheme, the soft switching range is expanded, reducing converter operating losses and improving overall efficiency. Furthermore, all switches can achieve zero-voltage turn-on, and only two switches experience losses during turn-off, while the others achieve zero-current turn-off, resulting in low overall losses. In addition, the modulation algorithm is similar to ordinary buck / boost modulation, without complex multi-phase shift control. The algorithm and control method are simple. By changing the pulse width of the primary-side switch and the phase shift of the secondary-side switch, a bidirectional resonant DC-DC converter achieves bidirectional wide-range voltage gain and flexible bidirectional energy flow without relying on the high-frequency transformer turns ratio. It also features no backflow power, and the low transformer turns ratio reduces the effective value of the resonant current, resulting in high converter operating efficiency.

[0091] To better illustrate the technical solution of the present invention, this embodiment is described in the following parts.

[0092] I. Working Mode 1

[0093] In this embodiment, the operating mode includes operating mode one, where power is transferred from the primary-side half-bridge circuit to the secondary-side full-bridge circuit via forward boost. In operating mode one, the secondary-side battery voltage increases, U s <2*nU p .

[0094] The optimized modulation scheme for working mode one is as follows:

[0095] The duty cycle of the PWM drive signal controlling the secondary-side switches Q1 to Q4 is 0.5. In the secondary-side full-bridge circuit, the upper and lower switches of the same bridge arm are complementary and conduct. The duty cycles of the PWM drive signals of the primary-side upper switch S1 and the primary-side lower switch S2 are equal and less than 0.5.

[0096] By using phase shifting, the falling edge of the PWM drive signal of the primary side switch S1 is aligned with the falling edge of the PWM drive signal of the second secondary side switch Q3, and the falling edge of the PWM drive signal of the primary side lower switch S2 is aligned with the falling edge of the PWM drive signal of the second secondary side lower switch Q4. The rising edge of the PWM drive signal of the first secondary side lower switch Q2 lags behind the rising edge of the PWM drive signal of the primary side switch S1, and the rising edge of the PWM drive signal of the first secondary side switch Q1 lags behind the rising edge of the PWM drive signal of the primary side lower switch S2.

[0097] After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on.

[0098] After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on.

[0099] When the upper switch Q1 on the first secondary side is turned off, a turn-off loss is generated, and the lower switch Q2 on the first secondary side achieves zero-voltage turn-on; when the lower switch Q2 on the first secondary side is turned off, a turn-off loss is generated, and the upper switch Q1 on the first secondary side achieves zero-voltage turn-on.

[0100] Figure 2 The PWM drive waveform, current flowing through the switching transistors, and resonant inductor L are shown in the forward boost power transfer mode from the primary half-bridge circuit to the secondary full-bridge circuit. r Resonant current i Lr .

[0101] In operating mode one, the bidirectional resonant DC-DC converter exhibits the following mode patterns during half a PWM control cycle (t0~t4): Figures 3 to 6 As shown.

[0102] like Figure 3 Description, Mode 1 [t0, t1]: At time t0, switches S1, Q1, and Q3 are turned on, and inductor L... r The resonant current i Lr The following expression exists:

[0103]

[0104] like Figure 4Description, Mode 2 [t1, t2]: At time t1, switches S1, Q2, and Q3 are turned on, with Q2 turning on at zero voltage, and inductor L... r The resonant current i Lr The following expression exists:

[0105]

[0106] like Figure 5 Description, Mode 3 [t2, t3]: At time t2, switch Q3 is turned off, where Q3 is turned off with near-zero current.

[0107] like Figure 6 Description, Mode 4 [t3, t4]: At time t3, switches Q2 and Q4 are turned on, with Q4 turning on at zero voltage. The current flowing through Q2 and Q4 is the magnetizing current i. Lm .

[0108] II. Working Mode Two

[0109] In this embodiment, the operating mode includes operating mode two, where power is transferred from the primary-side half-bridge circuit to the secondary-side full-bridge circuit via forward voltage reduction. In operating mode two, the secondary-side battery voltage decreases, U s ≥2*nU p .

[0110] The optimized modulation scheme for working mode two is as follows:

[0111] The duty cycle of the PWM drive signal controlling the secondary-side switches Q1 to Q4 is 0.5. In the secondary-side full-bridge circuit, the upper and lower switches of the same bridge arm are complementary and conduct. The duty cycles of the PWM drive signals of the primary-side upper switch S1 and the primary-side lower switch S2 are equal and less than 0.5.

[0112] By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1; the falling edge of the PWM drive signal of the second secondary side upper switch Q3 lags behind the falling edge of the PWM drive signal of the primary side upper switch S1, and the falling edge of the PWM drive signal of the second secondary side lower switch Q4 lags behind the falling edge of the PWM drive signal of the primary side lower switch S2.

[0113] After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on.

[0114] After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on.

[0115] After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on.

[0116] After the first secondary switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary switch Q2 is turned on in advance, achieving zero-voltage turn-on.

[0117] Figure 7 The PWM drive waveform, current flowing through the switching transistors, and resonant inductor L are shown in the forward buck power transfer mode from the primary half-bridge circuit to the secondary full-bridge circuit. r Resonant current i Lr .

[0118] In operating mode two, the bidirectional resonant DC-DC converter's mode during half a PWM control cycle (t0~t5) is as follows: Figures 8 to 12 As shown.

[0119] like Figure 8 Description, Mode 1 [t0, t1]: At time t0, switches S1, Q2, and Q3 are turned on, and inductor L... r The resonant current i Lr The following expression exists:

[0120]

[0121] like Figure 9 Description, Mode 2 [t1, t2]: At time t1, switches Q2 and Q3 are turned on, switch S1 is turned off, and inductor L... r The resonant current i Lr The following expression exists:

[0122]

[0123] like Figure 10 Description, Mode 3 [t2, t3]: At time t2, switch Q2 is turned on and switch Q3 is turned off.

[0124] like Figure 11 Description, Mode 4 [t3, t4]: At time t3, switches Q2 and Q4 are turned on, with Q4 turning on at zero voltage. The current flowing through Q2 and Q4 is the magnetizing current i. Lm .

[0125] like Figure 12 Description, Mode 5 [t4, t5]: At time t4, switch Q4 is turned on and switch Q2 is turned off.

[0126] III. Working Mode Three

[0127] In this embodiment, the operating mode includes operating mode three, where the secondary-side full-bridge circuit reverses voltage to transfer power to the primary-side half-bridge circuit via a step-down converter. In operating mode three, the secondary-side battery voltage increases, 2*nU p ≥U s .

[0128] The optimized modulation scheme for working mode three is as follows:

[0129] The duty cycle of the PWM drive signal controlling the secondary-side switches Q1 to Q4 is 0.5. In the secondary-side full-bridge circuit, the upper and lower switches of the same bridge arm are complementary and conduct. The duty cycles of the PWM drive signals of the primary-side upper switch S1 and the primary-side lower switch S2 are equal and less than 0.5.

[0130] By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1; the falling edge of the PWM drive signal of the second secondary side upper switch Q3 leads the falling edge of the PWM drive signal of the primary side upper switch S1, and the falling edge of the PWM drive signal of the second secondary side lower switch Q4 leads the falling edge of the PWM drive signal of the primary side lower switch S2.

[0131] After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on.

[0132] After the first secondary side upper switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side lower switch Q2 is turned on in advance, achieving zero-voltage turn-on;

[0133] After the upper switch Q3 on the second auxiliary side is turned off, a turn-off loss is generated, and the lower switch Q4 on the second auxiliary side achieves zero-voltage turn-on; after the lower switch Q4 on the second auxiliary side is turned off, a turn-off loss is generated, and the upper switch Q3 on the second auxiliary side achieves zero-voltage turn-on.

[0134] Figure 13 The PWM drive waveform, current flowing through the switching transistors, and resonant inductor L are shown in the reverse buck power transfer mode from the secondary-side full-bridge circuit to the primary-side half-bridge circuit. r Resonant current i Lr .

[0135] In operating mode three, the bidirectional resonant DC-DC converter's mode during half a PWM control cycle (t0~t4) is as follows: Figures 14 to 17 As shown.

[0136] like Figure 14Description, Mode 1 [t0, t1]: At time t0, switches S1, Q2, and Q3 are turned on, and inductor L... r The resonant current i Lr The following expression exists:

[0137]

[0138] like Figure 15 Description, Mode 2 [t1, t2]: At time t1, switches S1, Q2, and Q4 are turned on, switch Q3 is turned off, and Q4 is turned on with zero voltage. Inductor L r The resonant current i Lr The following expression exists:

[0139]

[0140] like Figure 16 Description, Mode 3 [t2, t3]: At time t2, switch S1 is turned off, and switches Q2 and Q4 are turned on. The current flowing through Q2 and Q4 is the magnetizing current i. Lm .

[0141] like Figure 17 Description, Mode 4 [t3, t4]: At time t3, switch Q2 is turned off and switch Q4 is turned on.

[0142] IV. Working Mode Four

[0143] In this embodiment, the operating mode includes operating mode four, where the secondary-side full-bridge circuit reverses and transfers power to the primary-side half-bridge circuit via a boost converter. In operating mode four, the secondary-side battery voltage decreases, and 2*nU... p s .

[0144] The optimized modulation scheme for working mode four is as follows:

[0145] The duty cycle of the PWM drive signal controlling the secondary-side switches Q1 to Q4 is 0.5. In the secondary-side full-bridge circuit, the upper and lower switches of the same bridge arm are complementary and conduct. The duty cycles of the PWM drive signals of the primary-side upper switch S1 and the primary-side lower switch S2 are equal and less than 0.5.

[0146] By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2; the falling edge of the PWM drive signal of the second secondary side lower switch Q4 lags behind the falling edge of the PWM drive signal of the primary side upper switch S1, and the falling edge of the PWM drive signal of the second secondary side upper switch Q3 lags behind the falling edge of the PWM drive signal of the primary side lower switch S2.​

[0147] After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on.

[0148] After the first secondary side upper switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side lower switch Q2 is turned on in advance, achieving zero-voltage turn-on;

[0149] After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on.

[0150] After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on.

[0151] Figure 18 The PWM drive waveform, current flowing through the switching transistors, and resonant inductor L are shown in the reverse boost power transfer mode from the secondary-side full-bridge circuit to the primary-side half-bridge circuit. r Resonant current i Lr .

[0152] In operating mode four, the bidirectional resonant DC-DC converter exhibits the following modes during half a PWM control cycle (t0~t5): Figures 19 to 23 As shown.

[0153] like Figure 19 Description, Mode 1 [t0, t1]: At time t0, switches S2, Q2, and Q3 are turned on, and inductor L... r The resonant current i Lr The following expression exists:

[0154]

[0155] like Figure 20 Description, Mode 2 [t1, t2]: At time t1, switches Q2 and Q3 are on, switch S2 is off, and inductor L... r The resonant current i Lr The following expression exists:

[0156]

[0157]

[0158] like Figure 21 Description, Mode 3 [t2, t3]: At time t2, switch Q2 is turned on and switch Q3 is turned off.

[0159] like Figure 22Description, Mode 4 [t3, t4]: At time t3, switches Q2 and Q4 are turned on, with Q4 turning on at zero voltage. The current flowing through Q2 and Q4 is the magnetizing current i. Lm .

[0160] like Figure 23 Description, Mode 5 [t4, t5]: At time t4, switch Q4 is turned on and switch Q2 is turned off.

[0161] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and not to limit the technical solutions. Those skilled in the art should understand that any modifications or equivalent substitutions to the technical solutions of the present invention without departing from the spirit and scope of the present invention should be covered within the scope of the claims of the present invention.

Claims

1. A modulation method for a bidirectional resonant DC-DC converter considering a wide voltage range variation, characterized in that: Based on a bidirectional resonant DC-DC converter; the bidirectional resonant DC-DC converter includes: a primary-side half-bridge circuit, located at power supply U. s The primary side circuit includes a half-bridge consisting of a primary-side upper switch S1 and a primary-side lower switch S2 connected in series; the secondary-side full-bridge circuit is located at power supply U. p The secondary side circuit includes an H-bridge consisting of a series-connected upper secondary switch Q1 and a series-connected lower secondary switch Q2, and a series-connected upper secondary switch Q3 and a series-connected lower secondary switch Q4 connected in parallel; a transformer is connected between the primary half-bridge circuit and the secondary full-bridge circuit; the magnetizing inductance of the transformer is L. m The resonant circuit, connected between the primary half-bridge circuit and the transformer, includes a series resonant inductor L. r and resonant capacitor C r ; The method includes: S1: Considering a wide range of voltage variations, based on the power supply U of the bidirectional resonant DC-DC converter. s and U p The voltage relationship between them establishes the corresponding operating mode; S2: Based on the equivalent circuit of the bidirectional resonant DC-DC converter, generate an optimized modulation scheme for the corresponding operating mode; S3: Modulate the bidirectional resonant DC-DC converter based on the optimized modulation scheme to enable it to enter the corresponding working mode, and at the same time use the excitation current flowing through the excitation inductor to realize the soft switching of the device. In step S1, the operating modes include operating mode one, in which the primary half-bridge circuit forward boosts and transfers power to the secondary full-bridge circuit. In step S2, the optimized modulation scheme for operating mode one is as follows: By using phase shifting, the falling edge of the PWM drive signal of the primary side switch S1 is aligned with the falling edge of the PWM drive signal of the second secondary side switch Q3, and the falling edge of the PWM drive signal of the primary side lower switch S2 is aligned with the falling edge of the PWM drive signal of the second secondary side lower switch Q4; the rising edge of the PWM drive signal of the first secondary side lower switch Q2 lags behind the rising edge of the PWM drive signal of the primary side switch S1, and the rising edge of the PWM drive signal of the first secondary side switch Q1 lags behind the rising edge of the PWM drive signal of the primary side lower switch S2. After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on. After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on. When the upper switch Q1 on the first secondary side is turned off, a turn-off loss is generated, and the lower switch Q2 on the first secondary side achieves zero-voltage turn-on; when the lower switch Q2 on the first secondary side is turned off, a turn-off loss is generated, and the upper switch Q1 on the first secondary side achieves zero-voltage turn-on.

2. The modulation method for a bidirectional resonant DC-DC converter considering a wide range of voltage variations as described in claim 1, characterized in that: In step S1, the operating mode includes operating mode two, in which the primary half-bridge circuit transmits power to the secondary full-bridge circuit via forward step-down.

3. The modulation method for a bidirectional resonant DC-DC converter considering a wide range of voltage variations as described in claim 2, characterized in that: In step S2, the optimized modulation scheme for operating mode two is as follows: By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1. The falling edge of the PWM drive signal of the second secondary switch Q3 lags behind the falling edge of the PWM drive signal of the primary switch S1, and the falling edge of the PWM drive signal of the second secondary switch Q4 lags behind the falling edge of the PWM drive signal of the primary switch S2. After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on. After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on. After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on. After the first secondary switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary switch Q2 is turned on in advance, achieving zero-voltage turn-on.

4. The modulation method for a bidirectional resonant DC-DC converter considering a wide range of voltage variations as described in claim 1, characterized in that: In step S1, the operating mode includes operating mode three, which involves reverse buck power transfer from the secondary full-bridge circuit to the primary half-bridge circuit.

5. The modulation method for a bidirectional resonant DC-DC converter considering a wide range of voltage variations as described in claim 4, characterized in that: In step S2, the optimized modulation scheme for operating mode three is as follows: By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1. The falling edge of the PWM drive signal of the second secondary switch Q3 leads the falling edge of the PWM drive signal of the primary switch S1, and the falling edge of the PWM drive signal of the second secondary switch Q4 leads the falling edge of the PWM drive signal of the primary switch S2. After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on. After the first secondary side upper switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side lower switch Q2 is turned on in advance, achieving zero-voltage turn-on; After the upper switch Q3 on the second auxiliary side is turned off, a turn-off loss is generated, and the lower switch Q4 on the second auxiliary side achieves zero-voltage turn-on; after the lower switch Q4 on the second auxiliary side is turned off, a turn-off loss is generated, and the upper switch Q3 on the second auxiliary side achieves zero-voltage turn-on.

6. The modulation method for a bidirectional resonant DC-DC converter considering a wide range of voltage variations as described in claim 1, characterized in that: In step S1, the operating mode includes operating mode four, in which the secondary full-bridge circuit reverses the voltage to transfer power to the primary half-bridge circuit.

7. The modulation method for a bidirectional resonant DC-DC converter considering a wide range of voltage variations as described in claim 6, characterized in that: In step S2, the optimized modulation scheme for operating mode four is as follows: By using phase shifting, the rising edge of the PWM drive signal of the primary side upper switch S1 is aligned with the rising edge of the PWM drive signal of the first secondary side upper switch Q1, and the rising edge of the PWM drive signal of the primary side lower switch S2 is aligned with the rising edge of the PWM drive signal of the first secondary side lower switch Q2; the falling edge of the PWM drive signal of the second secondary side lower switch Q4 lags behind the falling edge of the PWM drive signal of the primary side upper switch S1, and the falling edge of the PWM drive signal of the second secondary side upper switch Q3 lags behind the falling edge of the PWM drive signal of the primary side lower switch S2. After the first secondary side lower switch Q2 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side upper switch Q1 is turned on in advance, achieving zero-voltage turn-on. After the first secondary side upper switch Q1 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the first secondary side lower switch Q2 is turned on in advance, achieving zero-voltage turn-on; After the second auxiliary upper switch Q3 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary lower switch Q4 is turned on in advance, achieving zero-voltage turn-on. After the second auxiliary side lower switch Q4 is turned off, the resonant current flows through the magnetizing inductor, and the anti-parallel diode of the second auxiliary side upper switch Q3 is turned on in advance, achieving zero-voltage turn-on.

Images