High dynamic performance power control method applied to double active full-bridge three-phase bidirectional ac / dc converter

Abstract

The application discloses a high dynamic characteristic power control method applied to a double active full-bridge three-phase bidirectional AC / DC converter, which is characterized in that: a power error signal is obtained by subtracting a target power from a grid-side active power feedback quantity; an output value of the error signal after passing through a compensation network is calculated by a dichotomy method to obtain a phase-shifting angle and a duty cycle data required by the circuit at different time points, so as to realize feedback control; theoretical data of the phase-shifting angle and the duty cycle are calculated by the dichotomy method based on a theoretical power and a circuit hardware parameter value, and an expression between the phase-shifting angle and the duty cycle and a circuit phase and power is fitted to serve as a fitting calculation formula of feedforward control; the feedforward control is added on the basis of the feedback control, the phase-shifting angle and the duty cycle obtained by the feedforward control and the feedback control are added respectively, and a sum of the addition is used as the phase-shifting angle and the duty cycle finally acting on the main circuit. The control method is simple and can realize high dynamic characteristic precise control of power of the double active full-bridge three-phase bidirectional AC / DC converter.

Description

Technical Field

[0001] This invention relates to three-phase bidirectional AC / DC converter technology, and more specifically to a high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter. Background Technology

[0002] The concept of V2G (Vehicle to Grid) was proposed to address the aforementioned problems. First proposed by Amory Lovins in 1995, V2G technology refers to the bidirectional flow of energy between electric vehicle batteries and the power grid, using electric vehicle batteries as energy storage devices. Its core idea is to utilize the energy storage devices of numerous electric vehicles as an energy buffer between the power grid and renewable energy sources.

[0003] To enable V2G (Vehicle-to-Grid) connectivity for electric vehicles, key interface devices are needed between the power grid and the vehicle to allow for bidirectional energy flow. The connection between the electric vehicle and the power grid requires a bidirectional AC / DC converter based on power electronic devices.

[0004] Most scholars have studied the circuit structure and control strategies of dual active full-bridge three-phase bidirectional AC / DC converters, but few have considered the change in the required charging power during the charging process of electric vehicles. The method often used is simple feedback control, which has a slow response speed. Therefore, it is of great significance to develop a high dynamic characteristic power control method for the converter. Summary of the Invention

[0005] The purpose of this invention is to propose a high dynamic characteristic power control method for dual active full-bridge three-phase bidirectional AC / DC converters.

[0006] The technical solution to achieve the purpose of this invention is: a dual active full-bridge three-phase bidirectional AC / DC converter, including an AC-side filter inductor L. a L b and L c AC side filter capacitor C a C b and C c AC side three-phase bridge arm, energy transfer inductor L, high-frequency transformer T r DC-side full-bridge circuit, DC-side filter capacitor C dc DC-side filter inductor L dc ,in:

[0007] AC side filter inductor L a L b and L cOne end is connected to the three-phase AC power supply; the three bridge arms on the AC side include the first bridge arm, the second bridge arm, and the third bridge arm, with the first bridge arm consisting of a switching transistor S. a1 S a2 S a3 and S a4 The bridge is composed of series connections, with the second arm consisting of a switching transistor S. b1 S b2 S b3 and S b4 The bridge is composed of series connections, with the third arm consisting of a switching transistor S. c1 S c2 S c3 and S c4 Series connection; in the first bridge arm, the switching transistor S a1 The emitter and the switch S a2 The emitter is connected, and the switch S a3 The emitter and the switch S a4 The emitter is connected, and the switch S a2 collector and switch S a3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. a The other end is connected to capacitor C. a One end is connected; in the second bridge arm, the switch S b1 The emitter and the switch S b2 The emitter is connected, and the switch S b3 The emitter and the switch S b4 The emitter is connected, and the switch S b2 collector and switch S b3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. b The other end is connected to capacitor C. b One end is connected; in the third bridge arm, the switch S c1 The emitter and the switch S c2 The emitter is connected, and the switch S c3 The emitter and the switch S c4 The emitter is connected, and the switch S c2 collector and switch S c3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. c The other end is connected to capacitor C. c One end is connected; AC side filter capacitor C a C b and C c The other end is connected together; the switching transistor S a1 collector and switch S b1 S c1 The collectors of the transistors are connected, and the connection point serves as the first common port; the switching transistor S... a4 collector and switch Sb4 S c4 The collectors are connected, and the connection point serves as the second common port;

[0008] The DC-side full-bridge circuit consists of a fourth and a fifth bridge arm connected in parallel. The fourth bridge arm is formed by the series connection of the emitter of switch S1 and the collector of switch S2, with the connection point serving as the third common port. The fifth bridge arm is formed by the series connection of the emitter of switch S3 and the collector of switch S4, with the connection point serving as the fourth common port. The collectors of switch S1 and S3 are connected, and the connection point is then connected to the DC-side filter capacitor C. dc The positive terminal of transistor S2 is connected to the positive terminal of transistor S4; the emitter of transistor S2 is connected to the emitter of transistor S4, and the connection point is then connected to the DC-side filter capacitor C. dc The negative terminal is connected; DC side filter capacitor C dc The positive terminal and the DC side filter inductor L dc One end is connected, and the negative end is connected to the DC power supply; DC side filter inductor L dc The other end is connected to the DC power supply;

[0009] One end of the energy transfer inductor L is connected to the first common port, and the high-frequency transformer T... r The primary windings are connected to the other end of the energy transfer inductor L and the second common port, respectively; the secondary windings are connected to the third common port and the fourth common port, respectively; high-frequency transformer T r The primary winding is connected to the other end of the inductor L, and the secondary winding is connected to one end of the third common port, which are the same name terminals.

[0010] Furthermore, based on the direction of energy flow in the converter, power flow from the AC side to the DC side is defined as rectification, and power flow from the DC side to the AC side is defined as inversion.

[0011] A high dynamic characteristic power control method applied to the aforementioned dual active full-bridge three-phase bidirectional AC / DC converter includes the following steps:

[0012] Step 1, switch transistor S c1 collector and switching transistor S c4 The voltage difference between the collectors is defined as u p The terminal voltage of the primary winding of the high-frequency transformer is defined as u. s Define the phase voltage of phase A as e a =V m cos(φ), the phase voltage of phase B is e b =V m cos(φ+2π / 3), the phase voltage of phase C is e c =V m cos(φ-2π / 3), V mLet f be the phase voltage amplitude; define the power supply frequency as f, and one power frequency cycle as T; define the line voltage with the largest amplitude among the three-phase line voltages as u. max The second largest line voltage is located as u. med Define the second largest line voltage u within a switching cycle. med Duration and switching period T s The ratio is d m Define u s and u p The ratio of the phase difference to π / 2 is the phase shift percentage δ; the phase of one power frequency cycle is 2π, and the zero-crossing point of the A-phase voltage from negative to positive is defined as the starting point of phase φ, and the duration of each sector's phase is π / 6; based on the two-wire voltage modulation, the output power P and δ of the DC-side power supply and d are determined. m Relational expressions;

[0013] Step 2, input the target power P* and the grid-side active power feedback quantity P. f The difference is calculated to obtain the power error signal. After passing through the compensation network, the power error signal yields the adjustment value P*'. This adjustment value is then substituted into the binary search method for calculation to obtain δ1 and d in each switching cycle. m1 This allows the active power on the grid side to approach the target power P*, enabling feedback control. The compensation network uses PI or PID control methods.

[0014] Step 3, based on P and δ and d m Substituting the relational expression with different target powers P*, we can calculate d at different target powers. m The theoretical data of δ are fitted to d m The expressions for δ and phase φ and target power P*, which are the dependent variables, and are used as the fitting calculation formula for feedforward control;

[0015] Step 4: Substitute sector N, target power P*, and phase φ into the fitting formula of feedforward control to obtain δ2 and d. m2 ; δ1, d m1 and δ2 and d respectively m2 Summing, we get δ3 and d. m3 To control the duration of the maximum and second-largest line voltages during the switching cycle and u s u p The phase difference is used to complete the high dynamic characteristic power control by adding feedforward control to feedback control.

[0016] Further, in step 1, based on the two-wire voltage modulation, determine the output power P and δ of the DC-side power supply, as well as d. m The relational expression, specifically, is as follows:

[0017] 1) Divide one power frequency cycle T into 12 sectors on average.

[0018] Define sector 1 as e a >e b >e c And e b <0; Define sector 2 as e a >e b >e c And e b >0; Define sector 3 as e b >e a >e c And e a >0; Define sector 4 as e b >e a >e c And e a <0; Define sector 5 as e b >e c >e a And e c <0; Define sector 6 as e b >e c >e a And e c >0; Define sector 7 as e c >e b >e a And e b >0; Define sector 8 as e c >e b >e a And e b <0; Define sector 9 as e c >e a >e b And e a <0; Define sector 10 as e c >e a >e b And e a >0; Define sector 11 as e a >e c >e b And e c >0; Define sector 12 as e a >e c >e b And e c <0;

[0019] 2) Determine the maximum and second-largest line voltages in each sector.

[0020] The maximum and second-largest line voltages in each sector are shown in Table 1.

[0021] Table 1. Distribution of maximum and second-maximum line voltages in 112 sectors.

[0022]

[0023]

[0024] 3) Solve for P, δ, and d. m relational expressions

[0025] According to the two-wire voltage modulation strategy, P and d in each sector m The relationship between δ and δ is consistent, and can be expressed as:

[0026]

[0027] Among them, V dc It is the DC-side output voltage, f s It is the switching frequency, and the reference target average value of the B-phase current is equal to the average current of the B-phase, that is... The target power is equal to the average power on the DC side, i.e., P * =P, where n is the ratio of the number of turns in the primary winding to the number of turns in the secondary winding of the high-frequency transformer.

[0028] Further, in step 2, the target power P* and the grid-side active power feedback quantity P are... f The difference is calculated to obtain the power error signal. After passing through the compensation network, the power error signal yields the adjustment value P*'. This adjustment value is then substituted into the binary search method for calculation to obtain δ1 and d in each switching cycle. m1 This allows the active power on the grid side to approach the target power P*, achieving feedback control. The specific process of the bisection method is as follows:

[0029] S1: Initialize the loop count i = 0, the maximum loop count k, and the minimum value of δ. min =0, δ maximum value δ max =0.5;

[0030] S2: Calculate δ=(δ min +δ max ) / 2;

[0031] S3: Check if the loop count i is greater than k. If it is, output d directly. m (δ), otherwise proceed to S4;

[0032] S4: Determine if P(δ) is greater than P*'. If P(δ) is greater than P*', then update δ. max =δ, otherwise update δ min =δ;

[0033] S6: i = i + 1;

[0034] S7: Jump to S2 and proceed to the next iteration until the iteration count k is met, at which point δ and d are obtained. m The value of (δ).

[0035] Furthermore, the grid-side current i in the three-phase stationary coordinate system is... a i b and i c Perform a coordinate transformation to obtain the d-axis component I in the dq synchronous rotating coordinate system. d , representing the active component of the grid-side current; the grid-side voltage e in the three-phase stationary coordinate system a e b and e c Perform a coordinate transformation to obtain the d-axis component V in the dq synchronous rotating coordinate system. d , representing the active component of the grid-side voltage; multiplying the voltage and current components on the d-axis yields P. f .

[0036] Furthermore, in step 3, the circuit operating phase in sector 1 is... in Based on P, δ, and d m Substituting the relational expression into different target powers P*, we can calculate the d in sector 1 at different target powers. m The theoretical data of δ are fitted to d m δ is the dependent variable, and phase is the dependent variable. The expression for the target power P* as the independent variable is used as the fitting calculation formula for the feedforward control. The specific method is as follows:

[0037] Due to the use of two-wire voltage modulation, δ and d in each sector m The data are correlated, so we only need to solve the fitting calculation formula for the feedforward control of sector 1. The δ and d values ​​for the other sectors can then be calculated using the fitting formula for sector 1. m The specific process is as follows:

[0038] S1. The current sector is N. This represents the current operating phase of the circuit.

[0039] S2. Determine if sector N is odd;

[0040] S3. If the sector is an odd number of sectors, then the phase... The formula for converting to sector 1 phase is:

[0041]

[0042] S4. If the sector is an even number of sectors, then the phase... First convert to sector 2, then convert to sector 1. The formula is:

[0043] Converting φ to sector 2, the formula is:

[0044]

[0045] Will Converting to sector 1, the formula is:

[0046]

[0047] A high dynamic characteristic power control system for a dual active full-bridge three-phase bidirectional AC / DC converter is provided. The high dynamic characteristic power control method for the dual active full-bridge three-phase bidirectional AC / DC converter is implemented to achieve high dynamic characteristic power control of the dual active full-bridge three-phase bidirectional AC / DC converter.

[0048] A computer device includes a memory, a processor, and a computer program stored in the memory and executable on the processor. When the processor executes the computer program, it implements the high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter, thereby realizing high dynamic characteristic power control of the dual active full-bridge three-phase bidirectional AC / DC converter.

[0049] A computer-readable storage medium storing a computer program, which, when executed by a processor, implements the high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter, thereby realizing high dynamic characteristic power control of the dual active full-bridge three-phase bidirectional AC / DC converter.

[0050] Compared with the prior art, the significant advantages of this invention are that it can reduce the overshoot caused by changes in target power, and improve the dynamic response characteristics and stability of the circuit. Attached Figure Description

[0051] Figure 1 This is the topology diagram of a dual active full-bridge three-phase bidirectional AC / DC converter.

[0052] Figure 2 This is a high dynamic characteristic power control block diagram applied to a dual active full-bridge three-phase bidirectional AC / DC converter.

[0053] Figure 3 This is a schematic diagram of two-wire voltage modulation.

[0054] Figure 4 The calculation of sector i in sector 1 using the binary search method a i b and i c A schematic diagram.

[0055] Figure 5 It is a two-line voltage modulation rectification mode u p u s and i L The phase-shift control waveform diagram.

[0056] Figure 6 It is a dual-line voltage modulation inverter mode u p u s and i L The phase-shift control waveform diagram.

[0057] Figure 7 Waveform diagram of the rectifier mode switching transistor drive.

[0058] Figure 8 Waveform diagram of the switching transistor drive in inverter mode.

[0059] Figure 9 This is a flowchart of the bisection method calculation.

[0060] Figure 10 This is the δ-fit surface plot under different target powers.

[0061] Figure 11 d under different target power m Fit the surface plot.

[0062] Figure 12 Simulation waveforms of single feedback control from 1500W to 500W.

[0063] Figure 13 Simulation waveforms of high dynamic characteristic power control from 1500W to 500W.

[0064] Figure 14 Simulation waveforms of single feedback control from 500W to 1500W.

[0065] Figure 15 Simulation waveforms of high dynamic characteristic power control from 500W to 1500W. Detailed Implementation

[0066] To make the objectives, technical solutions, and advantages of this application clearer, the following detailed description is provided in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and not intended to limit the scope of this application.

[0067] The dual active full-bridge three-phase bidirectional AC / DC converter includes an AC-side filter inductor L. a L b and L c AC side filter capacitor C a C b and C c AC side three-phase bridge arm, energy transfer inductor L, high-frequency transformer T r DC-side full-bridge circuit, DC-side filter capacitor C dc DC-side filter inductor L dc ,in:

[0068] AC side filter inductor L a L b and L c One end is connected to the three-phase AC power supply; the three bridge arms on the AC side include the first bridge arm, the second bridge arm, and the third bridge arm, with the first bridge arm consisting of a switching transistor S. a1 S a2 S a3 and S a4 The bridge is composed of series connections, with the second arm consisting of a switching transistor S. b1 S b2 S b3 and S b4 The bridge is composed of series connections, with the third arm consisting of a switching transistor S. c1 S c2 S c3 and S c4 Series connection; in the first bridge arm, the switching transistor S a1 The emitter and the switch S a2 The emitter is connected, and the switch S a3 The emitter and the switch S a4 The emitter is connected, and the switch S a2 collector and switch S a3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. a The other end is connected to capacitor C. a One end is connected; in the second bridge arm, the switch S b1 The emitter and the switch S b2 The emitter is connected, and the switch S b3 The emitter and the switch S b4 The emitter is connected, and the switch S b2 collector and switch S b3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. b The other end is connected to capacitor C. b One end is connected; in the third bridge arm, the switch S c1 The emitter and the switch S c2 The emitter is connected, and the switch S c3 The emitter and the switch S c4 The emitter is connected, and the switch S c2 collector and switch S c3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. c The other end is connected to capacitor C. c One end is connected; AC side filter capacitor C a C b and C c The other end is connected together; the switching transistor S a1 collector and switch S b1S c1 The collectors of the transistors are connected, and the connection point serves as the first common port; the switching transistor S... a4 collector and switch S b4 S c4 The collectors are connected, and the connection point serves as the second common port;

[0069] The DC-side full-bridge circuit consists of a fourth and a fifth bridge arm connected in parallel. The fourth bridge arm is formed by the series connection of the emitter of switch S1 and the collector of switch S2, with the connection point serving as the third common port. The fifth bridge arm is formed by the series connection of the emitter of switch S3 and the collector of switch S4, with the connection point serving as the fourth common port. The collectors of switch S1 and S3 are connected, and the connection point is then connected to the DC-side filter capacitor C. dc The positive terminal of transistor S2 is connected to the positive terminal of transistor S4; the emitter of transistor S2 is connected to the emitter of transistor S4, and the connection point is then connected to the DC-side filter capacitor C. dc The negative terminal is connected; DC side filter capacitor C dc The positive terminal and the DC side filter inductor L dc One end is connected, and the negative end is connected to the DC power supply; DC side filter inductor L dc The other end is connected to the DC power supply;

[0070] One end of the energy transfer inductor L is connected to the first common port, and the high-frequency transformer T... r The primary windings are connected to the other end of the energy transfer inductor L and the second common port, respectively; the secondary windings are connected to the third common port and the fourth common port, respectively; high-frequency transformer T r The primary winding connected to the other end of inductor L and the secondary winding connected to one end of the third common port are called the same-name terminals. Based on the direction of energy flow in the converter, power flow from the AC side to the DC side is defined as rectification, and power flow from the DC side to the AC side is defined as inversion. Let S... c1 collector and S c4 The voltage difference between the collectors is defined as u p The terminal voltage of the primary winding of the high-frequency transformer is defined as u. s .

[0071] This circuit employs a two-wire voltage modulation strategy on the AC side bridge arm, which divides one power frequency cycle T into 12 sectors. The switching cycle T within each sector... s In this case, the line voltage u with the largest amplitude is selected. max and the second largest line voltage u med Define the second largest line voltage u during a switching cycle. med Duration and switching period T s The ratio is d m Define u s and u pThe ratio of the phase difference to π / 2 is the phase shift percentage δ. During the switching cycle, δ and d... m The values ​​of the two degrees of freedom determine the magnitude of the DC-side power supply output power.

[0072] To achieve high dynamic power control in this circuit, power feedforward and feedback control need to be combined. The specific method is as follows:

[0073] Step 1: Based on the two-wire voltage modulation, derive the output power P and δ of the DC-side power supply, as well as d. m After obtaining the relational expression, δ and d can be solved based on the DC-side power supply output power P. m The values ​​for the two degrees of freedom. The specific details of the two-wire voltage modulation are as follows:

[0074] (1) Divide the sector

[0075] First, a power frequency cycle T is divided into 12 sectors on average, and then the line voltages with the largest and second largest amplitudes are selected in each sector.

[0076] Define the phase voltage of phase A as The phase voltage of phase B is The phase voltage of phase C is V m This represents the phase voltage amplitude. Sector 1 is defined as e. a >e b >e c And e b <0; Define sector 2 as e a >e b >e c And e b >0; Define sector 3 as e b >e a >e c And e a >0; Define sector 4 as e b >e a >e c And e a <0; Define sector 5 as e b >e c >e a And e c <0; Define sector 6 as e b >e c >e a And e c >0; Define sector 7 as e c >e b >e a And e b >0; Define sector 8 as e c >e b >e a And eb <0; Define sector 9 as e c >e a >e b And e a <0; Define sector 10 as e c >e a >e b And e a >0; Define sector 11 as e a >e c >e b And e c >0; Define sector 12 as e a >e c >e b And e c <0. The division is shown in Table 1.

[0077] Table 112 Distribution of Maximum and Second Maximum Line Voltages in Sectors

[0078] sector Maximum line voltage Second largest line voltage sector Maximum line voltage Second largest line voltage Ⅰ <![CDATA[u ac ]]> <![CDATA[u ab ]]> Ⅶ <![CDATA[u ca ]]> <![CDATA[u ba ]]> Ⅱ <![CDATA[u ac ]]> <![CDATA[u bc ]]> Ⅷ <![CDATA[u ca ]]> <![CDATA[u cb ]]> Ⅲ <![CDATA[u bc ]]> <![CDATA[u ac ]]> Ⅸ <![CDATA[u cb ]]> <![CDATA[u ca ]]> Ⅳ <![CDATA[u bc ]]> <![CDATA[u ba ]]> Ⅹ <![CDATA[u cb ]]> <![CDATA[u ab ]]> Ⅴ <![CDATA[u ba ]]> <![CDATA[u bc ]]> ⅩⅠ <![CDATA[u ab ]]> <![CDATA[u cb ]]> Ⅵ <![CDATA[u ba ]]> <![CDATA[u ca ]]> ⅩⅡ <![CDATA[u ab ]]> <![CDATA[u ac ]]>

[0079] (2) Determine the maximum and second-largest line voltages acting on u during the switching cycle. p The sequence and the DC side voltage acting on u s The resulting waveform

[0080] To achieve volt-second balance, the voltage u across the transformer must be within each switching cycle. p u s It is symmetrical between positive and negative, V dc It is the DC-side output voltage, based on the voltage across inductor L and the current i flowing through L. L One switching cycle can be divided into 6 stages t1 to t6.

[0081] During one switching cycle, the DC side voltage acts on u s The order is as follows: voltage V dc and -V dc The duration of action is T. s / 2, δ is u s and u p The phase shift ratio ranges from [0, 0.5]. In rectification mode, the DC-side output voltage lags behind the AC-side output voltage, so the DC side continuously outputs δT first. s Time -V dc Then output T s / 2 time V dc Finally, output T. s / 2-δT s Time -V dcIn inverter mode, the DC-side output voltage leads the AC-side output voltage, so the DC side continuously outputs T first. s / 2-δT s V of time dc Then output T s / 2 time -V dc Finally, output δT s V of time dc ;

[0082] During one switching cycle, the voltage u max The duration of action is T s / 2-d m T s Voltage u med The duration of action is d m T s The maximum and second-largest line voltages act on u. p The sequence is as follows: in rectification mode, the output voltage u is output first. max Then output u med Due to the symmetry between positive and negative signs, -u is then output sequentially. max and -u med In inverter mode, the output voltage u is first... med Then output u max Due to the symmetry between positive and negative signs, -u is then output sequentially. med and -u max ;

[0083] The switching time distribution in each sector under rectification and inversion modes is shown in Table 2:

[0084] Table 2 Duration of each time period in rectification and inversion modes

[0085]

[0086] u in each sector under rectification and inversion modes p and u s The distribution during the switching cycle is shown in Table 3:

[0087] Table 3 u under rectification and inversion modes p and u s Values ​​within each time period

[0088]

[0089] Where n is the turns ratio of the high-frequency transformer, referring to the ratio of the primary and secondary windings of the high-frequency transformer, V dc This is the DC-side power supply voltage.

[0090] (3) Determine the switch status

[0091] Since the maximum and second-largest line voltages are selected differently in different sectors, the above-mentioned dual-line voltage modulation switching states are implemented in different sectors as follows.

[0092] Define the switching state of the switching transistor: S a1 and S a2 It is simultaneously turned on and off; S a3 and S a4 It is simultaneously turned on and off; S b1 and S b2 It is simultaneously turned on and off; S b3 and S b4 It is simultaneously turned on and off; S c1 and S c2 It is simultaneously turned on and off; S c3 and S c4 They are turned on and off simultaneously. In both rectification and inversion modes, in each of the 12 sectors, during each switching cycle T... s The internal switch status is shown in the table below:

[0093] Table 4 shows the switching cycle T of sector 1 in rectification mode. s Internal switch status

[0094]

[0095] Table 5 shows the switching cycle T of sector 1 in inverter mode. s Internal switch status

[0096]

[0097] Table 6. Switching cycle T of sector 2 in rectification mode. s Internal switch status

[0098]

[0099]

[0100] Table 7. Switching cycle T in sector 2 under inverter mode. s Internal switch status

[0101]

[0102] Table 8. Switching cycle T of sector 3 in rectification mode. s Internal switch status

[0103]

[0104] Table 9 shows the switching cycle T of sector 3 in inverter mode. s Internal switch status

[0105]

[0106]

[0107] Table 10 shows the switching cycle T of sector 4 in rectification mode. s Internal switch status

[0108]

[0109] Table 11 shows the switching cycle T of sector 4 in inverter mode. s Internal switch status

[0110]

[0111] Table 12 shows the switching cycle T of sector 5 in rectification mode. s Internal switch status

[0112]

[0113]

[0114] Table 13 shows the switching cycle T of sector 5 in inverter mode. s Internal switch status

[0115]

[0116] Table 14 shows the switching cycle T of sector 6 in rectification mode. s Internal switch status

[0117]

[0118] Table 15 shows the switching cycle T of sector 6 in inverter mode. s Internal switch status

[0119]

[0120] Table 16 shows the switching cycle T of sector 7 in rectification mode. s Internal switch status

[0121]

[0122] Table 17 shows the switching cycle T of sector 7 in inverter mode. s Internal switch status

[0123]

[0124] Table 18 shows the switching cycle T of sector 8 in rectification mode. s Internal switch status

[0125]

[0126] Table 19 shows the switching cycle T of sector 8 in inverter mode. s Internal switch status

[0127]

[0128]

[0129] Table 20 shows the switching cycle T of sector 9 in rectification mode. s Internal switch status

[0130]

[0131] Table 21 shows the switching cycle T of sector 9 in inverter mode. s Internal switch status

[0132]

[0133] Table 22 shows the switching cycle T of sector 10 in rectification mode. s Internal switch status

[0134]

[0135]

[0136] Table 23 shows the switching cycle T of sector 10 in inverter mode. s Internal switch status

[0137]

[0138] Table 24 shows the switching cycle T of sector 11 in rectification mode. s Internal switch status

[0139]

[0140] Table 25 shows the switching cycle T of sector 11 in inverter mode. s Internal switch status

[0141]

[0142]

[0143] Table 26 shows the switching cycle T of sector 12 in rectification mode. s Internal switch status

[0144]

[0145] Table 27 shows the switching cycle T of sector 12 in inverter mode.s Internal switch status

[0146]

[0147] Based on two-wire voltage modulation, the grid-side active power P and d are calculated under both rectification and inversion conditions. m The method for solving the relationship between δ and δ is the same, and its phase-shifting control waveform is shown in the figure below. Figure 5 , 6 As shown. According to Figure 5 The phase-shift control waveform diagram in rectification mode, defining the current flowing through inductor L at time t0 as i. L0 This allows us to obtain the current i passing through inductor L at various time intervals. L expression:

[0148]

[0149] Based on the expression for inductor current, the expression for grid-side active power P is derived:

[0150]

[0151] according to Figure 7 The switching state of sector 1 determines the three-phase current i during the switching cycle. a i b and i c The waveform, such as Figure 4 As shown. When the voltage u p Maximum line voltage u ac At that time, the three-phase current i a i b and i c The values ​​are i L , 0 and -i L When voltage u p The second largest line voltage u ab At that time, the three-phase current i a i b and i c The values ​​are i L , 0 and -i L .

[0152] Based on the expressions for the inductor current in each segment, the expression for the average current of phase B in sector 1 is derived:

[0153]

[0154] The control degrees of freedom δ and d can be solved using equations 2 and 3. m The solution method is as follows: First, in the system of equations, d mThe equation is transformed into an expression in terms of δ, resulting in the single-variable equation P = P(δ). Next, δ in P = P(δ) is solved. Finally, d is obtained using δ. m The following are the specific steps:

[0155] The nonlinear simultaneous equations are transformed into single-variable equations, where the reference target mean value of the phase B current is equal to the average phase B current, i.e. The target power equals the average power on the DC side, i.e., P* = P. Dividing the two equations, we get:

[0156]

[0157] By solving equation 4, the duty cycle d m It can be represented as:

[0158]

[0159] In equation 5, d m Substituting (δ) into Equation 3, we obtain the power expression for the single variable δ:

[0160]

[0161] Therefore, P and d can be derived through two-line voltage modulation. m The relationship between δ and P(δ) is consistent with Formula 6 for the remaining sectors. Based on d... m The DC output power of the circuit can be calculated using δ, and then the d-value that satisfies the target power P* can be calculated. m And δ.

[0162] Step 2: Implement feedback control of grid-side active power. This involves controlling the target power P* and the grid-side active power feedback quantity P. f The difference is calculated to obtain the power error signal. After the power error signal is passed through PI compensation control, the adjustment amount P*' is obtained. Substituting this into the binary search method, the d of the circuit in each switching cycle is obtained. m And δ, so that the active power on the grid side is close to the target power P* to achieve feedback control;

[0163] In power calculation using the bisection method, it is necessary to set boundary values ​​for the power. This takes into account a specific situation, namely when the voltage u... p Only when its maximum value u is reached max And d m When (δ) equals 0, the power expression simplifies to a quadratic function of δ. Due to modulation requirements, the range of δ is limited to [0, 0.5]. When δ is 0, no power transmission occurs; when δ reaches its upper limit of 0.5, the maximum transmitted power P is achieved. max =u max nV dc / 8f s L. The process of the bisection method is as follows:

[0164] S1: Initialize the loop count i = 0, the maximum loop count k, and the minimum value of δ. min =0, δ maximum value δ max =0.5;

[0165] S2: Calculate δ=(δ min +δ max ) / 2;

[0166] S3: Check if the loop count i is greater than k. If it is, output d directly. m (δ), otherwise proceed to S4;

[0167] S4: Determine if P(δ) is greater than P*'. If P(δ) is greater than P*', then update δ. max =δ, otherwise update δ min =δ;

[0168] S6: i = i + 1;

[0169] S7: Jump to S2 and proceed to the next iteration until the iteration count k is met, at which point δ and d are obtained. m The value of (δ).

[0170] Repeat the above calculation process until the number of iterations k is satisfied, and then stop the calculation to obtain d. m The value of (δ).

[0171] This algorithm can obtain the δ and d values ​​of the circuit at different times when a given target power is achieved. m .

[0172] Grid-side active power feedback P f The method for determining this is as follows: The grid-side current i in the three-phase stationary coordinate system... a i b and i c Perform a coordinate transformation to obtain the d-axis component I in the dq synchronous rotating coordinate system. d I d This represents the active component of the grid-side current. Then, the grid-side voltage e in the three-phase stationary coordinate system is... a e b and e c Perform a coordinate transformation to obtain the d-axis component V in the dq synchronous rotating coordinate system. d V d This represents the active component of the grid-side voltage. Multiplying the voltage and current components on the d-axis and then multiplying by the appropriate coefficient yields the feedback grid-side active power P. f The active power P on the feedback network side will be... fThe difference between the target power P* and the control value is used to obtain the adjustment value P*' after compensation control. The adjustment value P*' is then substituted into the binary division method for calculation to obtain the δ1 and d required by the circuit for different target powers. m1 Data. Ultimately, δ1 and d m1 The data is modulated by PWM to obtain the drive signal for the switching transistor, thereby achieving the purpose of controlling the grid-side power P.

[0173] Step 3: Solve the fitting calculation formula for feedforward control. The circuit operating phase in sector 1 is... in Based on P, δ, and d m Substituting the relational expression into different target powers P*, we can calculate the d in sector 1 at different target powers. m The theoretical data of δ are fitted to d m δ is the dependent variable, and phase is the dependent variable. The expression with the target power P* as the independent variable is used as the fitting calculation formula for feedforward control to improve the circuit response speed.

[0174] Due to the use of two-wire voltage modulation, δ and d in each sector m The data are correlated, so only a fitting formula for one sector is needed. The phase of one power frequency cycle is 2π, and the beginning of one power frequency cycle is the phase... At the beginning, the duration of each sector is π / 6, and the δ and d of each sector in each power frequency cycle under the same target power are... m The data is the same. Circuit phase. The circuit's operating time t can be converted into phase, overcoming the limitations of time-based fitting and facilitating the determination of the sector N based on phase. Furthermore, phase fitting can be extended to period calculations, allowing the use of a sector-specific fitting formula to solve for δ and d in different sectors with different power frequency periods through phase-period operations. m data.

[0175] Define sector N, Due to the use of two-wire voltage modulation, δ and d in each sector m The data are correlated, so we only need to solve the fitting calculation formula for the feedforward control of sector 1. The δ and d values ​​for the other sectors can then be calculated using the fitting formula for sector 1. m The specific process is as follows:

[0176] S1. The current sector is N. This represents the current operating phase of the circuit.

[0177] S2. Determine if sector N is odd;

[0178] S3. If the sector is an odd number of sectors, then the phase... The formula for converting to sector 1 phase is:

[0179]

[0180] S4. If the sector is an even number of sectors, then the phase... First convert to sector 2, then convert to sector 1. The formula is:

[0181] Will Converting to sector 2, the formula is:

[0182]

[0183] Will Converting to sector 1, the formula is:

[0184]

[0185] During the fitting process, the target power is first set to P. i *(i=1, 2, 3, 4, 5...n). Using formula 6 derived from two-wire voltage modulation, the values ​​of δ and d within sector 1 are calculated for different target powers. m The data is then fitted to obtain a surface, with δ and d respectively. m The dependent variable is the phase corresponding to this sector. and target power P i Formulas where * represents the independent variable:

[0186]

[0187] Therefore, under different target powers, the δ and d calculated by the bisection method can be used to calculate the... m The data is used to obtain a fitting formula for sector 1, which can be based on the phase corresponding to the sector. and the target power P in circuit theory i * Calculate δ2 and d within one switching cycle. m2 The calculation of the data enables feedforward control in the high dynamic characteristic power control method.

[0188] Step 4: Combine feedback control with feedforward control, considering sector N, target power P*, and the converted phase. Substituting into the fitting formula for feedforward control, we obtain δ2 and d. m2 Then, the δ1 and d calculated by feedback control are... m1 and δ2, d m2 Summing, we get δ3 and d. m3 δ3 and d are obtained within one switching cycle. m3 Then, the duration of the maximum and second-largest line voltages within the switching cycle and u can be controlled.s u p The phase difference enables high dynamic power control by incorporating feedforward control into feedback control.

[0189] Example

[0190] The circuit structure is as follows: Figure 1 As shown in the table below, its specific parameters are as follows.

[0191] Table 1 Circuit Parameters

[0192]

[0193]

[0194] like Figure 2 The diagram shows a high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter. The implementation steps are as follows:

[0195] Step 1: Determine the output power P and δ of the DC-side power supply, as well as d, based on the two-wire voltage modulation. m The relational expression, specifically, is as follows:

[0196] (1) Basic Definition

[0197] Figure 1 For the circuit structure used, based on the direction of energy flow in the converter, power flow from the AC side to the DC side is defined as rectification, and power flow from the DC side to the AC side is defined as inversion. Let S... c1 The collector and the Sth c4 The voltage difference between the collectors is defined as u p The terminal voltage of the primary winding of the high-frequency transformer is defined as u. s Define u s and u p The phase difference is δ.

[0198] (2) Divide into sectors

[0199] The AC power supply provides a three-phase sinusoidal voltage, such as Figure 3 The diagram shows that one power frequency cycle T is divided into 12 sectors. The line voltage with the largest amplitude among the three-phase line voltages is defined as u. max The second largest line voltage is defined as u. med Define the second largest line voltage u within a switching cycle. med The duration percentage is d m .

[0200] (3) Establish u p and u s Voltage distribution during the switching cycle

[0201] Each switching cycle T s Divided into 6 segments, in order to meet the volt-second balance, the voltage u across the transformer is required to... p u s It is symmetrical between positive and negative, such as Figure 5 , 6 As shown, phase shift control is performed in both rectification and inverter modes.

[0202] (4) Determine the switch status

[0203] In both rectification and inversion modes, each switching cycle T in sector 1 s Internal switch status as follows Figure 7 , 8 As shown:

[0204] Similarly, the on / off states of the remaining 11 sectors can be obtained under both modes.

[0205] (5) Solve for P, δ, and d. m relational expressions

[0206] According to the two-wire voltage modulation strategy, P and d in each sector m The relationship between δ and δ is consistent, and can be expressed as:

[0207]

[0208]

[0209] Step 2: Implement feedback control of the converter, and control the grid-side current i in the three-phase stationary coordinate system. a i b and i c Perform a coordinate transformation to obtain the d-axis component I in the dq synchronous rotating coordinate system. d , representing the active component of the grid-side current; the grid-side voltage e in the three-phase stationary coordinate system a e b and e c Perform a coordinate transformation to obtain the d-axis component V in the dq synchronous rotating coordinate system. d , representing the active component of the grid-side voltage; multiplying the voltage and current components on the d-axis yields the feedback grid-side active power P. f Then, the target power P* and the grid-side active power feedback P are calculated. f The difference is calculated to obtain the power error signal. After the power error signal is passed through PI compensation control, the adjustment value P*' is obtained. Substituting this value into the binary search method, δ1 and d are obtained for each switching cycle. m1 This allows the active power on the grid side to approach the target power P*, achieving feedback control. The specific process of the bisection method is as follows:

[0210] S1: Initialize the loop count i = 0, the maximum loop count is 50, and the minimum value of δ is δ.min =0, δ maximum value δ max =0.5;

[0211] S2: Calculate δ=(δ min +δ max ) / 2;

[0212] S3: Check if the loop count i is greater than 50. If it is, output d directly. m (δ), otherwise proceed to S4;

[0213] S4: Determine if P(δ) is greater than P*'. If P(δ) is greater than P*', then update δ. max =δ, otherwise update δ min =δ;

[0214] S6: i = i + 1;

[0215] S7: Jump to S2 and proceed to the next iteration until the iteration count k is met, at which point δ and d are obtained. m The value of (δ).

[0216] Repeat the above calculation process until the number of iterations reaches 50, at which point the calculation stops, yielding d. m The value of (δ).

[0217] This algorithm can obtain the δ and d values ​​of the circuit at different times when a given target power is achieved. m .

[0218] Step 3, the circuit operating phase in sector 1 is in Based on P, δ, and d m Substituting the relational expression into different target powers P*, we can calculate the d in sector 1 at different target powers. m The theoretical data of δ are fitted to d m δ is the dependent variable, and phase is the dependent variable. The expression for the target power P* as the independent variable is used as the fitting calculation formula for the feedforward control. The specific method is as follows:

[0219] The circuit's power supply frequency is 50Hz, and its power frequency period is 0.02s. One power frequency period is divided into 12 sectors, and the phase of one power frequency period is 2π. The number of sectors is N. Due to the use of two-wire voltage modulation, the δ and d values ​​of each sector are... m Since the data are correlated, we only need to solve the fitting calculation formula for the feedforward control of sector 1. Through phase period calculation, we can obtain the fitting calculation formulas for the feedforward control of different sectors. The process is as follows:

[0220] During the fitting process, the target power is first set to P. i*(i=1, 2, 3, 4, 5, 6, 7) are set to 1500W, 1300W, 1100W, 1000W, 900W, 700W, and 500W respectively. The values ​​of δ and d within sector 1 are calculated using formula 6 derived from the two-wire voltage modulation, yielding the values ​​for different target power. m The data is then fitted to obtain a surface, such as... Figure 10 , 11 As shown, the x-axis represents the phase, the y-axis represents the target power, and the z-axis of the two graphs represents the phase shift angle and duty cycle, respectively.

[0221] Due to the use of two-wire voltage modulation, δ and d in each sector m The data are correlated, so we only need to solve the fitting calculation formula for the feedforward control of sector 1. The δ and d values ​​for the other sectors can then be calculated using the fitting formula for sector 1. m The specific process is as follows:

[0222] S1. The current sector is N. This represents the current operating phase of the circuit.

[0223] S2. Determine if sector N is odd;

[0224] S3. If the sector is an odd number of sectors, then the phase... The formula for converting to sector 1 phase is:

[0225]

[0226] S4. If the sector is an even number of sectors, then the phase... First convert to sector 2, then convert to sector 1. The formula is:

[0227] Will Converting to sector 2, the formula is:

[0228]

[0229] Converting φ2 to sector 1, the formula is:

[0230]

[0231] Using δ and d respectively m The dependent variable is the phase corresponding to this sector. and target power P i Formulas where * represents the independent variable:

[0232]

[0233] Therefore, under different target powers, the δ and d calculated by the bisection method can be used to calculate the... mThe data is used to obtain a fitting formula for sector 1, which can be based on the phase corresponding to the sector. and the target power P in circuit theory i * Calculate δ2 and d within one switching cycle. m2 The calculation of the data enables feedforward control in the high dynamic characteristic power control method.

[0234] Step 4: Combine feedback control with feedforward control. Substitute sector N, target power P*, and the converted phase φ1 into the fitting formula of feedforward control to obtain δ2 and d. m2 Then, the δ1 and d calculated by feedback control are... m1 and δ2, d m2 Summing, we get δ3 and d. m3 δ3 and d are obtained within one switching cycle. m3 Then, the duration of the maximum and second-largest line voltages within the switching cycle and u can be controlled. s u p The phase difference enables high dynamic power control by incorporating feedforward control into feedback control.

[0235] Simulation verification shows that, as can be seen... Figure 12 , 14 The diagram shows the power switching waveform obtained from single feedback control. It can be seen that there is a significant delay in reaching steady state at the switching node compared to the reference power. In contrast, the control waveform of high dynamic characteristic power control is as follows: Figure 13 , 15 As shown, it can quickly reach the target power with almost no delay at the switching point. This demonstrates that the high dynamic characteristic power control method can improve the dynamic response characteristics and stability of the circuit.

Claims

1. A high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter, characterized in that, A dual active full-bridge three-phase bidirectional AC / DC converter, characterized in that it includes an AC-side filter inductor L. a L b and L c AC side filter capacitor C a C b and C c AC side three-phase bridge arm, energy transfer inductor L, high-frequency transformer T r DC-side full-bridge circuit, DC-side filter capacitor C dc DC-side filter inductor L dc ,in: AC side filter inductor L a L b and L c One end is connected to the three-phase AC power supply; the three bridge arms on the AC side include the first bridge arm, the second bridge arm, and the third bridge arm, with the first bridge arm consisting of a switching transistor S. a1 S a2 S a3 and S a4 The bridge is composed of series connections, with the second arm consisting of a switching transistor S. b1 S b2 S b3 and S b4 The bridge is composed of series connections, with the third arm consisting of a switching transistor S. c1 S c2 S c3 and S c4 Series connection; in the first bridge arm, the switching transistor S a1 The emitter and the switch S a2 The emitter is connected, and the switch S a3 The emitter and the switch S a4 The emitter is connected, and the switch S a2 collector and switch S a3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. a The other end is connected to capacitor C. a One end is connected; in the second bridge arm, the switch S b1 The emitter and the switch S b2 The emitter is connected, and the switch S b3 The emitter and the switch S b4 The emitter is connected, and the switch S b2 collector and switch S b3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. b The other end is connected to capacitor C. b One end is connected; in the third bridge arm, the switch S c1 The emitter and the switch S c2 The emitter is connected, and the switch S c3 The emitter and the switch S c4 The emitter is connected, and the switch S c2 collector and switch S c3 The collector is connected to the inductor L, and the connection point is connected to the inductor L. c The other end is connected to capacitor C. c One end is connected; AC side filter capacitor C a C b and C c The other end is connected together; the switching transistor S a1 collector and switch S b1 S c1 The collectors of the transistors are connected, and the connection point serves as the first common port; the switching transistor S... a4 collector and switch S b4 S c4 The collectors are connected, and the connection point serves as the second common port; The DC-side full-bridge circuit consists of a fourth and a fifth bridge arm connected in parallel. The fourth bridge arm is formed by the series connection of the emitter of switch S1 and the collector of switch S2, with the connection point serving as the third common port. The fifth bridge arm is formed by the series connection of the emitter of switch S3 and the collector of switch S4, with the connection point serving as the fourth common port. The collectors of switch S1 and S3 are connected, and the connection point is then connected to the DC-side filter capacitor C. dc The positive terminal of transistor S2 is connected to the positive terminal of transistor S4; the emitter of transistor S2 is connected to the emitter of transistor S4, and the connection point is then connected to the DC-side filter capacitor C. dc The negative terminal is connected; DC side filter capacitor C dc The positive terminal and the DC side filter inductor L dc One end is connected, and the negative end is connected to the DC power supply; DC side filter inductor L dc The other end is connected to the DC power supply; One end of the energy transfer inductor L is connected to the first common port, and the high-frequency transformer T... r The primary winding is connected to the other end of the energy transfer inductor L and the second common port, respectively; the secondary winding is connected to the third common port and the fourth common port, respectively; high-frequency transformer T r The primary winding is connected to the other end of the inductor L, and the secondary winding is connected to one end of the third common port, which are the same name terminals; A high dynamic characteristic power control method includes the following steps: Step 1, switch transistor S c1 collector and switching transistor S c4 The voltage difference between the collectors is defined as u p The terminal voltage of the primary winding of the high-frequency transformer is defined as u. s Define the phase voltage of phase A as: The phase voltage of phase B is The phase voltage of phase C is V m Let f be the phase voltage amplitude; define the power supply frequency as f, and one power frequency cycle as T; define the line voltage with the largest amplitude among the three-phase line voltages as u. max The second largest line voltage is located as u. med Define the second largest line voltage u within a switching cycle. med Duration and switching period T s The ratio is d m Define u s and u p The ratio of the phase difference to π / 2 is the phase shift percentage δ; the phase of one power frequency cycle is 2π, and the zero-crossing point of the A-phase voltage from negative to positive is defined as the phase. The starting point is determined by the phase duration of each sector being π / 6; based on the two-wire voltage modulation, the output power P and δ of the DC-side power supply and d are determined. m Relational expressions; Step 2, input the target power P* and the grid-side active power feedback quantity P. f The difference is calculated to obtain the power error signal. After passing through the compensation network, the power error signal yields the adjustment value P*'. This adjustment value is then substituted into the binary search method for calculation to obtain δ1 and d in each switching cycle. m1 This allows the active power on the grid side to approach the target power P*, enabling feedback control. The compensation network uses PI or PID control methods. Step 3, based on P and δ and d m Substituting the relational expression with different target powers P*, we can calculate d at different target powers. m The theoretical data of δ are fitted to d m δ is the dependent variable, and phase is the dependent variable. The expression with the target power P* as the independent variable is used as the fitting calculation formula for feedforward control; Step 4: Set sector N, target power P*, and phase. Substituting into the fitting formula for feedforward control, we obtain δ2 and d. m2 ; δ1, d m1 and δ2 and d respectively m2 Summing, we get δ3 and d. m3 To control the duration of the maximum and second-largest line voltages during the switching cycle and u s u p The phase difference is used to complete the high dynamic characteristic power control by adding feedforward control to feedback control.

2. The high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter according to claim 1, characterized in that, Based on the direction of energy flow in the converter, power flow from the AC side to the DC side is defined as rectification, and power flow from the DC side to the AC side is defined as inversion.

3. The high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter according to claim 1, characterized in that, Step 1: Determine the output power P and δ of the DC-side power supply, as well as d, based on the two-wire voltage modulation. m The relational expression, specifically, is as follows: 1) Divide one power frequency cycle T into 12 sectors on average. Define sector 1 as e a >e b >e c And e b <0; Define sector 2 as e a >e b >e c And e b >0; Define sector 3 as e b >e a >e c And e a >0; Define sector 4 as e b >e a >e c And e a <0; Define sector 5 as e b >e c >e a And e c <0; Define sector 6 as e b >e c >e a And e c >0; Define sector 7 as e c >e b >e a And e b >0; Define sector 8 as e c >e b >e a And e b <0; Define sector 9 as e c >e a >e b And e a <0; Define sector 10 as e c >e a >e b And e a >0; Define sector 11 as e a >e c >e b And e c >0; Define sector 12 as e a >e c >e b And e c <0; 2) Determine the maximum and second-largest line voltages in each sector. The maximum and second-largest line voltages in each sector are shown in Table 1. Table 1. Distribution of maximum and second-maximum line voltages in 112 sectors. 3) Solve for P, δ, and d. m relational expressions According to the two-wire voltage modulation strategy, P and d in each sector m The relationship between δ and δ is consistent, and can be expressed as: Among them, V dc It is the DC-side output voltage, f s It is the switching frequency, and the reference target average value of the B-phase current is equal to the average current of the B-phase, that is... The target power is equal to the average power on the DC side, i.e., P * =P, where n is the ratio of the number of turns in the primary winding to the number of turns in the secondary winding of the high-frequency transformer.

4. The high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter according to claim 1, characterized in that, Step 2, input the target power P* and the grid-side active power feedback quantity P. f The difference is calculated to obtain the power error signal. After passing through the compensation network, the power error signal yields the adjustment value P*'. This adjustment value is then substituted into the binary search method for calculation to obtain δ1 and d in each switching cycle. m1 This allows the active power on the grid side to approach the target power P*, achieving feedback control. The specific process of the bisection method is as follows: S1: Initialize the loop count i = 0, the maximum loop count k, and the minimum value of δ. min =0, δ maximum value δ max =0.5; S2:calculationδ=(d min +d max ) / 2; S3: Check if the loop count i is greater than k. If it is, output d directly. m (δ), otherwise proceed to S4; S4: Determine if P(δ) is greater than P*'. If P(δ) is greater than P*', then update δ. max =δ, otherwise update δ min =δ; S6: i = i + 1; S7: Jump to S2 and proceed to the next iteration until the iteration count k is met, at which point δ and d are obtained. m The value of (δ).

5. The high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter according to claim 4, characterized in that, The grid-side current i in the three-phase stationary coordinate system a i b and i c Perform a coordinate transformation to obtain the d-axis component I in the dq synchronous rotating coordinate system. d , representing the active component of the grid-side current; the grid-side voltage e in the three-phase stationary coordinate system. a e b and e c Perform a coordinate transformation to obtain the d-axis component V in the dq synchronous rotating coordinate system. d , representing the active component of the grid-side voltage; multiplying the voltage and current components on the d-axis yields P. f .

6. The high dynamic characteristic power control method applied to a dual active full-bridge three-phase bidirectional AC / DC converter according to claim 3, characterized in that, Step 3, the circuit operating phase in sector 1 is in Based on P, δ, and d m Substituting the relational expression into different target powers P*, we can calculate the d in sector 1 at different target powers. m The theoretical data of δ are fitted to d m δ is the dependent variable, and phase is the dependent variable. The expression for the target power P* as the independent variable is used as the fitting calculation formula for the feedforward control. The specific method is as follows: Due to the use of two-wire voltage modulation, δ and d in each sector m The data are correlated, so we only need to solve the fitting calculation formula for the feedforward control of sector 1. The δ and d values ​​for the other sectors can then be calculated using the fitting formula for sector 1. m The specific process is as follows: S1. The current sector is N. This represents the current operating phase of the circuit. S2. Determine if sector N is odd; S3. If the sector is an odd number of sectors, then the phase... The formula for converting to sector 1 phase is: S4. If the sector is an even number of sectors, then the phase... First convert to sector 2, then convert to sector 1. The formula is: Will Converting to sector 2, the formula is: Will Converting to sector 1, the formula is:

7. A high dynamic characteristic power control system applied to a dual active full-bridge three-phase bidirectional AC / DC converter, characterized in that, Implement the high dynamic characteristic power control method for a dual active full-bridge three-phase bidirectional AC / DC converter as described in any one of claims 1-6 to achieve high dynamic characteristic power control of the dual active full-bridge three-phase bidirectional AC / DC converter.

8. A computer device, comprising a memory, a processor, and a computer program stored in the memory and executable on the processor, wherein when the processor executes the computer program, it implements the high dynamic characteristic power control method for a dual active full-bridge three-phase bidirectional AC / DC converter as described in any one of claims 1-6, thereby realizing high dynamic characteristic power control of the dual active full-bridge three-phase bidirectional AC / DC converter.

9. A computer-readable storage medium having a computer program stored thereon, wherein when the computer program is executed by a processor, it implements the high dynamic characteristic power control method for a dual active full-bridge three-phase bidirectional AC / DC converter as described in any one of claims 1-6, thereby realizing high dynamic characteristic power control of the dual active full-bridge three-phase bidirectional AC / DC converter.

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