Bidirectional isolation converter

Abstract

The invention relates to the technical field of power electronics and power conversion, and discloses a bidirectional isolation converter which comprises a direct current-alternating current conversion module, an alternating current-direct current conversion module, a controller and an isolation transformer, the converter adopts a dual-active-bridge topological structure, bidirectional transmission and isolation of power are achieved through a soft switching technology, and the isolation transformer is connected with the controller. Soft switching techniques include zero voltage switching and zero current switching. According to the invention, through simplifying the control strategy, the SPS control mode controls the flow of the current by adjusting the phase difference, so that the energy conversion is realized. The control mode is relatively simple and easy to implement, the control complexity is reduced through SPS control, and the workload of design and debugging is reduced; and the efficiency is improved: under the SPS control, the dual-active bridge DC-DC converter can effectively improve the efficiency of the system on the premise of basically not increasing the current stress. Through a reasonable control strategy, the efficiency of an experimental prototype during transmission of 6kW power is absolutely increased by 0.83%, and the current stress is only increased by 0.5 A.

Description

Technical Field

[0001] The present invention relates to the technical fields of power electronics and power conversion, and more particularly, to a bidirectional isolation converter. Background Art

[0002] In the field of power electronics, DC-DC converters are important devices for realizing DC voltage regulation. The dual-active-bridge isolated DC-DC converter has received extensive attention due to its superior electrical performance and design flexibility. Compared with traditional non-isolated DC-DC converters, the dual-active-bridge isolated design has the following several remarkable advantages:

[0003] Bidirectional power transfer: The dual-active-bridge structure supports bidirectional energy transfer and can achieve bidirectional flow of electrical energy, which is particularly important in applications such as electric vehicle charging and discharging, and renewable energy systems. By adjusting the control of the switches, the energy flow can be flexibly managed to improve the overall energy efficiency of the system.

[0004] Electrical isolation: The isolation characteristic effectively prevents direct contact between high voltage and low voltage, enhancing the safety and reliability of the system. This characteristic is of great significance for applications requiring electrical isolation, such as medical devices and high-voltage industrial equipment.

[0005] High-efficiency conversion: This design allows operation within a wide range of input / output voltage while ensuring high efficiency, helping to reduce energy losses. In modern power management and renewable energy integration, the demand for high-efficiency converters is increasing.

[0006] Compatibility and flexibility: The dual-active-bridge architecture supports various control strategies, such as phase-shift modulation and frequency modulation, enabling designers to optimize according to specific application requirements.

[0007] The above dual-active-bridge isolated DC-DC converter also has the following defects:

[0008] (1) The DAB converter using the traditional control strategy, the single-phase-shift (SPS) method, may not meet the requirements of transient response;

[0009] (2) Control complexity: In different load conditions, the traditional dual-phase-shift (DPS) control strategy may need to be further optimized to reduce the reverse power and improve the system transmission efficiency. Although the three-phase-shift (TPS) control can further reduce the reverse power, due to the existence of three free variables, the control is relatively complex and the system dynamic response is affected;

[0010] (3) Since the DC bus voltage may vary due to various reasons, such as sudden line faults, voltage sags, and surges, in the case of a DC connection system, the DC bus voltage may fluctuate due to changes in the irradiance of the PV system, thus hindering the performance of the DAB converter. Summary of the Invention

[0011] In order to overcome the deficiencies of the prior art, the present invention provides a bidirectional isolation converter to solve the problems raised in the above-mentioned background art.

[0012] To achieve the above object, the present invention provides the following technical solution: a bidirectional isolation converter, including a DC-AC conversion module, an AC-DC conversion module, a controller, and an isolation transformer. The converter adopts a dual-active-bridge topology structure to achieve bidirectional power transmission and isolation through soft-switching technology, and the soft-switching technology includes zero-voltage switching (ZVS) and zero-current switching (ZCS).

[0013] Preferably, the controller performs peak current control through a feed-forward loop to suppress load current disturbances and input voltage fluctuations, thereby enhancing the stability and dynamic response performance of the output voltage. The feed-forward loop uses the input voltage and load current as control signals to achieve fast dynamic response and efficient power regulation.

[0014] Preferably, the controller adopts a single-phase shift (SPS) control strategy to control the direction and magnitude of power by adjusting the phase shift angle between the main bridge and the secondary bridge. The SPS control strategy realizes zero-voltage switching (ZVS) in a wide operating range by improving logical operations and optimizing control algorithms, thereby improving system efficiency.

[0015] Preferably, the isolation transformer adopts a high-frequency EE-type magnetic core structure, and its transformer turns ratio is designed according to the input and output voltage ranges to adapt to different load requirements. The leakage inductance of the transformer is optimized to reduce energy losses during high-frequency switching and improve the overall system efficiency.

[0016] Preferably, the DC bus voltage range of the converter is 36V to 60V, the output voltage is 400V, and the switching frequency is 50kHz, which can adapt to different input and output voltage change conditions.

[0017] Preferably, the controller adopts an STM32F334C8T6 single-chip microcomputer to generate accurate PWM signals through an integrated high-resolution timer for driving the main power circuit.

[0018] Preferably, the converter further includes a protection circuit module, which immediately turns off the power switch tube to protect the circuit and the load when overcurrent or overvoltage conditions are detected.

[0019] Preferably, both the DC-AC conversion module and the AC-DC conversion module use MOS transistors as the main switching devices. The primary switching device is selected as BSC123N10LS, and the secondary switching device is selected as IRFP460L to adapt to different voltage levels and power requirements.

[0020] Preferably, the converter has good adaptability in renewable energy applications such as photovoltaic systems. By optimizing the control strategy, the adaptability to the DC bus voltage fluctuation is effectively improved.

[0021] Preferably, the controller designs a type compensator control strategy. By adjusting the zeros and poles of the transfer function, fast dynamic response and system stability are achieved.

[0022] Compared with the prior art, the beneficial effects of the present invention are as follows:

[0023] (1) Simplify the control strategy: The SPS control method controls the flow of current by adjusting the phase difference, thereby realizing the conversion of energy. This control method is relatively simple and easy to implement. Compared with the more complex TPS (triple phase shift) control, the SPS control reduces the control complexity and the workload of design and debugging;

[0024] (2) Improve efficiency: Under SPS control, the dual active bridge DC-DC converter can effectively improve the system efficiency without significantly increasing the current stress. Through a reasonable control strategy, the efficiency of the experimental prototype can be absolutely increased by 0.83% when transmitting 6 kW of power, while the current stress only increases by 0.5 A;

[0025] (3) Reduce costs: Due to the simplicity of the SPS control method, the demand for high-performance control devices can be reduced, thereby reducing costs. The complexity of the control method affects the workload of design and debugging, and thus affects the cost of the product;

[0026] (4) Provide a more advanced control strategy: The load current and input voltage are used as feedforward signals in the control loop to adapt to rapidly changing load and system requirements;

[0027] (5) Improve the adaptability of the DAB converter to the DC bus voltage fluctuation. Especially in renewable energy applications such as photovoltaic systems, compared with the prior art, it can better maintain performance and efficiency. Description of the Drawings

[0028] Figure 1 It is the circuit structure diagram of the DAB converter of the present invention;

[0029] Figure 2 It is the simplified circuit structure diagram of the DAB converter;

[0030] Figure 3 Schematic diagram of the working range a of ZVS of the present invention;

[0031] Figure 4 Schematic diagram of the working range b of ZVS of the present invention;

[0032] Figure 5 Bode plot of the power stage transfer function of the DAB converter of the present invention;

[0033] Figure 6 Step response diagram of the control-to-output transfer function of the present invention;

[0034] Figure 7 Bode plot of the power stage transfer function of the DAB converter of the present invention;

[0035] Figure 8 Bode plot of the system after introducing the Type II controller of the present invention;

[0036] Figure 9 Bode plot of the system after introducing gain of the present invention;

[0037] Figure 10 PWM drive waveform diagram of phase-shift control of the present invention;

[0038] Figure 11 System and control architecture diagram of the DAB converter of the present invention;

[0039] Figure 12 Peak current mode control diagram of the DAB converter based on the power reference feedforward function of the present invention;

[0040] Figure 13 Circuit diagram of the 15V auxiliary voltage of the present invention;

[0041] Figure 14 Circuit diagram of the 10V auxiliary voltage of the present invention;

[0042] Figure 15 Circuit diagram of the 3.3V auxiliary voltage of the present invention;

[0043] Figure 16 Circuit diagram of the drive of the DC-AC part of the present invention;

[0044] Figure 17 Circuit diagram of the drive of the AC-DC part of the present invention;

[0045] Figure 18 Main power circuit diagram of the DC-AC part of the present invention;

[0046] Figure 19 Main power circuit diagram of the AC-DC part of the present invention. Detailed implementation

[0047] The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all the embodiments. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts shall fall within the protection scope of the present invention.

[0048] As Figures 1 to 19 shown, the present invention provides a bidirectional isolation converter, which includes a DC-AC conversion module, an AC-DC conversion module, a controller, and an isolation transformer. The converter adopts a dual-active-bridge topology structure, and realizes bidirectional power transmission and isolation through soft-switching technology. The soft-switching technology includes zero-voltage switching (ZVS) and zero-current switching (ZCS);

[0049] The dual-active-bridge topology structure and soft-switching technology are adopted to improve efficiency and reduce switching losses. Through zero-voltage switching (ZVS) and zero-current switching (ZCS) technologies, the voltage and current stresses during switching are reduced, electromagnetic interference is lowered, and the overall efficiency of the system is improved.

[0050] Among them, the controller performs peak current control through a feedforward loop to suppress load current disturbances and input voltage fluctuations, thereby enhancing the stability and dynamic response performance of the output voltage. The feedforward loop uses the input voltage and load current as control signals to achieve fast dynamic response and efficient power regulation;

[0051] By controlling through the feedforward loop to suppress load current disturbances and input voltage fluctuations, the transient response performance of the system is improved to meet the requirements of rapidly changing loads and the system.

[0052] Among them, the controller adopts a single-phase-shift (SPS) control strategy. By adjusting the phase-shift angle between the main bridge and the secondary bridge, the direction and magnitude of the power are controlled. The SPS control strategy realizes zero-voltage switching (ZVS) within a wide operating range by improving logical operations and optimizing control algorithms, thereby improving the system efficiency;

[0053] By the single-phase-shift (SPS) control strategy, the direction and magnitude of power transmission are accurately controlled, and zero-voltage switching (ZVS) is realized within a wide operating range, reducing the reverse power and optimizing the system transmission efficiency under different load conditions.

[0054] Among them, the isolation transformer adopts a high-frequency EE-type magnetic core structure. Its transformer turns ratio is designed according to the input and output voltage ranges to adapt to different load requirements. The leakage inductance of the transformer is optimized to reduce energy losses during high-frequency switching and improve the overall efficiency of the system;

[0055] Improve the overall system efficiency and reduce energy losses during high-frequency switching by optimizing the transformer design.

[0056] Among them, the DC bus voltage range of the converter is from 36V to 60V, the output voltage is 400V, and the switching frequency is 50kHz, which can adapt to different input and output voltage change conditions.

[0057] Among them, the controller uses the STM32F334C8T6 single-chip microcomputer to generate accurate PWM signals through an integrated high-resolution timer for driving the main power circuit;

[0058] Improve the control accuracy of the power switch tube and achieve more efficient power conversion.

[0059] Among them, the converter further includes a protection circuit module. When overcurrent or overvoltage conditions are detected, the power switch tube is immediately turned off to protect the circuit and the load;

[0060] Improve the safety and reliability of the system and prevent damage caused by abnormal conditions.

[0061] Among them, both the DC-AC conversion module and the AC-DC conversion module use MOS tubes as the main switching devices. The primary switching device selects BSC123N10LS, and the secondary switching device selects IRFP460L to adapt to different voltage levels and power requirements;

[0062] Improve the power handling capacity and voltage adaptability of the converter to ensure the stable operation of the converter under various conditions.

[0063] Among them, the converter has good adaptability in renewable energy applications such as photovoltaic systems. By optimizing the control strategy, the adaptability to DC bus voltage fluctuations is effectively improved.

[0064] Among them, the controller designs a type compensator control strategy. By adjusting the zeros and poles of the transfer function, fast dynamic response and system stability are achieved;

[0065] By adjusting the zeros and poles of the transfer function, improve the dynamic performance and stability of the system to meet the high-performance control requirements.

[0066] 1. System topology selection

[0067] The system topology requires bidirectional conversion and isolation topology, with an input voltage range of 36Vdc-60Vdc, an output voltage of 400Vdc, and a rated output power of 1KW. Common isolated bidirectional DC / DC converters include forward bidirectional DC / DC converters, flyback bidirectional DC / DC converters, push-pull bidirectional DC / DC converters, half-bridge bidirectional DC / DC converters, and full-bridge bidirectional DC / DC converters. Forward and flyback bidirectional DC / DC converters have simple structures and relatively low costs, and are more suitable for use in low-power applications. Push-pull bidirectional DC / DC converters are more suitable for medium- and low-voltage high-power applications. Half-bridge bidirectional DC / DC converters are more suitable for use in low-power applications due to their own high stress characteristics. The switch tube voltage and current stress of the full-bridge bidirectional DC / DC converter are relatively small, and are more suitable for use in high-power applications.

[0068] In addition to the above-mentioned bidirectional DC / DC converter, there is also a dual active bridge bidirectional DC / DC converter (DAB). This converter consists of a primary and secondary bridge circuit, a high-frequency transformer, capacitors and inductors. Figure 1 The full-bridge DAB converter shown has an increased number of switch tubes compared to the half-bridge DAB converter, but a more flexible control method, which is suitable for disturbances in a wide input range and a wide load range.

[0069] 2. DAB converter design parameters

[0070] The main simulation electrical parameters of the dual active bridge DC / DC converter are shown in Table 1

[0071] Table 1 Main simulation electrical parameters of dual active bridge DC / DC converter

[0072]

[0073] 3. DAB converter circuit model

[0074] The conduction rule of the four switches Q1-Q8 on the primary and secondary sides is that the two switches on the same bridge arm are complementary, and the two switches on the diagonal of the H bridge are turned on at the same time. In this way, the voltage at the midpoint of the primary and secondary bridge arms is an AC square wave voltage. The square wave voltage on the secondary side is converted to the primary side of the transformer. At this time, the circuit structure of the DAB converter can be simplified as follows Figure 2 The equivalent circuit of two AC square wave voltage sources added to both sides of the inductor L is shown. The duty cycle of the switch tube is constant at 50% and a dead zone is left to prevent the upper and lower bridge arms from being turned on at the same time and causing a short circuit. The direction and magnitude of power transmission can be controlled by controlling the phase difference (phase shift angle δ) of the two AC square wave voltages of the original and secondary bridge arms. The ratio of the phase shift angle δ to π is defined as the shift to the duty cycle

[0075]

[0076] 3.1 Average and RMS Values of the Primary Current of the DAB Converter

[0077]

[0078] 3.2 ZVS Operating Range of the DAB Converter

[0079] Define the voltage transfer ratio (voltage gain) M

[0080]

[0081] where M is the voltage transfer ratio (voltage gain), V o is the output voltage, V i is the input voltage, and n is the transformer turns ratio.

[0082] To achieve zero-voltage turn-on, the conditions that need to be satisfied are that the diode conducts before the switch is turned on, and the current maintains the original direction of flow, i.e., I 1 > 0, I 2 > 0.

[0083] Combining the current analysis in 3.2 and the voltage transfer ratio M, the inductor current can be expressed in terms of the phase-shift ratio and the voltage transfer ratio:

[0084]

[0085]

[0086] Let 2DM + 1 - M > 0, 2D + M - 1 > 0, and the range of the phase-shift ratio D can be obtained:

[0087]

[0088] Use Matlab to plot the operating range that satisfies ZVS (as shown in Figure 3 ). Only when M = 1 can ZVS be achieved over the full load range. Therefore, a suitable transformer turns ratio n can be selected during design so that the DAB converter can also achieve ZVS over a wide operating range.

[0089] 3.3 High-Frequency Transformer Turns Ratio and Inductance Value of the DAB Converter

[0090] The design specifications are a rated input voltage of 40V and a rated output voltage of 400V. Let the voltage transfer ratio M = 1, and calculate the transformer turns ratio n = 10. Determine the voltage transfer ratio M according to the range of the input voltage.

[0091]

[0092] where M 1, M 2 is the voltage transfer ratio, V o : output voltage, V i_nin , V i_max are the minimum input voltage and the maximum input voltage respectively, and n is the transformer turns ratio.

[0093] According to the working range of ZVS and M 1 and M 2 values to determine the intersection points D zvs1 and D zvs2 . As Figure 4 shown, D zvs1 = 0.05, D zvs2 = 0.35.

[0094] Let it can be obtained that:

[0095]

[0096] where M is the voltage transfer ratio, V out : output voltage, I out : output current, f s : switching frequency, D: phase shift ratio, n: transformer turns ratio.

[0097] When the DAB converter works under half - load conditions, I out = 1.25 A, and the solution is:

[0098] K = 23.3918, and the minimum value of the inductor can be obtained as 1.368 μH and the maximum value as 3.276 μH.

[0099] 3.4 DAB Converter Switching Model

[0100] Analyze the primary - side active half - bridge of the DAB converter. There are only 2 switching states:

[0101] When S 1 = 0, V A = - V in , when S 1 = 1, V A = + V in ;

[0102] When S 2 = 0, V B = - V in , when S 2 = 1, V B = + V in ;

[0103] Analyze the primary - side active full - bridge of the DAB converter. There are a total of 4 switching states:

[0104] When S 1 = 0, S 2 = 0, V A = -V in , V B = -V in , V 1 = 0;

[0105] When S 1 = 0, S 2 = 1, V A = -V in , V B = +V in , V 1 = -2V in ;

[0106] When S 1 = 1, S 2 = 0, V A = +V in , V B = -V in , V 1 = 2V in ;

[0107] When S 1 = 1, S 2 = 1, V A = +V in , V B = +V in , V 1 = 0.

[0108] The functional relationship between the output voltage V1 of the primary-side active bridge and the switching logic variables of the switching transistors Q1 - Q4 can be obtained as follows:

[0109] V 1 = 2V in {S 1 -S 2}#(2.3)

[0110] Express the switching function as a function related to time:

[0111] S k (t) = {0,1} k = 1,2,…,8#(2.4)

[0112] Similarly, analyzing the secondary-side full bridge, the output voltages at the midpoints of the primary and secondary bridge arms can be expressed as:

[0113] V 1 = 2V in {S 1 -S 2}#(2.5)

[0114] V 2 = 2V out {S 5 -S 6}#(2.6)

[0115] 3.5DAB Converter Small-Signal Model

[0116] The voltage and current relationship of the output capacitor is as follows:

[0117]

[0118] At the secondary output side, according to KCL:

[0119] i c (t) = i DC (t) - i r (t)#(2.8)

[0120] where i DC (t) is the DC current output by the secondary bridge arm, i r (t) is the load resistance, i c (t) is the capacitor current.

[0121] The DC current output by the secondary bridge arm is related to the switching state of the secondary bridge arm switching tube and the inductor current i L is relevant.

[0122] 4DAB Converter Controller Design

[0123] 4.1DAB Converter Transfer Function

[0124] The main simulation electrical parameters of the dual active bridge DC / DC converter are shown in Table 2.

[0125] Table 2 Main Simulation Electrical Parameters of Dual Active Bridge DC / DC Converter

[0126]

[0127] The transfer function from control to output obtained from the DAB converter small-signal model is:

[0128]

[0129] Use Matlab to plot the Bode plot of the open-loop transfer function of the DAB converter, as Figure 5 shown. It can be seen that this is a self-stable system, but the cut-off frequency is much higher than the switching frequency. Therefore, a controller needs to be introduced to correct the system.

[0130] Then plot the step response of the transfer function from control to output as Figure 6As shown. The step response of the system indicates that the system is a stable system without oscillation.

[0131] 4.2 DAB Converter Type Ⅱ Compensator Design

[0132] Substituting the design parameters in Table 1 into G(s), we can get:

[0133]

[0134] The transfer function of the Type Ⅱ compensator is:

[0135]

[0136] where K g is the gain coefficient, ω p , ω z are zero-pole designs.

[0137] For the system to have a fast dynamic response, the crossover frequency is usually selected as Therefore, the crossover frequency is selected as 5 kHz and the phase margin is 45°. From Figure 7 it can be seen that the phase at the crossover frequency of 5 kHz is -1.73°, and the Type Ⅱ compensator has a built-in phase reduction of -90°, so the phase to be increased is:

[0138] boost = 45° - [(-1.73° - 90°) + 180°] = -43.27°

[0139] Use the K coefficient method to determine the positions of the zero and pole:

[0140] K = tan(45° + boost / 2) = 0.432

[0141] The positions of the zero and pole can be determined as:

[0142] ω c = 31416 rad / s, ω p = 13572 rad / s, ω z = 72722 rad / s

[0143] First, assume the gain K g = 1, then the transfer function of the Type Ⅱ compensator can be expressed as:

[0144]

[0145] Use Matlab to draw the Bode plot of G(s)H(s) as Figure 8 shown. It can be seen that the phase margin at the crossover frequency is 45° and the amplitude is -83.6 dB which is not zero. Calculate K gAdjust the value to the total gain, raise the phase by 83.6 dB, and solve for K g = 15137, redraw the Bode plot of G(s)H(s) as Figure 9 shown. It can be seen that the amplitude is -0.0232 dB at 5 kHz and the phase margin is 45°, meeting the design requirements.

[0146]

[0147] 4.3 Discretization of the Type II compensator for the DAB converter

[0148] The expression of the discretized controller is:

[0149]

[0150] STM32F334C8T6 realizes SPS single-phase shift:

[0151] The PWM drive waveform of the phase-shift control is as Figure 10 shown. Q1 and Q2 are complementary in driving and a dead zone is inserted; Q3 and Q4 are complementary in driving and a dead zone is inserted. The duty cycle of all PWM waveforms is fixed (50%), and Q3 / Q4 is phase-shifted relative to Q1 / Q2, and the phase magnitude is determined by the control loop Type II compensator.

[0152] STM32F334 integrates a high-precision Timer (HRTIM) internally, which can be frequency-multiplied up to 32 times. This enables the timer to have a high resolution even when operating at a high frequency. Turn on the Master Timer as the clear synchronization signal for the four sub-timers. For example, to achieve a 30° phase shift, the update event of the Master Timer can be set as the count reset signal for Sub-timer A, and at this time, the phase shift is set to 0 degrees; the comparison value 1 of the Master Timer is used as the count reset signal for Sub-timer B. If the set phase shift is 30 degrees and the frequency of the Master Timer is set to 50 kHz, the repetition count value of the Master Timer should be 20480 at this time, that is, 20480 corresponds to a 360° phase shift, then setting the comparison value 1 to 1706 can achieve a 30° phase shift. The DAB converter system and control architecture are as Figure 11 shown. During forward transmission, the voltage at the output end is collected and compared with the target voltage, and the angle of the required phase shift is calculated through the Type II compensator, and the waveform is generated and controlled through the high-resolution timer.

[0153] Adopt a new feedforward strategy to achieve peak current mode control. The reference of the peak current is generated by the reference formulas of the input power and the terminal voltage.

[0154] 5. MCU selection

[0155] This solution needs to consider generating high-frequency PWM signals to drive MOS transistors, and the PWM needs to meet requirements such as high enough precision, low power consumption, and energy saving. The requirements for MCU selection are as follows: Provide multiple energy-saving modes and power consumption optimization options, suitable for applications that need to run for a long time or are battery-powered. Therefore, the STM32 series of single-chip microcomputers from ST company are selected.

[0156] STM32F1 series: Cortex-M3 core, 72MHz main frequency, no FPU, low cost, no HRTIM.

[0157] STM32F3 series: Cortex-M4F core, 72MHz main frequency, has FPU, and some models have HRTIM.

[0158] STM32F4 series: Cortex-M4F core, up to 168MHz main frequency, has FPU, no HRTIM.

[0159] STM32G4 series: Cortex-M4F core, 170MHz main frequency, has FPU, higher cost, and some models have HRTIM.

[0160] Combined with the requirements, STM32F334 is used as the MCU of the operating system control board for the bidirectional DC isolation converter.

[0161] 6. Selection of auxiliary circuits

[0162] The auxiliary circuit is mainly divided into three voltage levels: 15V, 10V, and 3.3V;

[0163] (1) 15V part:

[0164] Solution 1: LT1676 (Vmax: 60V, Imax: 0.46A, switching frequency: 100kHz.)

[0165] Solution 2: XL7005A (Vmax: 80V, Imax: 0.4A, switching frequency: 150kHz.)

[0166] This design is a digital power supply. In order to better meet the hard condition that the maximum Vcc of the experiment is 60V, after calculation and estimation, the current we need is about 350mA. Solution 2 is selected here. See Figure 13 .

[0167] (2) 10V part:

[0168] Solution 1: LT1676 (Vmax: 60V, Imax: 0.46A, switching frequency: 100kHz.)

[0169] Plan 2: XL7005A (Vmax: 80V, Imax: 0.4A, switching frequency: 150kHz.)

[0170] This design is a digital power supply. To better meet the hard condition that the maximum Vcc of the experiment is 60V, after calculation and estimation, the current we need is about 350mA. Here, Plan 2 is selected. See Figure 14 .

[0171] (3) 3.3V part:

[0172] Plan 1: LP6460A (Vmax: 16V, I 静 : 400uA, switching frequency: 600kHz.)

[0173] Plan 2: SY8120B (Vmax: 18V, I 静 : 450uA)

[0174] Combined with the experimental requirements and cost considerations, adopting Plan 2 as the chip for this 3.3V part is more in line with the experimental requirements. See Figure 15 .

[0175] 7. Selection of drive circuit

[0176] Plan 1: Characteristics of IR2104: It has independent high and low side inputs, built-in logic inverter, and can provide low and high level drive signals; the gate drive supply voltage range is 10 to 20V, with built-in dead time, the high side output is in phase with the HIN input, the shutdown input will turn off both channels, and the two channels have a matching propagation delay.

[0177] Plan 2: Characteristics of EG3112: The highest frequency supports 500kHz; the low-end VCC voltage range is 2.8V - 20V; the output current capacity Io + / - 2A / 2.5A; it has functions such as overcurrent protection; few peripheral devices; the static current is less than 1μA, which is very suitable for battery applications.

[0178] Although EG3112 is relatively new and the supply situation may not be as stable as IR2104, due to the efficiency requirements of the topic, to improve efficiency, it is necessary to minimize power loss as much as possible. Here, Plan 2 is selected. The circuit design details can be seen in Figure 16 、 17 .

[0179] 8. Selection of main power circuit

[0180] The bidirectional DC isolation converter is a circuit used to achieve bidirectional energy transfer in a DC power supply system. Its main power circuit usually includes two full-bridge converters, one for forward conversion and the other for reverse conversion.

[0181] Select the appropriate N-channel MOSFET of BSC123N10LS and IRFP460 according to the requirements.

[0182] 9. Selection of Switching Devices

[0183] The maximum voltage on the primary side of the converter is 60V, and the maximum voltage on the secondary side is 400V at the maximum transmission power. Considering the overshoot in the transient process, at least a one-fold margin is left when selecting the switching device. Considering the requirements of the hardware circuit design and the product parameters in the market comprehensively, finally, the Mos device of BSC123N10LS is selected as the primary-side switching device, with a voltage rating of 100V and a current of 80A; the Mos device of IRFP460L is selected for the secondary side, with a voltage rating of 500V and a current of 20A.

[0184] 10. Design of High-Frequency Transformer

[0185] The parameter design and selection of the high-frequency transformer are the key points of the hardware design of the DAB converter, which need to meet multiple requirements such as transmission power, voltage rating, operating frequency, and magnetic flux density of the magnetic core. At the same time, it is necessary to maximize the power density and reduce the volume and mass. The transformer design mainly includes: the selection of magnetic core materials, structures, types, as well as the selection of the number of turns and wire types of the coil.

[0186] 10.1 Selection of Magnetic Core Materials

[0187] Currently, the main magnetic core materials for transformers are ferrite materials and amorphous and nanocrystalline materials. Among them, ferrite materials have low losses at medium and high frequencies, are cheap, and are easy to be designed into various shapes. However, they have the disadvantages of low saturation magnetic flux density and low magnetic flux rate. On the contrary, amorphous and nanocrystalline materials have good comprehensive magnetic properties, but are expensive and not easy to design the outer shape. Considering comprehensively, manganese-zinc ferrite is selected as the magnetic core material of the transformer.

[0188] 10.2 Selection of Magnetic Core Structures

[0189] The selection criteria for magnetic core structures mainly include: whether it can reduce the leakage inductance of the high-frequency transformer, whether it can increase the heat dissipation area of the winding coil, and whether it is conducive to winding and assembly wiring. Common magnetic core structures include EI type, EE type, EC type, Q type, PO type, etc. Among them, the EE type structure has the advantages of large window, good heat dissipation, simple and mature manufacturing process, and low cost, and has become the most commonly used magnetic core structure in high-frequency transformers. The EE type magnetic core structure is selected for this design.

[0190] 10.3 Selection of Magnetic Core Types

[0191] After determining the magnetic core materials and structures, it is necessary to further determine parameters such as the size of the window, that is, the magnetic core type. The AP method is used in this article to determine the magnetic core type of the transformer.

[0192] The equivalent power P of the high-frequency transformer can be obtained from the basic parameters of the DAB converter:

[0193]

[0194] where P T is the apparent power, P is the transformer power, and η is the converter efficiency.

[0195] Among them, η is the designed efficiency of the converter, and η = 95% is selected in this design. The product of the core area reflects the power transmission ability of the high-frequency transformer. The expression of the core area is:

[0196]

[0197] where A e and A w are the window area and cross-sectional area of the core respectively, P T is the apparent power, K 0 is the window utilization factor of the transformer, K f is the waveform factor, f s is the switching frequency, B w is the magnetic flux density, and K j is the coefficient of the current density.

[0198] It should be noted that in this article, relational terms such as first and second are only used to distinguish one entity or operation from another entity or operation, and do not necessarily require or imply any actual relationship or order between these entities or operations. Moreover, the term "comprising", "including" or any other variant thereof is intended to cover non-exclusive inclusion, so that a process, method, article or device comprising a series of elements not only includes those elements, but also includes other elements not expressly listed, or also includes elements inherent to such process, method, article or device.

[0199] Although the embodiments of the present invention have been shown and described, for those of ordinary skill in the art, it can be understood that various changes, modifications, substitutions and variations can be made to these embodiments without departing from the principles and spirit of the present invention. The scope of the present invention is defined by the appended claims and their equivalents.

Claims

1. A bidirectional isolation converter, comprising a DC-AC conversion module, an AC-DC conversion module, a controller and an isolation transformer, characterized in that: The converter adopts a dual active bridge topology structure and realizes bidirectional power transmission and isolation through soft switching technology, and the soft switching technology includes zero voltage switching (ZVS) and zero current switching (ZCS).

2. The bidirectional isolation converter according to claim 1, characterized in that: The controller performs peak current control through a feedforward loop to suppress load current disturbances and input voltage fluctuations, thereby enhancing the stability and dynamic response performance of the output voltage. The feedforward loop uses the input voltage and load current as control signals to achieve fast dynamic response and efficient power regulation.

3. The bidirectional isolation converter according to claim 1, characterized in that: The controller adopts a single phase shift (SPS) control strategy to control the direction and size of power by adjusting the phase shift angle between the main bridge and the auxiliary bridge. The SPS control strategy realizes zero voltage switching (ZVS) in a wide working range by improving logical operations and optimizing control algorithms, thereby improving system efficiency.

4. The bidirectional isolation converter according to claim 1, characterized in that: The isolation transformer adopts a high-frequency EE-type magnetic core structure, and its transformer ratio is designed according to the input and output voltage ranges to adapt to different load requirements. The leakage inductance of the transformer is optimized to reduce energy loss during high-frequency switching and improve the overall efficiency of the system.

5. The bidirectional isolation converter according to claim 1, characterized in that: The DC bus voltage range of the converter is 36V to 60V, the output voltage is 400V, and the switching frequency is 50kHz, which can adapt to different input and output voltage change conditions.

6. The bidirectional isolation converter according to claim 1, characterized in that: The controller adopts the STM32F334C8T6 single-chip microcomputer and realizes accurate PWM signal generation by integrating a high-resolution timer for driving the main power circuit.

7. The bidirectional isolation converter according to claim 1, characterized in that: The converter further includes a protection circuit module, which immediately turns off the power switch tube to protect the circuit and the load when an over-current or over-voltage condition is detected.

8. The bidirectional isolation converter according to claim 1, characterized in that: The DC-AC conversion module and the AC-DC conversion module both use MOS tubes as main switch devices, wherein the primary switch device is BSC123N10LS and the secondary switch device is IRFP460L to adapt to different voltage levels and power requirements.

9. The bidirectional isolation converter according to claim 1, characterized in that: The converter has good adaptability in renewable energy applications such as photovoltaic systems, and can effectively improve its adaptability to DC bus voltage fluctuations by optimizing control strategies.

10. The bidirectional isolation converter according to claim 1, characterized in that: The controller is designed with a compensator control strategy to achieve fast dynamic response and system stability by adjusting the zeros and poles of the transfer function.

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